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Power and Thermal Management Design - Sylvain LARRIBE

1. RADIATION RADIATION FROM COUPLING VIA COMMON FROM I O POWER WIRING GROUND IMPEDANCE WIRING Figure 8 70 A key point in minimizing noise problems in a design is to choose devices no faster than actually required by the application Many designers assume that faster is better fast logic is better than slow high bandwidth amplifiers are clearly better than low bandwidth ones and fast DACs and ADCs are better even if the speed is not required by the system Unfortunately faster is not better but worse where EMI is concerned Many fast DACs and ADCs have digital inputs and outputs with rise and fall times in the nanosecond region Because of their wide bandwidth the sampling clock and the digital inputs can respond to any form of high frequency noise even glitches as narrow as 1 to 3ns These high speed data converters and amplifiers are easy prey for the high frequency noise of microprocessors digital signal processors motors switching regulators hand held radios electric jackhammers etc With some of these high speed devices a small amount of input output filtering may be required to desensitize the circuit from its EMI RFI environment Adding a small ferrite bead just before the decoupling capacitor as shown in Figure 8 71 is very effective in filtering high frequency noise on the supply lines For those circuits that require bipolar supplies this technique should be applied to both positive and n
2. O 1 V O FAN2 5v ADM9240 VCCP1 O y VCCP1 NN 10ko0 45V VCCP2 MRD e O 2 Q v 901 BACKUP NT BATTERY a 74HC132 MUTO 12VIN p v 12V PROCESSOR V 470k0 12V 5VIN 45V OP295 amp NTESTIN V 2N2219A 43 3VIN O V 43 3V 0 RESET 2 5VIN O 42 5V 82kQ gt 10kQ GNDA GNDD Q Q Z Z V V Figure 7 14 In hardware monitoring circuits it is often desirable to use a high resolution low cost measurement ADC for tasks such as monitoring battery voltages during charging The AD7705 is a 16 bit sigma delta ADC with a two channel multiplexed input as shown in Figure 7 15 Key specifications are given in Figure 7 16 The AD7705 has a programmable gain amplifier which can be set for a gain of 1 to 128 The inputs and outputs are handled with a three wire serial interface The device has an on chip digital filter and a programmable output rate from 20Hz to 500Hz An application of the AD7705 as a cell monitor in a battery charging circuit is shown in Figure 7 17 7 11 HARDWARE MONITORING AD7705 16 BIT ADC BATTERY MONITOR REF IN REF IN Vpp Q CHARGE BALANCING ADC SIGMA DELTA DIGITAL MODULATOR FILTER CLOCK GENERATION REGISTER BANK AD7705 Figure 7 15 AD7705 ADC KEY SPECIFICATIONS 2 Channel Charge Balancing ADC 16 bits No Missing Codes 0 0129
3. INPUT OO CURRENT di 1A bis dt 100ns 0 Equivalent 3 5MHz ESL 20nH cere VPEAK ESLe t ESR 400mV C 100uF OUTPUT VOLTAGE Xe 0 00050 o 3 5MHz ESR 200 0 Figure 8 16 Regarding inductors Ferrites non conductive ceramics manufactured from the oxides of nickel zinc manganese or other compounds are extremely useful in power supply filters Reference 9 At low frequencies lt 100kHz ferrites are inductive thus they are useful in low pass LC filters Above 100kHz ferrites become resistive an important characteristic in high frequency filter designs Ferrite impedance is a function of material operating frequency range DC bias current number of turns size shape and temperature Figure 8 18 summarize a number of ferrite characteristics 8 23 HARDWARE DESIGN TECHNIQUES 8 24 ELECTROLYTIC CAPACITOR IMPEDANCE VERSUS FREQUENCY C 100pF ESL 20nH d REGION REGION EM LOG 121 ESR 0 20 REGION ESR 0 20 lt lt 10 2 1 2 LOG FREQUENCY Figure 8 17 FERRITES SUITABLE FOR HIGH FREQUENCY FILTERS Ferrites Good for Frequencies Above 25kHz B Many Sizes and Shapes Available Including Leaded Resistor Style Ferrite Impedance at High Frequencies Primarily Resistive Ideal for HF Filtering Low DC Loss Resistance of Wire Passing Through Ferrite is Very Low High Saturation Current
4. VIN 9v 33 33 33 HF YF Vout T ADP3000 ADJ L1 12 5pH 5V 100 4 16 100 33yF x 3 1 5817 33pF 16V L1 COILTRONICS CTX25 4 C1 C2 SPRAGUE 293D SERIES SURFACE MOUNT TANTALUM 7 8 27 8 32 HARDWARE DESIGN TECHNIQUES ADP3000 BUCK INPUT WAVEFORM a Vout 5V 100mA 60mV p p ADP3000 BUCK REG CIRCUIT C1 100uF C2 33yF 16V 20 0 5 0008 Chi e 33pF 16V x 3 VERTICAL SCALE 20mV DIV HORIZ SCALE 5us DIV C1 33uF 16V x3 SPRAGUE 293D SURFACE MOUNT TANTALUM C2 33pF 16V SPRAGUE 293D SURFACE MOUNT TANTALUM Figure 8 28 ADP3000 BUCK OUTPUT WAVEFORM Vout 5V 100mA 30mV p p ADP3000 BUCK REG CIRCUIT C1 100pF C2 33pF 16V 10 0 5 5 005 Chi 7 3 8 16 x 3 VERTICAL SCALE 20 DIV HORIZ SCALE 5ps DIV C1 33uF 16V x3 SPRAGUE 293D SURFACE MOUNT TANTALUM C2 33yF 16V SPRAGUE 293D SURFACE MOUNT TANTALUM Figure 8 29 8 33 HARDWARE DESIGN TECHNIQUES ADP3000 BUCK FILTERED OUTPUT V ADP3000 BUCK REG 3 CIRCUIT TI 10 0mv M5 00us Chi 7 3 8mV C1 100pF C2 33pF 16V S3pF 16V x 3 VERTICAL SCALE 10mV DIV HORIZ SCALE 5ys DIV C1 33uF 16V x3 SPRAGUE 293D SURFACE MOUNT TANTALUM C2 33pF 16V SPRAGUE 293D SURFACE MOUNT TANTALUM OUTPUT FILTER Lr 12 5pH COILTRONICS 25 4 Cr
5. Vout _ lin B vin iout Energy Must be Conserved Figure 3 2 Design engineers unfamiliar with IC switching regulators are sometimes confused by what exactly these devices can do for them Figure 3 3 summarizes what to expect from a typical IC switching regulator It should be emphasized that these are typical specifications and can vary widely but serve to illustrate some general characteristics Input voltages may range from 0 8 to beyond 30V depending on the breakdown voltage of the IC process Most regulators are available in several output voltage options 12V 5V 3 3V and 3V are the most common and some regulators allow the output voltage to be set using external resistors Output current varies widely but regulators with internal switches have inherent current handling limitations that controllers with external switches do not Output line and load regulation is typically about 50mV The output ripple voltage is highly dependent upon the external output capacitor but with care can be limited to between 20mV and 100mV peak to peak This ripple is at the switching frequency which can range from 20kHz to 1MHz There are also high frequency components in the output current of a switching regulator but these can be minimized with proper external filtering layout and grounding Efficiency can also vary widely with up to 95 sometimes being achievable 3 4 SWITCHING REGULATORS WHAT TO EXPECT FROM A SWITCHING R
6. 16 SPRAGUE 293D SURFACE MOUNT TANTALUM Cro 10NF 16V SURFACE MOUNT TANTALUM T491C SERIES Figure 8 30 ADP1148 9V TO 3 3V 1A BUCK REGULATOR The circuit for the ADP1148 9V to 3 3V 1A buck regulator is shown in Figure 8 31 and the input waveform in Figure 8 32 The input waveform is characteristic of the PWM buck regulator The decaying portion of the waveform occurs when the inductor is connected to the input The flat portion is when the input is disconnected from the inductor The fundamental switching frequency is approximately 150kHz The output waveform of the ADP1148 buck regulator is shown in Figure 8 33 Note that the output filter capacitors consist of two leaded OS CON types with very low ESR approximately 0 02Q each This results in a low ripple of 6mV peak to peak which is acceptable without further filtering 8 34 HARDWARE DESIGN TECHNIQUES ADP1148 BUCK REGULATOR APPLICATION Vin 9V O O 220uF 25V 10nF Vin IRF7204 V P DRIVE9 ADP1148 3 3 L 90 Your SHUTDOWN 20 olg SENSE C2 1kO 1000pF e Pc SENSE Cr ETS 220pF 220y 3300pF N CH 10V 10V N DRIVEo lt 2 470pF SGND PGND C1 220pF 25V GEN PURPOSE AL ELECTROLYTIC EFSOIPIBONIGS 1 CERAMIC C2 220uF 10V OSCON 2 Figure 8 31 ADP1148 BUCK INPUT WAVEFORM CONDITION 1 Vout 3 3V 1A 200mvV p p ADP1148 BUCK REG CIRCUIT C1 220 u
7. V d ESL IDEAL ACTUAL Figure 4 3 If an ideal capacitor is charged with an ideal voltage source as shown in Figure 4 4 A the capacitor charge buildup occurs instantaneously corresponding to a unit impulse of current A practical circuit Figure 4 4 B will have resistance in the switch Rgw as well as the equivalent series resistance ESR of the capacitor In addition the capacitor has an equivalent series inductance ESL The charging current path also has an effective series inductance which can be minimized with proper component layout techniques These parasitics serve to limit the peak current and also increase the charge transfer time as shown in the diagram Typical switch resistances can range from 10 to 500 and ESRs between 50mQ and 200m2 Typical capacitor values may range from about 0 1yF to 1O0pF and typical ESL values 1 to 5nH Although the equivalent RLC circuit of the capacitor can be underdamped or overdamped the relatively large switch resistance generally makes the final output voltage response overdamped 4 4 SWITCHED CAPACITOR VOLTAGE CONVERTERS CHARGING A CAPACITOR FROM A VOLTAGE SOURCE IDEAL A ACTUAL B Vout Figure 4 4 The law of conservation of charge states that if two capacitors are connected together the total charge on the parallel combination is equal to the sum of the original charges on the capacitors Figure 4 5 shows two capacitors C1 and C2 each charged to voltages
8. 100 95 EFFICIENCY 90 SCHOTTKY 85 80 0 01 0 03 0 1 0 3 1 3 OUTPUT CURRENT A Figure 3 41 SWITCHING REGULATORS ADP1147 STEP DOWN REGULATOR CONTROLLER KEY SPECIFICATIONS Input Voltage Range 3 5V to 14V 16V Max Output Voltage Options 3 3V 5V Current Mode Control Circuit Constant Off Time 5ys Variable Frequency P Channel MOSFET Gate Drive Output Power Saving Mode 160 Typical Up to 95 Efficiency 8 Pin SOIC and DIP Packages Figure 3 42 In order to achieve even higher efficiency the Schottky diode can be replaced with an N channel MOSFET switch as shown in Figure 3 43 This configuration is referred to as a synchronous rectifier or synchronous switch because the switching of the N channel MOSFET switch must be synchronized to the switching of the P channel MOSFET switch so that it essentially passes the current in one direction and blocks it in the other direction just like a rectifier or diode This terminology does not imply that the switching frequency of the regulator is synchronized to an external clock The gate drive signals from the controller must be non overlapping to prevent cross conduction current spikes in the switches This means that there is a period of time when both switches are off The external Schottky diode prevents the body diode of the N channel MOSFET from conducting during this time It is not always necessary to add the Schottky diode but it will increase overall efficie
9. 9CS 98 where is the junction to case thermal resistance the case to heat sink thermal resistance and the heat sink to ambient thermal resistance Each term is multiplied by the device power dissipation Pp to determine the temperature rise associated with each thermal resistance Ty Pp 9cg 88A 8 51 HARDWARE DESIGN TECHNIQUES In most situations the maximum junction temperature T MAX maximum ambient temperature TA M AX and Pp are known quantities and it is desired to calculate the required heat sink thermal resistance 08A which will limit the junction temperature to T j w AX under the specified conditions We know that the junction to ambient thermal resistance 7 can be expressed in terms of TJ MAX and Pr as follows TJ MAX TA MAX Pp 9JA We also know that be expressed in terms of 07A and TJ MAX TA MAX SA JA JC OCS Pp 0JC 9Cs In most cases can be less than 1 C W with the use of thermal grease and the expression for the maximum allowable heat sink to ambient resistance reduces to _ TA MAX _ Pp 9JC A design example will help clarify the process and identify the various tradeoffs Consider the low dropout linear regulator circuit shown in Figure 8 51 based on the ADP3310 LDO THERMAL DESIGN EXAMPLE GATE Vour ADP3310 3 3
10. Other difficulties in prototyping may occur with op amps or other linear devices having bandwidths greater than a few hundred megahertz Small variations in parasitic capacitance 1pF between the prototype and the final board may cause subtle differences in bandwidth and settling time Oftentimes prototyping is done with DIP packages when the final production package is an SOIC This can account for differences between prototype and final PC board performance However this option may not be available as many new ICs are only being introduced in surface mount packages EVALUATION BOARDS Walt Kester Most manufacturers of analog ICs provide evaluation boards usually at a nominal cost which allow customers to evaluate products without constructing their own prototypes Regardless of the product the manufacturer has taken proper precautions regarding grounding layout and decoupling to ensure optimum device performance The artwork or CAD file is usually made available free of charge 8 9 HARDWARE DESIGN TECHNIQUES should the customer wish to copy the layout directly or make modifications to suit the application Figure 8 7 shows the very compact evaluation board for the ADP3300 50mA low dropout linear regulator The ADP3300 is located in the center of the board and is in a SOT 23 6 lead package The input capacitor C1 and output capacitor C2 are both 0 47uF low inductance surface mount devices The Noise Reduction capacitor
11. BATTERY CHARGERS The value of the charge current is controlled by the feedback loop comprised of the external DC DC converter and the DC voltage at the VCTRL input The actual charge current is set by the voltage VcTRL and is dependent upon the choice for the values of Rcg and R3 according to ICHARGE hice dure Rcs 80kQ Typical values Rcg 0 250 and R3 20kQ which result in a charge current of 1 0A for a control voltage of 1 0V The 80kQ resistor is internal to the IC and it is trimmed to its absolute value The positive input of GM1 is referenced to ground forcing the point to a virtual ground The low side sense resistor Rog converts the charging current into a voltage which is applied to the pin If the charge current increases above its programmed value the GM1 stage forces the current IOUT to increase The higher IOUT decreases the duty cycle of the DC DC converter reducing the charging current and balancing the feedback loop As the battery approaches its final charge voltage the voltage control loop takes over The system becomes a voltage source floating the battery at constant voltage thereby preventing overcharging The voltage control loop is comprised of R1 R2 GM2 and the DC DC converter The final battery voltage is simply set by the ratio of R1 to R2 according to VBAT 2 000V 1 If the battery voltage rises above its programm
12. Johnson Howard W 8 13 8 88 Jung Walt 2 1 2 24 25 3 69 8 1 8 14 8 19 8 44 8 45 8 86 8 88 K Kelvin sensing 2 55 circuit 2 16 Kerridge Brian 5 25 Kester Walt 1 1 2 1 3 1 4 1 5 1 6 1 6 38 7 1 8 1 8 2 8 9 8 14 8 17 8 19 8 45 8 88 Kimmel Bill 8 88 INDEX Kovacs Gregory T A 2 57 KRL Bantry Components 2 57 L Laptop computers circuit redundancy 1 5 6 design challenges 1 4 off line flyback battery charger 5 17 replacing desktop systems 1 4 thermal and power management critical 1 5 Law of conservation of charge 4 5 Law of Intermediate Metals 6 6 LDO regulator ADP330X series diagram 2 43 advantages 2 25 board layout general guidelines 2 45 46 capacitor ESR zones 2 37 controller 2 48 56 basic design 2 49 50 circuit 2 56 current limit sense voltage errors 2 55 differences 2 48 49 multiple low ESR capacitors 2 55 printed circuit copper resistance design 2 54 DC and AC design issues 2 36 frequency compensation pole splitting fatal flaw 2 40 41 linear voltage power efficiency enhancement 2 27 pass device application problem 2 35 high output impedance 2 35 inverting mode 2 33 34 saturation capability 2 36 stability criteria 2 35 PCB layout issues 2 55 Kelvin sensing 2 55 routing techniques 8 14 15 sensing resistors 2 53 55 thermal considerations 2 45 47 traditional architecture 2 34 performance 2 35 Lee Seri 8 58 Lenk John D 3 69 Li Ala
13. TEMPERATURE SENSORS The TMP03 TMP04 are ideal for monitoring the thermal environment within electronic equipment For example the surface mounted package will accurately reflect the thermal conditions which affect nearby integrated circuits The TO 92 package on the other hand can be mounted above the surface of the board to measure the temperature of the air flowing over the board The TMP03 and TMP04 measure and convert the temperature at the surface of their own semiconductor chip When they are used to measure the temperature of a nearby heat source the thermal impedance between the heat source and the sensor must be considered Often a thermocouple or other temperature sensor is used to measure the temperature of the source while the TMP03 TMP04 temperature is monitored by measuring T1 and T2 Once the thermal impedance is determined the temperature of the heat source can be inferred from the TMP03 TMP04 output One example of using the TMP04 to monitor a high power dissipation microprocessor or other IC is shown in Figure 6 32 The TMP04 in a surface mount package is mounted directly beneath the microprocessor s pin grid array PGA package In a typical application the TMP04 s output would be connected to an ASIC where the mark space ratio would be measured The TMP04 pulse output provides a significant advantage in this application because it produces a linear temperature output while needing only one I O pin and without requi
14. perhaps a feedback controller if it just generates the feedback signal to the switch modulator It is important to know what you are getting in your controller and to know if your switching regulator is really a regulator or is it just the controller function Also like switchmode power conversion linear power conversion and charge pump technology offer both regulators and controllers So within the field of power conversion the terms regulator and controller can have wide meaning The most basic switcher topologies require only one transistor which is essentially used as a switch one diode one inductor a capacitor across the output and for practical but not fundamental reasons another one across the input A practical converter however requires several additional elements such as a voltage reference error amplifier comparator oscillator and switch driver and may also include optional features like current limiting and shutdown capability Depending on the power level modern IC switching regulators may integrate the entire converter except for the main magnetic element s usually a single inductor and the input output capacitors Often a diode the one which is an essential element of basic switcher topologies cannot be integrated either In any case the complete power conversion for a switcher cannot be as integrated as a linear regulator for example The requirement of a magnetic element means that system de
15. 2 1 2 Regulator circuit grounding techniques 8 14 18 Regulators low dropout linear 2 1 57 Resistance temperature detector 6 11 15 6 11 19 characteristics 6 12 configuration 6 13 four wire Kelvin connection 6 14 interfacing high resolution ADC 6 15 passive sensor 6 12 Seebeck coefficient 6 11 12 voltage drop 6 12 13 Resistance temperature devices 6 2 RFI analog circuits instrument disruption 8 65 68 minimizing 8 68 rectification 8 67 electric field strength 8 66 equipment sensitivity electric field intensity 8 66 immunity 8 66 frequency bands separation 8 69 instrument disruption 8 65 70 shielded cables 8 65 noise currents inductive capacitive coupling 8 66 67 Rich A 8 77 8 86 RTD see Resistance temperature detector S SAE EMI standards 8 61 Scaled reference voltage reference 2 19 20 Schottky diode external 3 43 loss 3 41 manufacturer listing 3 71 replaced by N channel MOSFET 3 43 44 Schweber Bill 5 25 7 15 Sealed lead acid battery 5 8 4 5 6 fast charging characteristics 5 6 slow charging characteristics 5 6 Seebeck coefficient resistance temperature detector 6 11 12 thermocouples 6 3 5 Type J thermocouple 6 5 Type K thermocouple 6 8 Type S thermocouple 6 11 12 Semiconductor temperature sensor 6 2 6 19 38 Index 10 basic relationships 6 20 current voltage output type 6 21 26 digital output type 6 26 29 on chip 6 36 38 in process control 1 7 theory 6
16. 4ESRcq 1 f C1 Typical values for switch resistances are between 1 20Q and ESRs between 50 and 200m2 The values of C1 and f are generally chosen such that the term 1 f C1 is less than 10 For instance 10 100kHz yields R 10 The dominant sources of power loss in most inverters are therefore the switch resistances and the ESRs of the pump capacitor and output capacitor The ADP3603 3604 3605 3607 series regulators have a shutdown control pin which can be asserted when load current is not required When activated the shutdown feature reduces quiescent current to a few tens of microamperes Power losses in a voltage doubler circuit are shown in Figure 4 13 and the analysis is similar to that of the inverter 4 12 SWITCHED CAPACITOR VOLTAGE CONVERTERS VOLTAGE DOUBLER POWER LOSSES PLoss louT 2ViN VOUT IqViN louT Rout aVin Rout 8Rsw 4 58 2 5 Figure 4 13 UNREGULATED INVERTER DOUBLER DESIGN EXAMPLE The ADM660 is a popular switched capacitor voltage inverter doubler IC see Figure 4 14 Switching frequency is selectable 25kHz 120kHz using the FC input When the FC input is open the switching frequency is 25kHz When it is connected to V the frequency increases to 120kHz Only two external electrolytic capacitors ESR should be less than 200m are required for operation see Figure 4 14 The choice of the value of these capacitors is somewhat flexible For a 25k
17. 6 2 3 common types 6 3 description 6 2 output voltage 6 4 5 principles 6 2 11 in process control 1 7 Seebeck coefficient vs temperature 6 4 temperature difference measurement 6 6 temperature sensors 6 2 11 Type J Seebeck coefficient 6 5 Type J K S 6 3 4 Type K output conditioning 6 8 9 Seebeck coefficient 6 8 Type S Seebeck coefficient 6 11 12 voltage generation 6 6 voltage temperature curves 6 3 4 Thermoelectric e m f 6 5 dissimilar metals 6 6 Thermoelectric voltage 6 5 01 programmable setpoint controller 6 31 32 circuit 6 31 control relay driver 6 31 key features 6 32 TMP03 TMP04 digital output sensors 6 26 29 circuit 6 27 output format 6 27 TMP03 TMP 04 electronic equipment thermal monitoring 6 29 TMP04 high power dissipation monitoring 6 29 microcontroller interfacing 6 28 TMP12 in airflow monitor 6 32 35 airflow temperature sensor IC 6 32 35 circuit diagram 6 34 35 parasitic temperature errors 6 35 programming for airspeed control 6 34 setpoint controller circuit 6 35 temperature errors 6 35 temperature relationships 6 33 TMP12 setpoint temperature programming 6 34 TMP17 current output sensors 6 21 22 TMP35 cold junction compensation 6 9 6 11 voltage output sensor 6 8 TMP35 TMP36 TMP37 voltage output sensors 6 23 263 thermal resistance 8 56 Tolerance voltage references 2 14 Transmitted power electric field strength 8 66 Trickle char
18. 8 20 22 Ambient temperature 8 45 Amplifier linearized thermistor 6 19 Amplifier Applications Guide Analog Devices 8 13 8 77 Analog circuit definition 8 1 prototyping 8 2 9 bird s nest 8 4 deadbug 8 3 Mini Mount 8 5 Solder Mount 8 5 advantages 8 5 components 8 5 RFI minimization 8 68 RFI sensitivity 8 66 simulation considerations 8 2 Antenna effect 8 85 Antenna gain electric field strength 8 66 Antognetti Paolo 8 13 anyCAP capacitor size 2 46 LDO regulator series 2 38 47 comparison 2 42 controller block diagram 2 49 diagram 2 43 Thermal Coastline packaging 2 47 thermal performance 2 46 LDO topology benefits 2 42 pole splitting topology 2 41 42 voltage regulation wide range 2 44 Avalanche diode breakdown 2 3 AVX TPS series capacitors 3 66 B Bandgap temperature sensor 6 21 Bandgap voltage reference 2 4 9 basic circuit 2 4 characteristics 2 13 shunt circuit 2 7 two terminal 2 5 Barrow Jeff 8 87 Battery boost regulator 3 7 capacity 5 2 charge discharge cycles 5 2 charger 5 1 25 linear 5 17 18 offline isolated flyback 5 14 17 switch mode dual 5 18 22 universal 5 22 24 Index 3 INDEX diagram 5 23 charging 5 5 13 fast 5 5 6 generalized circuit 5 5 termination methods 5 7 8 trickle 5 5 6 current 5 2 discharge profiles 5 4 discharge rate 5 2 disposal 5 4 environmental concerns 5 4 fundamentals 5 2 4 internal multiplexer final voltage sel
19. Long term drift in precision analog circuits is a random walk phenomenon and increases with the square root of the elapsed time this supposes that drift is due to random micro effects in the chip and not some over riding cause such as contamination The 1 year figure will therefore be about V8 766 3 times the 1000 hour figure and the ten year value will be roughly 9 times the 1000 hour value In practice things are a little better even than this as devices tend to stabilize with age 2 14 REFERENCES AND LOW DROPOUT LINEAR REGULATORS The accuracy of an ADC or DAC can be no better than that of its reference Reference temperature drift affects fullscale accuracy as shown in Figure 2 13 This table shows system resolution and the TC required to maintain 1 2 LSB error over an operating temperature range of 100 C For example a TC of about 1ppm C is required to maintain 1 2LSB error at 12 bits For smaller operating temperature ranges the drift requirement will be less The last three columns of the table show the voltage value of 1 2 LSB for popular full scale ranges REFERENCE TEMPERATURE DRIFT REQUIREMENTS FOR VARIOUS SYSTEM ACCURACIES 1 2 LSB CRITERIA 100 C SPAN 1 LSB WEIGHT mV 10 5 AND 2 5V FULLSCALE RANGES REQUIRED BITS DRIFT ppm C 10V 5V 2 5V 8 19 53 19 53 9 77 4 88 9 9 77 9 77 4 88 2 44 10 4 88 4 88 2 44 1 22 11 2 44 2 44 1 22 0 61 12 1 22 1 22 0 61 0 3
20. V cell 2 27 1 50 1 50 4 1 or 4 2 Time hr 24 16 16 16 Temp Range 0 45 5 45 5 40 5 40 Termination None None Timer Voltage Limit Figure 5 7 Fast charging batteries charge time less than 3 hours requires much more sophisticated techniques Figure 5 8 summarizes fast charging characteristics for the four popular battery types The most difficult part of the process is to correctly determine when to terminate the charging Undercharged batteries have reduced capacity while overcharging can damage the battery cause catastrophic outgassing of the electrolyte and even explode the battery BATTERY CHARACTERISTICS FOR FAST CHARGING lt 3HOURS 5 6 SLA NiCd NiMH Li lon Current gt 1 5C gt 1C gt 1C 1C Voltage V cell 2 45 1 50 1 50 4 1 or 4 2 50mV Time hours lt 1 5 lt 3 lt 3 2 5 Temp Range C 0 30 15 to 40 15 to 40 10 to 40 Primary Imin AV dT dt Imin Termination ATCO dT dt dV dt 0 Voltage Limit Secondary Timer TCO TCO TCO Termination ATCO Timer Timer Timer C Normal Capacity Minimum Current Threshold Termination TCO Absolute Temperature Cutoff Figure 5 8 ATCO Temperature Rise Above Ambient BATTERY CHARGERS Because of the importance of proper charge termination a primary and secondary method is generally used Depending on the battery type the charge may be terminated based on monitoring battery voltage voltage c
21. or TO 92 packaged voltage output temperature sensors with a 10mV C TMP35 36 or 20mV C TMP37 scale factor see Figure 6 26 Supply current is below providing very low self heating less than 0 1 C in still air A shutdown feature is provided which reduces the current to 0 5pA The TMP35 provides a 250mV output at 25 C and reads temperature from 10 C to 125 C The TMP36 is specified from 40 C to 125 C and provides a 750mV output at 25 C Both the TMP35 and TMP36 have an output scale factor of 10mV C The TMP37 is intended for applications over the range 5 C to 100 C and provides an output scale factor of 20mV C The TMP37 provides a 500mV output at 25 C ABSOLUTE VOLTAGE OUTPUT SENSORS WITH SHUTDOWN Vg 2 7V TO 5 5V SHUTDOWN ALSO SO 8 OR TO 92 Vour SOT 23 5 B Vout TMP35 250mV 25 C 10mV C 10 C to 125 C TMP36 750mV 25 C 10mV C 40 to 125 C TMP37 500mV 25 C 20mV C 5 C to 100 C B 2 C Error Over Temp Typical 0 5 C Non Linearity Typical Specified 40 C to 125 C 50pA Quiescent Current 0 5pA in Shutdown Mode Figure 6 26 6 23 TEMPERATURE SENSORS The ADT45 ADT50 are voltage output temperature sensors packaged in a SOT 23 3 package designed for an operating voltage of 2 7V to 12V see Figure 6 27 The devices are specified over the range of 40 C to 125 C The output scale factor for both devic
22. or LO is defined as regulation active or vice versa The error output ERR is useful within a system to detect regulator overload such as saturation of the pass device thermal overload etc The remaining functions shown are always part of an IC power regulator 2 27 REFERENCES AND LOW DROPOUT LINEAR REGULATORS BLOCK DIAGRAM OF A VOLTAGE REGULATOR m ViN VpnoPour Vmi 7 Vout CURRENT PASS LIMIT DEVICE OVERLOAD SATURATION SENSOR R1 VOUT vner a OVERTEMP SENSOR R2 SHUTDOWN CONTROL COMMON Figure 2 24 In operation a voltage reference block produces a stable voltage VREF which is almost always a bandgap based voltage typically 1 2V which allows output voltages of 3V or more from supplies as low as 5V This voltage is presented to one input of an error amplifier with the other input connected to the VOUT sensing divider R1 R2 The error amplifier drives the pass device which in turn controls the output The resulting regulated voltage is then simply R1 VOUT VREF 1 x With a typical bandgap reference voltage of 1 2V the R1 R2 ratio will be approximately 3 1 for a 5V output When standby power is critical several design steps will be taken The resistor values of the divider will be high the error amplifier and pass device driver will be low power and the reference current will also be low By these means the regulator s unloaded standby current can
23. references etc Reference 2 and 3 These models represent approximations to the actual circuit and parasitic effects such as package capacitance and inductance and PC board layout are rarely included The models are designed to work with various versions of SPICE simulation programs such as PSpice Reference 4 and run on workstations or personal computers The models are simple enough so that circuits using multiple ICs can be simulated in a reasonable amount of computation time and with good certainty of convergence Consequently SPICE modeling does not always reproduce the exact performance of a circuit and should always be verified experimentally using a carefully built prototype Finally there are mixed signal ICs such as A D and D A converters which have no SPICE models or if they exist the models do not simulate dynamic performance Signal to noise effective bits etc and prototypes of circuits using them should always be built In addition to mixed signal ICs switching regulators do not lend themselves to SPICE modelling The dynamics of either magnetic based or switched capacitor based regulators are far too complex for simple macromodels ANALOG CIRCUIT SIMULATION CONSIDERATIONS B ADSpice Macromodels Over 500 Exist for the Following Amplifiers Analog Multipliers Multiplexers and Switches Voltage References No Practical SPICE Macromodels for ADCs DACs Switching Regulators There is No Substitute fo
24. usually 0 C as shown in Figure 6 8 Ideally the compensation voltage should be an exact match for the difference voltage required which is why the diagram gives the voltage as f T2 a function of T2 rather than KT2 where K is a simple constant In practice since the cold junction is rarely more than a few tens of degrees from 0 C and generally varies by little more than 10 C a linear approximation V KT2 to the more complex reality is sufficiently accurate and is what is often used The expression for the output voltage of a thermocouple with its measuring junction at T C and its reference at 0 C is a polynomial of the form V K9T2 but the values of the coefficients Ko Kg etc are very small for most common types of thermocouple References 8 and 9 give the values of these coefficients for a wide range of thermocouples 6 6 TEMPERATURE SENSORS CLASSICAL COLD JUNCTION COMPENSATION USING AN ICE POINT 0 C REFERENCE JUNCTION METAL A METAL A V1 W 0 C T1 METAL B ICE BATH Figure 6 7 USING A TEMPERATURE SENSOR FOR COLD JUNCTION COMPENSATION TEMPERATURE COMPENSATION CIRCUIT o RB oci METAL A TEMP SENSOR T1 METAL B ISOTHERMAL BLOCK V COMP f T2 V OUT zV T1 V T2 V COMP IF V COMP V T2 V 0 C THEN V OUT V T1 V 0 C Figure 6 8 6 7 TEMPERATURE SENSORS When electronic cold junction compensation is used
25. y typ max mA 25 C All SO 8 are Singles have 496 Full Thermal NR SD ERR coastline Dual no NR ADP3300 0 08 0 17 50 0 8 1 4 SOT 23 6 Single ADP3301 0 10 0 2 100 0 8 1 4 SO 8 Single ADP3302 0 10 0 2 100 0 8 1 4 SO 8 Dual ADP3303 0 18 0 4 200 0 8 1 4 SO 8 Single ADP3307 0 13 0 22 100 0 8 1 4 SOT 23 6 Single Figure 2 35 REFERENCES AND LOW DROPOUT LINEAR REGULATORS Functional Diagram and Basic 50 mA LDO Regulator A functional diagram common to the various devices of the ADP330X series LDO regulators is shown by Figure 2 36 Operation of the various pins and internal functions is discussed below anyCAP SERIES LDO REGULATORS FUNCTIONAL DIAGRAM IN Qi O OUT THERMAL PROTECTION Q2 DRIVER R2 BANDGAP REF O GND Figure 2 36 In application the use of the anyCAP series of LDOs is simple as shown by a basic 50mA ADP3300 regulator in Figure 2 37 This circuit is a general one to illustrate points common to the entire device series The ADP3300 is a basic LDO regulator device designed for fixed output voltage applications while operating from sources over a range of 3 to 12V and a temperature range of 40 to 85 C The actual ADP3300 device ordered would be specified as ADP3300ART YY where the YY is a voltage designator suffix such as 2 7 3 3 2 3 3 or 5 for those respective voltages The ART portion of the part number desi
26. 1 7 REFERENCES THERMAL MANAGEMENT 8 58 Power Consideration Discussions AD815 Data Sheet Analog Devices Heat Sinks for Multiwatt Packages AAVID Thermal Technologies Inc One Kool Path Laconia NH 03246 603 528 3400 General Catalog AAVID Thermal Technologies Inc One Kool Path Laconia NH 03246 603 528 3400 Seri Lee How to Select a Heat Sink AAVID Thermal Technologies Available on internet site http www aavid com Seri Lee Optimum Design and Selection of Heat Sinks 11 IEEE SEMI THERM Symposium 1995 Also available on internet site http www aavid com Alan Li et all Maximum Power Enhancement Techniques for SO 8 Power MOSFETs Fairchild Semiconductor application note AN1029 April 1996 http www fairchildsemi com Rob Blattner Wharton McDaniel Thermal Management in On Board DC to DC Power Conversion Temic application note http www temic com HARDWARE DESIGN TECHNIQUES EMI RFI CONSIDERATIONS Electromagnetic interference EMI has become a hot topic in the last few years among circuit designers and systems engineers Although the subject matter and prior art have been in existence for over the last 50 years or so the advent of portable and high frequency industrial and consumer electronics has provided a comfortable standard of living for many EMI testing engineers consultants and publishers With the help of EDN Magazine and Kimmel Gerke Associates this section will highli
27. 10 ADM9261 key features 7 4 pager power system application circuit 7 6 specifications 7 5 triple comparator and reference 7 5 triple power supply monitor IC 7 4 ADM9264 application circuit 7 8 block diagram 7 7 error output signals 7 6 quad power supply monitor IC 7 6 specifications 7 7 ADM9268 hex voltage monitor 7 8 monitor voltage in Pentium II processor 7 8 similar to ADM9264 7 8 specifications 7 8 ADP1147 buck converter controller 3 41 44 efficiency losses 3 41 42 specifications 3 43 step down application 3 41 42 buck pulse wave modulation regulator 3 53 sleep power saving mode 3 31 switch modulator 3 26 ADP1148 buck pulse wave modulation regulator 3 53 evaluation board 8 10 12 switching regulator 8 10 11 8 16 buck application circuit 8 34 35 driving ADP3310 linear low dropout post regulator 8 37 38 waveforms 8 38 filtered output 8 37 input output waveforms 8 35 36 synchronous step down regulator controller 3 44 46 application circuit 3 44 45 efficiency losses 3 45 specifications 3 46 ADP3000 low dropout linear regulator 8 9 switching regulator 3 28 8 6 8 block diagram 3 36 boost application circuit 3 38 8 28 experiment 8 27 31 filtered output 8 30 31 waveforms 8 29 buck application circuit 3 38 8 32 Index 2 experiment 8 32 34 filtered output 8 34 input output waveforms 8 33 NBN switching 3 34 pulse burst modulation 3 32 3 34 38 pulse burst modulation
28. 1000 2000uA channel with the OP284 and OP279 The former series is most useful for very light loads lt 2mA while the latter series provide device dependent outputs up to 50mA Various devices can be used in the circuit as shown and their key specs are summarized in Figure 2 18 2 19 REFERENCES AND LOW DROPOUT LINEAR REGULATORS OP AMPS USEFUL IN LOW VOLTAGE RAIL RAIL REFERENCES AND REGULATORS DEVICE 14 mA Vsat Vsat Isc mA per channel V 2 V max mA min OP181 281 481 0 003 4 93 9 0 05 0 075 0 05 3 5 OP193 293 493 0 017 4 20 9 1 0 280 1 typ 8 OP196 296 496 0 045 4 30 9 1 0 400 1 4 typ OP295 495 0 150 4 50 1 0 110 1 11 OP191 291 491 0 300 4 80 2 5 0 075 2 5 8 75 AD820 822 0 620 4 89 2 0 055 2 15 OP184 284 484 1 250 4 85 2 5 0 125 2 5 7 5 AD8531 32 34 1 400 4 90 10 0 100 10 250 OP279 2 000 4 80 10 0 075 10 45 Typical device specifications Vs 5V 25 C unless otherwise noted Maximum Figure 2 18 In Figure 2 17 without gain scaling resistors R2 R3 VoyT is simply equal to VREF With the use of the scaling resistors can be set anywhere between a lower limit of VREF and an upper limit of the positive rail due to the op amp s rail rail output swing Also note that this buffered reference is inherently low dropout allowing a 4 5V or more reference outp
29. 10uF 16V tantalum capacitor Kemet T491C series is also shown outside the box as it might be used on a more conventional LDO circuit It is several times the size of output capacitor C2 Recent developments in packaging have led to much improved thermal performance for power management ICs The anyCAPTM LDO regulator family capitalizes on this most effectively using a thermally improved leadframe as the basis for all 8 pin devices This package is called a Thermal Coastline design and is shown in Figure 2 39 The foundation of the improvement in heat transfer is related to two key parameters of the leadframe design distance and width The payoff comes in the reduced thermal resistance of the leadframe based on the Thermal Coastline only 90 C W versus 160 C W for a standard SO 8 package The increased dissipation of the Thermal Coastline allows the anyCAP series of SO 8 regulators to support more than one watt of dissipation at 25 C 2 46 REFERENCES AND LOW DROPOUT LINEAR REGULATORS anyCAP SERIES REGULATORS IN SO 8 USE THERMAL COASTLINE PACKAGES 1 8 1 8 2 7 2 7 3 6 3 6 STANDARD LEADFRAME SOIC THERMAL COASTLINE SOIC 0JA 160 C W 0JA 90 C W Figure 2 39 Additional insight into how the new leadframe increases heat transfer can be appreciated by Figure 2 40 In this figure it can be noted how the spacing of the Thermal Coastline paddle and leads shown on the right is reduced while the width of the lead ends
30. 20 22 tantalum electrolytic 3 63 8 20 22 considerations 3 59 66 continuous switching 4 6 steady state 4 7 electrolytic impedance versus frequency 8 23 24 ripple current rating 3 64 65 equivalent circuit pulse response 8 23 fundamentals 3 8 10 and law of conservation of charge 4 5 manufacturer listing 3 70 parasitics 4 3 4 8 23 pump diagram 4 7 stored charge diagram 4 4 theoretical 4 3 4 types 8 20 22 Cell phones low dropout linear regulators 1 5 6 shutdown features 1 5 Ceramic capacitor multilayer 3 63 64 8 20 22 Charge control 3 31 Charge transfer capacitors 4 3 7 Chestnut Bill 8 1 8 19 Chryssis George 3 69 Circuit high precision 8 3 high speed 8 3 mixed signal 8 3 reference bandgap based 2 4 series 2 4 shunt 2 4 zener diode based 2 4 see also Voltage reference INDEX Clelland Ian 3 69 8 44 Cold junction 6 6 compensation electronic 6 8 ice point reference 6 7 temperature sensor 6 7 Contact potential 6 5 Core manufacturer listing 3 70 Current output sensor 6 21 22 Current mode control 3 29 30 disadvantages 3 30 31 D DAC high speed EMI RFI noise 8 73 Dan Pnina 5 25 Data acquisition board block diagram 1 3 4 Deadbug prototyping 8 3 4 Decoupling 1 7 2 14 Designing for EMC Workshop Notes 8 77 8 86 Device junction temperature reference point 8 45 Digital output sensor nominal output frequency 6 26 sigma delta ADC modulator 6 26 29 Diode re
31. 2ppm C AD588 and AD586 and the lowest noise as a percent of full scale i e 100nV VHz or less On the downside the operating current of zener type references is usually relatively high typically on the order of several mA An important general point arises when comparing noise performance of different references The best way to do this is to compare the ratio of the noise within a given bandwidth to the DC output voltage For example a 10V reference with a 100nV VHz noise density is 6dB more quiet in relative terms than is a 5V reference with the same noise level XFET REFERENCES A third and brand new category of IC reference core design is based on the properties of junction field effect JFET transistors Somewhat analogous to the bandgap reference for bipolar transistors the JFET based reference operates a pair of junction field effect transistors with different pinchoff voltages and amplifies the differential output to produce a stable reference voltage One of the two JFETs uses an extra ion implantation giving rise to the name XFET eXtra implantation junction Field Effect Transistor for the reference core design The basic topology for the XFET reference circuit is shown in Figure 2 9 J1 and J2 are the two JFET transistors which form the core of the reference J1 and J2 are driven at the same current level from matched current sources I1 and I2 To the right J1 is the JFET with the extra implantation which ca
32. 755254 52 R1 kT V 2 1 8 BE Q1 R2 q 1 205V Note that J1 current density in Q1 J2 current density in Q2 and J1 J2 8 However because of the presence of the R4 R5 laser trimmed thin film divider and the op amp the actual voltage appearing at can be scaled higher in the AD580 case 2 5V Following this general principle Vout be raised to other practical levels such as for example in the AD584 with taps for precise 2 5 5 7 5 and 10V operation The AD580 provides up to 10mA output current while operating from supplies between 4 5 and 30V It is available in tolerances as low as 10mV with TCs as low as 10ppm C 2 6 REFERENCES AND LOW DROPOUT LINEAR REGULATORS Many of the recent developments in bandgap references have focused on smaller package size and cost reduction to address system needs for smaller more power efficient and less costly reference ICs Among these are several recent bandgap based IC references The AD1580 is a shunt mode IC reference which is functionally quite similar to the classic shunt IC reference the AD589 mentioned above A key difference is the fact that the AD1580 uses a newer small geometry process enabling its availability within the tiny SOT 23 package The very small size of this package allows use in a wide variety of space limited applications and the low operating current lends itself to portable battery powered uses The AD1580 circuit is
33. 8 87 European Community EMC standards 8 60 European Community standards see EMC Evaluation board 8 9 13 uses 8 12 Extra implantation junction field effect transistor 2 10 F FDA EMI standards 8 60 Federal Communications Commission see FCC Ferrites characteristics 8 23 24 impedance 8 24 in inductor power supply filters 8 23 8 73 materials bead 8 24 leaded ferrite bead 8 24 PSpice models 8 24 Ferromagnetic core 3 56 Film capacitor 3 63 64 Filter broadband 8 69 70 Index 6 failure 8 69 70 high pass 8 69 low pass 8 67 effectiveness 8 68 leakage 8 67 multistage 8 69 non zero ground effectiveness 8 70 Filtering reference noise performance 2 18 Flyback charger charge current vs voltage scheme 5 16 Flyback converter circuit 3 25 Forward converter circuit 3 26 Forward biased diode 2 3 G Gated oscillator control 3 31 34 Gauss 3 55 Gelbach Herman 8 86 Gerber files 8 12 Gerke Daryl 8 88 German EMI regulations by Verband Deutscher Electrotechniker VDE 8 59 Goodenough Frank 2 57 5 25 Gottlieb Irving M 3 69 Graham Martin 8 13 8 88 Grant Doug 8 88 Grounding 1 7 H Hageman Steve 8 44 Handbook of Chemistry and Physics 6 38 Hardware design techniques 1 8 8 1 85 analog circuit simulation 8 1 2 layout 1 7 Hardware monitoring 7 1 15 brownout 1 1 design techniques 1 7 microprocessor supply voltage 1 1 overview 1 2 parameters 1 1 7 1 proce
34. 85V to 5 75V 1 2mA Typical Supply Current 10A in Shutdown 24 pin SOIC Package Figure 7 13 Figure 7 14 shows a generic application circuit using the AD9240 The analog inputs are connected to the power supplies and processor core voltage VID inputs are connected to the processor Voltage ID pins There are two inputs from fans and the analog output is controlling the speed of a third fan A chassis intrusion latch with a phototransistor as the sensor is connected to the CI input In an actual application every input and output may not be used in which case unused analog and digital inputs should be tied to analog or digital ground as appropriate The chassis intrusion circuit could use a microswitch that opens or closes when the cover is removed a reed switch operated by a magnet fixed to the cover a Hall effect switch operated by a magnet fixed to the cover or a phototransistor that detects light when the cover is removed In the circuit shown in Figure 7 14 light falling on the phototransistor when the PC cover is removed will cause it to turn on and pull up the input of N1 thus setting the latch N2 N3 After the cover is replaced a low reset on the ADM9240 CI output will pull down the input of N3 thus resetting the latch 7 10 HARDWARE MONITORING ADM9240 APPLICATION CIRCUIT 12V 9 5V o NTESTOUT OS O A0 Q 12 9 45Vo ibat V 55 52 50 SDA VID PINS Dd SERIAL BUS VID2 P7 OF
35. 8A LDO regulator controller can be configured as shown in Figure 2 45 This circuit uses an ADP3310 2 8 to produce a 2 8V output The sense resistor is dropped to 5 milliohms which supports currents of up to 10A or about 6 7A with current limiting active Four terminal wiring should be used with the sense resistor to minimize errors The most significant change over the more generic schematic of Fig 2 43 is the use of multiple low ESR input and output bypass capacitors At the output C2 is a bank of 4 x 220uF OS CON type capacitors in parallel with 2 x 10uF MLCC chip type capacitors These are located right at the load point with minimum inductance wiring plus separate wiring back to the VouyT pin of the ADP3310 and the drain of the pass device This wiring will maximize the DC output accuracy while the multiple capacitors will minimize the transient errors at the point of load In addition multiple bypasses on the regulator input in the form of C1 minimizes the transient errors at the regulator s VIN pin 2 55 REFERENCES AND LOW DROPOUT LINEAR REGULATORS A 2 8V 8A LDO REGULATOR CONTROLLER Rs NDP6020P OR NDB6020P 2 8V 8A Vin on FAIRCHILD C2 4 x 220yF 2 x 220 OS CON OS CON Ag ATE IN Vout 2x10pF MLC 2x10pF MLC ADP3310 2 8 Figure 2 45 Heat sink requirements for the pass device in this application will be governed by the loading and input voltage and should be calculated by the procedur
36. Build an Ultra Low Noise Voltage Reference Electronic Design Analog Applications Issue June 24 1993 Walt Jung Getting the Most from IC Voltage References Analog Dialogue 28 1 1994 pp 13 21 REFERENCES AND LOW DROPOUT LINEAR REGULATORS LOW DROPOUT REGULATORS Walt Jung INTRODUCTION Linear IC voltage regulators have long been standard power system building blocks After an initial introduction in 5 V logic voltage regulator form they have since expanded into other standard voltage levels spanning from 3 to 24 V handling output currents from as low as 100 mA or less to as high as 5 A or more For several good reasons linear style IC voltage regulators have been valuable system components since the early days One reason is the relatively low noise characteristic vis a vis the switching type of regulator Others are a low parts count and overall simplicity compared to discrete solutions But because of their power losses these linear regulators have also been known for being relatively inefficient Early generation devices of which many are still available required 2V or more of unregulated input above the regulated output voltage making them lossy in power terms More recently however linear IC regulators have been developed with more liberal i e lower limits on minimum input output voltage This voltage known more commonly as dropout voltage has led to what is termed the Low DropOut regulator or more popularly
37. C2 charges linearly by an amount equal to IQUT 2f C2 When C1 is connected back to the input the ripple waveform reverses direction as shown in the diagram The total peak to peak output ripple voltage is therefore IOUT 2f C2 VRIPPLE 2100 ESRC2 4 9 SWITCHED CAPACITOR VOLTAGE CONVERTERS VOLTAGE INVERTER WAVEFORMS Vout loUT OUT VRIPPLE 2louT ESRc2 JOUT 2f C2 C VRIPPLE Figure 4 10 VOLTAGE DOUBLER WAVEFORMS gt Vout 2Vin VRIPPLE 2louT ESRc2 JOUT 2f C2 C VRIPPLE Figure 4 11 4 10 SWITCHED CAPACITOR VOLTAGE CONVERTERS The current and voltage waveforms for a simple voltage doubler are shown in Figure 4 11 and are similar to those of the inverter Typical voltage ripple for practical switched capacitor voltage inverter doublers range from 25mV to 100mV but can be reduced by filtering techniques as described in Section 8 of this book Note that the input current waveform has an average value of 21g jT because VIN is connected to C1 during C1 s charge cycle and to the load during C1 s discharge cycle The expression for the ripple voltage is identical to that of the voltage inverter SWITCHED CAPACITOR VOLTAGE CONVERTER POWER LOSSES The various sources of power loss in a switched capacitor voltage inverter are shown in Figure 4 12 In addition to the inherent switched capacitor resistance R 1 f C1 there are
38. C3 is 10nF Resistor R1 is a pullup resistor for the open collector error output pin The entire active circuit located within the small square is approximately 0 6 by 0 6 15 2mm by 15 2mm EVALUATION BOARD FOR ADP3300 LOW DROPOUT REGULATOR ANALOG DEVICES ADP3300 SANTA CLARA DIVISION Figure 8 7 Switching regulators such as the ADP1148 place more exacting demands on layout and decoupling The evaluation board for the ADP1148 is shown in Figures 8 8 top view and 8 9 bottom view ground plane The board size is approximately 2 by 2 5 1cm by 5 1cm The input decoupling capacitors C1 and C1A are each 220nF 25V general purpose aluminum electrolytic capacitors Also there is tantalum in parallel with C1 and C1A labeled C2 The output capacitors C6 and C6A are each low ESR OS CON 220nF 10V The ADP1148 is located to the right of the center of the board and is in a 14 pin SOIC surface mount package The energy transfer inductor L1 is a 68uH surface mount part from Coiltronics Other readily visible components making up the regulator are two power MOSFETs Q1 and Q2 and a 0 050 current sense resistor R2 8 10 HARDWARE DESIGN TECHNIQUES EVALUATION BOARD FOR ADP1148 SWITCHING REGULATOR TOP VIEW ANALOG 1148 2 6 95 SANTA CLARA DIVISION Figure 8 8 EVALUATION BOARD FOR ADP1148 SWITCHING REGULATOR BOTTOM VIEW GROUND PLANE 2 e g m o m Figure 8 9 8
39. Capacitor if Required B Efficiency gt 90 Achievable Optimized for Doubling or Inverting Supply Voltage Efficiency Degrades for Other Output Voltages Low Cost Compact Low Profile Height Parts with Voltage Regulation are Available ADP3603 ADP3604 ADP3605 ADP3607 Figure 4 2 The voltage inverter is useful where a relatively low current negative voltage is required in addition to the primary positive voltage This may occur in a single supply system where only a few high performance parts require the negative voltage Similarly voltage doublers are useful in low current applications where a voltage greater than the primary supply voltage is required CHARGE TRANSFER USING CAPACITORS A fundamental understanding of capacitors theoretical and real is required in order to master the subtleties of switched capacitor voltage converters Figure 4 3 shows the theoretical capacitor and its real world counterpart If the capacitor is charged to a voltage V then the total charge stored in the capacitor q is given by q CV Real capacitors have equivalent series resistance ESR and inductance ESL as shown in the diagram but these parasitics do not affect the ability of the capacitor to store charge They can however have a large effect on the overall efficiency of the switched capacitor voltage converter 4 3 SWITCHED CAPACITOR VOLTAGE CONVERTERS STORED CHARGE IN A CAPACITOR Q STORED CHARGE ESR 4
40. External Components Required 16 pin Narrow SOIC Package 150mil wide Figure 7 9 7 7 HARDWARE MONITORING ADM9264 APPLICATION CIRCUIT Vcc y o ADAPTER CHIP ERRX ADM9264 ADT05 TEMP ERRY SENSOR DIS Figure 7 10 The ADM9268 block diagram not shown is similar to the ADM9264 but monitors six power supplies in a desktop PC and outputs the status information on an industry standard two wire I2C compatible serial interface One input of the ADM9268 is designed to monitor the CPU core voltage of a Pentium II processor The range of CPU voltage options is from 1 3V to 3 5V and is set by a 5 bit VID code which is inputted via the serial I2C compatible interface This makes the ADM9268 compatible with all the CPUs currently available in the marketplace Key specifications for the ADM9268 are summarized in Figure 7 11 ADM9268 HEX VOLTAGE MONITOR Monitors All Six Desktop PC Power Supplies Simultaneously with Hex Window Comparators Monitors 12V 696 5V 7 3 3V X796 2 5V or 3 3V 7 1 5V 796 and CPU Core Voltage 596 5 bit VID Code Sets Core Monitor Voltage 1 3V to 3 5V Standard two wire I2C Compatible Serial Interface Operates on from 2 5V to 6V 16 pin Narrow 150mil SOIC Package Figure 7 11 7 8 HARDWARE MONITORING The ADM9240 see Figure 7 12 is a complete high level system hardware monitor for microprocessor based systems providing m
41. FCC and VDE standards are less stringent with respect to conducted interference by a factor of 10 over radiated levels The FCC Part 15 and VDE 0871 regulations group commercial equipment into two classes Class A for all products intended for business environments and Class B for all products used in residential applications For example Figure 8 58 illustrates the electric field emission limits of commercial computer equipment for both FCC Part 15 and VDE 0871 compliance 8 59 HARDWARE DESIGN TECHNIQUES RADIATED EMISSION LIMITS FOR COMMERCIAL COMPUTER EQUIPMENT Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA Frequency MHz Class A at 3m Class B at 3m 30 88 300 uV m 100 pV m 88 216 500 uV m 150 V m 216 1000 700 uV m 200 uV m Figure 8 58 In addition to the already stringent VDE emission limits the European Community EMC standards IEC and IEEE now requires mandatory compliance to these additional EMI threats Immunity to RF fields electrostatic discharge and power line disturbances All equipment systems marketed in Europe must exhibit an immunity to RF field strengths of 1 10V m IEC standard 801 3 electrostatic discharge generated by human contact or through material movement in the range of 10 15kV IEC standard 801 2 and power line disturbances of 4kV EFTs extremely fast transients IEC standard 801 4 and 6kV lightning surges IEE
42. GND O FET NDP6020P OR NDB6020P FAIRCHILD jc 2 3 C W Pp 5V 3 3V 3A 5 1W 125 5 TA MAX 50 C T E OSA MA BMAD c 125 89 3 14 7 3 11 7 C W Figure 8 51 8 52 HARDWARE DESIGN TECHNIQUES The power dissipated in the FET pass transistor Fairchild NDP6020P or NDB6020P due to the 1 7V drain to source voltage drop and the 3A output current is 5 1W Now assume that we want to hold the maximum transistor junction temperature to T MAX 125 C at an ambient temperature of 50 C The junction to case thermal resistance of the FET is specified by the manufacturer to be 3 C W We can now calculate the maximum allowable heat sink case to ambient thermal resistance neglecting the case to heat sink thermal resistance TJ MAX TA MAX _ Pp 125 50 147 3 117 C W 0SA lt The 6020P Fairchild FET is available in two packages as shown Figure 8 52 The TO 220 style has a junction to ambient thermal resistance of 53 C W no airflow and has a metal tab which is designed to be bolted to a heat sink The TO 263 style has a junction to ambient thermal resistance of 73 C W no airflow and is designed for surface mounting The metal drain tab of the surface mount package is designed to be soldered directly to the PC board pad which acts as a heat sink TO 220 AND TO 263 D PAK PACKAGES FOR FAIRCHILD NDP6020P NDB6020P FETs TO 263 NDB SERI
43. However it is important for the design engineer to understand at least some of the fundamental issues relating to inductors This discussion while by no means complete will give some insight into the relevant magnetics issues Selecting the actual value for the inductor in a switching regulator is a function of many parameters Fortunately in a given application the exact value is generally not all that critical and equations supplied on the data sheets allow the designer to calculate a minimum and maximum acceptable value That s the easy part Unfortunately there is more to a simple inductor than its inductance Figure 3 49 shows an equivalent circuit of a real inductor and also some of the many considerations that go into the selection process To further complicate the issue most of these parameters interact thereby making the design of an inductor truly more of an art than a science INDUCTOR CONSIDERATIONS E Inductance Value L ACTUAL MODEL Saturation Current IDEAL APPROXIMATE Inductor Losses Hysteresis Loss Eddy Current Loss Winding Loss R E Heating C B EMI RFI GORE Self Resonant Frequency Core Material Form Factor Core Volume Number of Turns Wire Size Spacing Temperature Operating Current Operating Frequency O HEHEHEHEHE E Figure 3 49 3 48 SWITCHING REGULATORS Probably the easiest inductor problem to solve is selecting the proper value In most switching regulator appli
44. LPKF CAD CAM Systems Inc 1800 NW 169th Place Beaverton OR 97006 and T Tech Inc 5591 B New Peachtree Road Atlanta GA 34341 11 Howard W Johnson and Martin Graham High Speed Digital Design PTR Prentice Hall 1993 12 Practical Analog Design Techniques Analog Devices 1995 13 High Speed Design Techniques Analog Devices 1996 8 13 HARDWARE DESIGN TECHNIQUES GROUNDING TECHNIQUES FOR REGULATOR CIRCUITS Walt Kester Walt Jung The importance of maintaining a low impedance large area ground plane is critical to practically all analog circuits today especially high current low dropout linear regulators or switching regulators The ground plane not only acts as a low impedance return path for high frequency switching currents but also minimizes EMI RFI emissions In addition it serves to minimize unwanted voltage drops due to high load currents Because of the shielding action of the ground plane the circuit s susceptibility to external EMI RFI is also reduced When using multilayer PC boards it is wise to add a power plane In this way low impedances can be maintained on both critical layers Figure 8 10 shows a grounding arrangement for a low dropout linear regulator such as the ADP3310 It is important to minimize the total voltage drop between the input voltage and the load as this drop will subtract from the voltage dropped across the pass transistor and reduce its headroom For this reason these runs should
45. Precision 2 5V 8mV Reference 400p A Quiescent Current 1A in Shutdown Packages 8 Pin Dip 8 Pin SOIC 8 Pin TO 99 Other Setpoint Controllers Dual Setpoint Controllers ADT20 21 22 Versions of 01 with Internal Hysteresis Quad Setpoint Controller ADT14 Figure 6 35 The ADT20 21 22 series are similar to the TMPO01 but have internal hysteresis and are designed to operate on a 3V supply A quad ADT14 setpoint controller is also available An Airflow Monitor Based on the TMP12 For large power dissipation and or to maintain low T s forced air movement be used to increase air flow and aid in heat removal In its most simple form this can consist of a continuously or thermostatically operated fan directed across high temperature high wattage dissipation devices such as CPUs DSP chips etc Quite often however more sophisticated temperature control is necessary Recent temperature monitoring and control ICs such as the TMP12 an airflow temperature sensor IC lend themselves to such applications The TMP12 includes on chip two comparators a voltage reference a temperature sensor and a heater The heater is used to force a predictable internal temperature rise to match a power IC such as a microprocessor The temperature sensing and control portions of the IC can then be programmed to respond to the temperature changes and control an external fan so as to maintain some range of temperature Compared to a sim
46. SHUTDOWN 5V O SENSE ep Ro 1kQ c Cg P SENSE o 3300pF a PD 390pF N DRIVEO 10BQ040 470pF SGND PGND Rgense KRL SL 1 C1 ORO50L V L COILTRONICS CTX 68 4 Figure 3 44 ADP1148 TYPICAL EFFICIENCY LOSSES 100 95 EFFICIENCY 96 90 85 80 0 01 0 03 0 1 0 3 1 3 OUTPUT CURRENT A Figure 3 45 3 45 SWITCHING REGULATORS ADP1148 HIGH EFFICIENCY SYNCHRONOUS SWITCH REGULATOR CONTROLLER KEY SPECIFICATIONS Input Voltage Range 3 5V to 18V 20V Max Output Voltage Options 3 3V 5V and Adjustable Current Mode Control Circuit Non Overlapping P and N Channel MOSFET Gate Drive Outputs Constant Off Time 5ys Variable Frequency Power Saving Mode 160 Typical Up to 95 Efficiency Possible 14 Pin SOIC and DIP Packages Figure 3 46 The ADP3153 is a 5 bit programmable synchronous switching regulator controller suitable for the Pentium II processor An application circuit is shown in Figure 3 47 and key specifications are given in Figure 3 48 The ADP3153 is optimized for applications where 5V is stepped down to a digitally controlled output voltage between 1 8V and 3 5V Using a 5 bit DAC to read a voltage identification VID code directly from the processor the ADP3153 generates the precise output voltage by using a current mode constant off time topology to drive two N channel MOSFETs at a nominal switching frequency of 250kHz The constant off time topology maintains constant
47. Switching Power Supplies Butterworth Heinemann 1995 Keith Billings Switchmode Power Supply Handbook McGraw Hill 1989 George Chryssis High Frequency Switching Power Supplies Theory and Design Second Edition McGraw Hill 1989 Abraham I Pressman Switching Power Supply Design McGraw Hill 1991 Tantalum Electrolytic and Ceramic Families Kemet Electronics Box 5828 Greenville SC 29606 803 963 6300 Type HFQ Aluminum Electrolytic Capacitor and Type V Stacked Polyester Film Capacitor Panasonic 2 Panasonic Way Secaucus NJ 07094 201 348 7000 OS CON Aluminum Electrolytic Capacitor 93 94 Technical Book Sanyo 3333 Sanyo Road Forest City AK 72335 501 633 6634 Ian Clelland Metalized Polyester Film Capacitor Fills High Frequency Switcher Needs PCIM June 1992 Type 5MC Metallized Polycarbonate Capacitor Electronic Concepts Inc Box 1278 Eatontown NJ 07724 908 542 7880 Walt Jung Dick Marsh Picking Capacitors Parts 1 and 2 Audio February March 1980 3 69 SWITCHING REGULATORS 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 29 3 70 Capacitor Manufacturers AVX Corporation 801 17 Ave S Myrtle Beach SC 29577 803 448 9411 Sprague 70 Pembroke Road Concord NH 03301 603 224 1961 Panasonic 2 Panasonic Way Secaucus NJ 07094 201 392 7000 Sanyo Corporation 2001 Sanyo Ave San Diego CA 92173 619 661 6835 Kemet El
48. The equation assumes a voltage of at least a few hundred mV on the collector and ignores Early effects If we take N transistors identical to the first see Figure 6 22 and allow the total current I to be shared equally among them we find that the new base emitter voltage is given by the equation kT I VN In lt d q 6 19 TEMPERATURE SENSORS BASIC RELATIONSHIPS FOR SEMICONDUCTOR TEMPERATURE SENSORS lc lc ONE TRANSISTOR N TRANSISTORS VBE VN kT 3 kT lc VBE In VN In EE t neg E VBE VN KT inn q INDEPENDENT OF Ic 15 Figure 6 22 Neither of these circuits is of much use by itself because of the strongly temperature dependent current Is but if we have equal currents in one BJT and N similar BJTs then the expression for the difference between the two base emitter voltages is proportional to absolute temperature and does not contain Is VBE VN EL lc q q 15 The circuit shown in Figure 6 23 implements the above equation and is known as the Brokaw Cell see Reference 10 The voltage VN appears across resistor R2 The emitter current Q2 is therefore VN R2 The op amp s servo loop and the resistors R force the same current to flow through Q1 The Q1 and Q2 currents are equal and are summed and flow into resistor R1 The corresponding voltage developed across R1 is proportional to absolute temperatur
49. Thermal Technologies Inc com Wo ee C 0 100 200 300 400 500 AIRFLOW LFPM Figure 8 54 Now consider the alternate package the surface mount TO 263 package Because the drain pad connection acts as a heat sink the thermal resistance is a function of the drain pad area on the PC board Figure 8 55 shows the thermal resistance of the package as a function of PC board drain pad area which is acting as the heat sink Note that even with 2 square inches of pad area the thermal resistance is still 30 C W which is well above the calculated maximum allowable value of 11 7 C W 8 55 HARDWARE DESIGN TECHNIQUES The THERMAL RESISTANCE OF TO 263 D PAK VS DRAIN PAD AREA NO AIRFLOW 70 Courtesy AAVID Thermal Technologies Inc 60 NR EUR esses mee C W 50 Nauta Mtn Wa NE c EON dote c AM M mese T mM 30 0 0 5 1 0 1 5 2 0 0 323 645 967 1290 DRAIN PAD PCB AREA in mm Figure 8 55 situation can be improved by the addition of a surface mount heat sink as shown in Figure 8 56 AAVID part number 573300 This heat sink solders to two pads on the PC board which are extensions of the drain pad connecting area The thermal resistance of this combination as a function of airflow is shown in Figure 8 57 Note that with the addition of the surface mount heat sink the thermal resistance of the c
50. Transducer Interfacing Handbook Analog Devices Inc 1980 Walt Kester Editor 1992 Amplifier Applications Guide Section 2 3 Analog Devices Inc 1992 Walt Kester Editor System Applications Guide Section 1 6 Analog Devices Inc 1993 Jim Williams Thermocouple Measurement Linear Technology Application Note 28 Linear Technology Corporation Dan Sheingold Nonlinear Circuits Handbook Analog Devices Inc James Wong Temperature Measurements Gain from Advances in High precision Op Amps Electronic Design 15 May 1986 OMEGA Temperature Measurement Handbook Omega Instruments Inc Handbook of Chemistry and Physics Chemical Rubber Co Paul Brokaw A Simple Three Terminal IC Bandgap Voltage Reference IEEE Journal of Solid State Circuits Vol SC 9 December 1974 HARDWARE MONITORING SECTION 7 HARDWARE MONITORING Walt Kester INTRODUCTION Today s computers require that hardware as well as software operate properly in spite of the many things that can cause a system crash or lockup The purpose of hardware monitoring is to monitor the critical items in a computing system and take corrective action should problems occur Microprocessor supply voltage and temperature are two critical parameters If the supply voltage drops below a specified minimum level further operations should be halted until the voltage returns to acceptable levels In some cases it is desirable to reset the microprocessor under browno
51. V1 and V2 respectively When the switch is closed an impulse of current flows and the charge is redistributed The total charge on the parallel combination of the two capacitors is C1 V1 C2 V2 This charge is distributed between the two capacitors so the new voltage across the parallel combination is equal to qT C1 C2 or vp 2T ANE OVE C2 e CI C2 CLC2 C1 C2 This principle may be used in the simple charge pump circuit shown in Figure 4 6 Note that this circuit is neither a doubler nor inverter but only a voltage replicator The pump capacitor is C1 and the initial charge on C2 is zero The pump capacitor is initially charged to When it is connected to C2 the charge is redistributed and the output voltage is VIN 2 assuming C1 C2 On the second transfer cycle the output voltage is pumped to V N 2 VIN 4 On the third transfer cycle the output voltage is pumped to VIN 2 VIN 4 VIN 8 The waveform shows how the output voltage exponentially approaches VIN 4 5 SWITCHED CAPACITOR VOLTAGE CONVERTERS CHARGE REDISTRIBUTION BETWEEN CAPACITORS q Ct V1 qz C2 V2 CONSERVATION V2 CHARGE C1 C2 C1 V1 C2 V2 EET Hs y C1 C2 C1 C2 Figure 4 5 CONTINUOUS SWITCHING 7 Vin oe p eaten 0 t Figure 4 6 4 6 SWITCHED CAPACITOR VOLTAGE CONVERTERS Figure 4 7 shows a pump capaci
52. VOUT which in this case is 1V For the NDP6020P used in Fig 2 43 see Reference 10 this device achieves an Rpg Qn of 70 milliohms max with a of 2 7V a voltage drive appreciably less than the ADP3310 s VGg DRIVE of 5V The dropout voltage VIN of this regulator configuration is the sum of two series voltage drops the FET s drop plus the drop across or VMIN RpsioN Bs In the design here the two resistances are roughly comparable to one another so the net VIN will be 1A x 50 70 milliohms 120mV For a design safety margin use FET with a rated at the required Rpg with a substantial headroom between the applicable ADP3310 VGsg DRIVE and the applicable rating for the FET In the case here there is ample margin with 5V of drive and a Vg of 2 7V It should be borne in mind that the FET s Vg and RDS ON Will change over temperature but for the NDP6020P device even these variations and of 4 5V are still possible with the circuit as shown With a rated minimum DC input of 6V this means that the design is conservative with 5V output In practice the circuit will typically operate with input voltage minimums on the order of VoyT plus the dropout of 120mV or 5 12V Since the NDP6020P is also a fairly low threshold device it will typically operate at lower output voltages down to about 3V In the event the output is shorted to ground the pass device chosen must be able to con
53. Voltage Minimizes Power Loss due to Wiring Resistance B Flexible Multiple Output Voltages Easily Obtained AC Power Transformer Design Easier Only One Winding Required Regulation Not Critical Selective Shutdown Techniques Can Be Used for Higher Efficiency Eliminates Safety Isolation Requirements for DC DC Converters Figure 3 5 Batteries are the primary power source in much of today s consumer and communications equipment Such systems may require one or several voltages and they may be less or greater than the battery voltage Since a battery is a self contained power source power converters seldom require isolation Often then the basic switcher topologies are used and a wide variety of switching regulators are 3 6 SWITCHING REGULATORS available to fill many of the applications Maximum power levels for these regulators typically can range up from as low as tens of milliwatts to several watts Efficiency is often of great importance as it is a factor in determining battery life which in turn affects practicality and cost of ownership Often of even greater importance though often confused with efficiency is quiescent power dissipation when operating at a small fraction of the maximum rated load e g standby mode For electronic equipment which must remain under power in order to retain data storage or minimal monitoring functions but is otherwise shut down most of the time the quiescent dissipation
54. WAVEFORMS Hes AROF T IFR7404 10nF Vin IRF7204 sev SINT e 1 V P DRIVE ES ADP1148 L 68HH 096 3 75V SD IN lg SENSE 4 o a ae ADP3310 3 3 I 1000 Cc c SENSE O rem C1 100uF 2200 1 N DRIVEO iC 1 20 470pF 5 9 1 0 V Figure 8 36 WAVEFORMS FOR ADP1148 BUCK REGULATOR DRIVING ADP3310 LOW DROPOUT REGULATOR ADP1148 OUTPUT ADP3310 INPUT ADP3310 OUTPUT TOME 5222 boma 157005 Chi 200 BED boma Ms 00gs Chi 7 200pV VERTICAL SCALE 10mV DIV VERTICAL SCALE 10mV DIV HORIZ SCALE 5us DIV HORIZ SCALE 5us DIV Figure 8 37 8 38 HARDWARE DESIGN TECHNIQUES ADP3605 5V to 3V 100mA Switched Capacitor Voltage Converter Figure 8 38 shows the application circuit for the ADP3605 switched capacitor voltage converter All three capacitors input output and pump are 10 surface mount tantalum Kemet T491C series Input and output waveforms are shown in Figure 8 39 where the output ripple is approximately 120mV peak to peak The addition of a 10uH 10pF output filter reduced the ripple to approximately 5mV as shown in Figure 8 40 ADP3605 SWITCHED CAPACITOR VOLTAGE CONVERTER APPLICATION CIRCUIT WAVEFORMS T Vin 4 5 TO 6V Vout 3V 100mA ADP3605 C1 C2 100F 16V SURFACE MOUNT TANTALUM KEMET T491C SERIES Figure 8 38 8 39 HARDWARE D
55. a maximum output frequency of 100MHz corresponds to a risetime of 3 5ns and a track carrying this signal greater than 7 inches should be treated as a transmission line 8 75 HARDWARE DESIGN TECHNIQUES LINE TERMINATION SHOULD BE USED WHEN LENGTH OF PCB TRACK EXCEEDS 2 inches ns Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA DIGITAL IC tr ts PCB TRACK LENGTH PCB TRACK LENGTH FAMILY ns inches cm GaAs 0 1 0 2 0 5 ECL 0 75 1 5 3 8 Schottky 6 15 FAST 3 6 15 AS 3 6 15 AC 4 8 20 ALS 6 12 30 LS 8 16 40 TTL 10 20 50 HC 18 36 90 t rise time of signal in ns t fall time of signal in ns For analog signals 9 fmax calculate t t 0 35 fmax Figure 8 73 Equation 8 4 can be used to determine the characteristic impedance of a PCB track separated from a power ground plane by the board s dielectric microstrip transmission line Zo Q 8 in 5 984 Eq 8 4 ley 1 41 0 89w t where dielectric constant of printed circuit board material d thickness of the board between metal layers in mils w width of metal trace in mils and t thickness of metal trace in mils The one way transit time for a single metal trace over a power ground plane can be determined from Eq 8 5 tpd ns ft 1 017 0 475ey 0 67 Eq 8 5 For example a standard 4 layer PCB board might use 8 mil wide 1 o
56. be described as follows 2 38 REFERENCES AND LOW DROPOUT LINEAR REGULATORS In this circuit VREp is defined as a reference voltage existing at the output of a zero impedance divider of ratio R1 R2 In the figure this is depicted symbolically by the dotted unity gain buffer amplifier fed by R1 R2 which has an output of VREF This reference voltage feeds into a series connection of dotted R11 R2 then actual components D1 R3 R4 etc The error amplifier shown here as a gm stage is actually a PNP input differential stage with the two transistors of the pair operated at different current densities so as to produce a predictable PTAT offset voltage Although shown here as a separate block Vog this offset voltage is inherent to a bipolar pair for such operating conditions The causes a current IpTAT to flow in R4 which is simply V IPTAT ve Note that this current also flows in series connected R4 R3 and the Thevenin resistance of the divider R11 R2 so VPTAT IPTAT R4 R1IIR2 The total voltage defined as is the sum of two component voltages VREF where the scaled voltages across R4 and R11 R2 produce a net PTAT voltage and the diode voltage Vp is a voltage As in a standard bandgap reference the PTAT and CTAT components add up to a temperature stable reference voltage of 1 25V In this case however the reference v
57. circuit devoted simply to dampening resonances via power dissipation is called a snubber If the ringing generates EMI RFI problems it may be damped with a suitable RC snubber However this will cause additional power dissipation and reduced efficiency If the load current of a standard buck converter is low enough the inductor current becomes discontinuous The current at which this occurs can be calculated by observing the waveform shown in Figure 3 13 This waveform is drawn showing the inductor current going to exactly zero at the end of the switch off time Under these conditions the average output current is IOUT 2 We have already shown that the peak inductor current is VIN VOUT ston I PEAK L Thus discontinuous operation will occur if VIN V However Vout and VIN are related by t VOUT VIN D VIN LES on to Solving for ton _ VOUT alia dog oS OUE Y on to VIN f 3 14 SWITCHING REGULATORS Substituting this value for tp into the previous equation for Io iT Vi ME 1 50 oes T IOUT lt 2Lf Criteria for discontinuous operation buck converter BUCK CONVERTER POINT OF DISCONTINUOUS OPERATION INDUCTOR CURRENT AND OUTPUT CURRENT IPEAK Vin VOUT VOUT DISCONTINUOUS MODE IF VIN VOUT ton 1 1 OUT lt 2 PEAK 2L VOUT V 1 390 our VIN 1 211 ton toff IOUT lt Figure 3 13 IDEAL STEP UP BOOST CONVERTER The ba
58. cm2 From the standpoint of the source path receptor model the strength of the electric field E surrounding the receptor is a function of transmitted power antenna gain and distance from the source of the disturbance An approximation for the electric field intensity for both near and far field sources in these terms is given by Equation 8 3 E X ss fra Eq 8 3 m d 1 where E Electric field intensity in V m Transmitted power in mW cm2 Ga Antenna gain numerical and d distance from source in meters For example a 1W hand held radio at a distance of 1 meter can generate an electric field of 5 5V m whereas a 10kW radio transmission station located 1km away generates a field smaller than 0 6V m Analog circuits are generally more sensitive to RF fields than digital circuits because analog circuits operating at high gains must be able to resolve signals in the microvolt millivolt region Digital circuits on the other hand are more immune to RF fields because of their larger signal swings and noise margins As shown in Figure 8 62 RF fields can use inductive and or capacitive coupling paths to generate noise currents and voltages which are amplified by high impedance analog instrumentation In many cases out of band noise signals are detected and rectified by these circuits The result of the RFI rectification is usually unexplained offset voltage shifts in the circuit or in the system 8 66 HARDWAR
59. expected temperature range of the filter should be known as ferrite impedance varies with temperature Third the DC current flowing through the ferrite must be known to ensure that the ferrite does not saturate Although models and other analytical tools may prove useful the general guidelines given above coupled with some experimentation with the actual filter connected to the supply output under system load conditions should lead to a proper ferrite selection Using proper component selection low and high frequency band filters can be designed to smooth a noisy switcher s DC output so as to produce an analog ready 5V supply It is most practical to do this over two and sometimes more stages each stage optimized for a range of frequencies A basic stage can be used to carry all of the DC load current and filter noise by 60dB or more up to a 1 10MHz range This larger filter is used as a card entry filter providing broadband filtering for all power entering a PC card Smaller more simple local filter stages are also used to provide higher frequency decoupling right at the power pins of individual stages 8 25 HARDWARE DESIGN TECHNIQUES SWITCHING REGULATOR EXPERIMENTS In order to better understand the challenge of filtering switching regulators a series of experiments were conducted with representative devices see Figure 8 19 The first series of experiments were conducted on a low power switching regulator the ADP3000 The
60. films generally is very high in terms of Q In fact this can cause problems of spurious resonance in filters requiring damping components 8 21 HARDWARE DESIGN TECHNIQUES Typically using a wound layer type construction film capacitors can be inductive which can limit their effectiveness for high frequency filtering Obviously only non inductively made film caps are useful for switching regulator filters One specific style which is non inductive is the stacked film type where the capacitor plates are cut as small overlapping linear sheet sections from a much larger wound drum of dielectric plate material This technique offers the low inductance attractiveness of a plate sheet style capacitor with conventional leads see References 4 5 6 Obviously minimal lead length should be used for best high frequency effectiveness Very high current polycarbonate film types are also available specifically designed for switching power supplies with a variety of low inductance terminations to minimize ESL Reference 7 Dependent upon their electrical and physical size film capacitors can be useful at frequencies to well above 10MHz At the highest frequencies only stacked film types should be considered Some manufacturers are now supplying film types in leadless surface mount packages which eliminates the lead length inductance Ceramic is often the capacitor material of choice above a few MHz due to its compact size low loss and ava
61. for low frequency circuits and the socket pins allow easy point to point wiring There is a commercial breadboarding system which has most of the advantages of the above techniques robust ground screening ease of circuit alteration low capacitance and low inductance and several additional advantages it is rigid components are close to the ground plane and where necessary node capacitances and line impedances can be calculated easily This system is made by Wainwright Instruments and is available in Europe as Mini Mount and in the USA where the trademark Mini Mount is the property of another company as Solder Mount Reference 8 Solder Mount consists of small pieces of PCB with etched patterns on one side and contact adhesive on the other These pieces are stuck to the ground plane and components are soldered to them They are available in a wide variety of patterns including ready made pads for IC packages of all sizes from 3 pin SOT 23 packages to 64 pin DILs strips with solder pads at intervals which intervals range from 0 040 to 0 25 the range includes strips with 0 1 pad spacing which may be used to mount DIL devices strips with conductors of the correct width to form microstrip transmission lines 500 600 75Q or 1000 when mounted on the ground plane and a variety of pads for mounting various other components Self adhesive tinned copper strips and rectangles LO PADS are also available as tie points for connectio
62. gain phase characteristics 2 35 REFERENCES AND LOW DROPOUT LINEAR REGULATORS If the two poles of such a system are widely separated in terms of frequency stability may not be a serious problem The emitter follower output of a classic regulator like the LM309 is an example with widely separated pole frequencies as the very low ZouT of the NPN follower pushes the output pole due to load capacitance far out in frequency where it does little harm The internal compensation capacitance C1 of Fig 2 27 again then forms part of a dominant pole which reduces loop gain to below unity at the much higher frequencies where the output pole does occur Thus stability is not necessarily compromised by load capacitance in this type of regulator Figure 2 30 summarizes the various DC and AC design issues of LDOs DC AND AC DESIGN ISSUES IN LOW DROPOUT REGULATORS DC AC Lateral PNP Pass Device Two Pole Compensation System High y Vertical PNP Pass Device C ESR Critical to Stability Low Low VMIN B Requires Large C B PMOS Pass Device Lowest Ignouwp Variation Requires Zoned C ESR Low Max Min ESR Limits Over Time Ampere Level Output and Temperature Currents Figure 2 30 By their nature however LDOs simply can t afford the luxury of emitter follower outputs they must instead operate with pass devices capable of saturation Thus given the existence of two or more pole
63. gated oscillator 3 37 specifications 3 37 ADP3050 switching regulator buck converter application circuit 3 39 NPN switch 3 39 40 specifications 3 40 ADP3153 5 bit programmable synchronous switching regulator controller for Pentium II 3 46 47 schematic diagram 3 47 specifications 3 47 ADP330X anyCAP low dropout regulators 2 38 47 design AC performance 2 40 DC performance 2 39 40 error amplifier 2 39 high gain vertical PNP pass device 2 38 merged amplifier reference design 2 38 voltage calculation 2 39 40 ADP3300 LDO regulator basic circuit 2 44 ADP3310 LDO regulator 8 14 15 controller 2 48 51 circuit 2 50 features 2 48 driven by ADP1148 buck regulator circuit 8 38 waveforms 8 38 sensing resistors 2 53 simplest 2 54 thermal design example 8 52 ADP3603 3604 3605 application circuit 4 17 features 4 17 ripple voltage equations 4 10 4 16 voltage inverter regulated output 4 16 18 voltage regulator boost switched capacitor 4 18 ADP3603 3604 3605 3607 regulator shutdown feature 4 12 ADP3605 switched capacitor voltage converter 8 39 41 application circuit 8 39 filtered output 8 40 input output waveforms 8 40 ADP3607 application circuit 4 20 regulated voltage circuit diagram 4 21 switched capacitor boost regulator diagram 4 18 specifications 4 19 voltage regulator switched capacitor 4 18 21 ADP3801 3802 battery charging ICs 4 22 5 18 buck battery charger diag
64. generally configured in a four resistor bridge circuit The bridge output is amplified by an instrumentation amplifier for further processing However high resolution measurement ADCs such as the AD77XX series allow the RTD output to be digitized directly In this manner linearization can be performed digitally thereby easing the analog circuit requirements 6 13 TEMPERATURE SENSORS FOUR WIRE OR KELVIN CONNECTION TO Pt RTD FOR ACCURATE MEASUREMENTS FORCE SENSE R LEAD LEAD LEAD 1000 TO HIGH Z PtRTD OR ADC SENSE LEAD Figure 6 15 Figure 6 16 shows a 1000 Pt RTD driven with a 400pA excitation current source The output is digitized by one of the AD77XX series ADCs Note that the RTD excitation current source also generates the 2 5V reference voltage for the ADC via the 6 25kQ resistor Variations in the excitation current do not affect the circuit accuracy since both the input voltage and the reference voltage vary ratiometrically with the excitation current However the 6 25kQ resistor must have a low temperature coefficient to avoid errors in the measurement The high resolution of the ADC and the input PGA gain of 1 to 128 eliminates the need for additional conditioning circuits The ADT70 is a complete Pt RTD signal conditioner which provides an output voltage of 5mV C when using a RTD see Figure 6 17 The Pt RTD and the 1k reference resistor are both excited with 1mA matched current source
65. if linearization techniques are not used It is possible to use a thermistor over a wide temperature range only if the system designer can tolerate a lower sensitivity to achieve improved linearity One approach to linearizing a thermistor is simply shunting it with a fixed resistor Paralleling the thermistor with a fixed resistor increases the linearity significantly As shown in Figure 6 20 the parallel combination exhibits a more linear variation with temperature compared to the thermistor itself Also the sensitivity of the combination still is high compared to a thermocouple or RTD The primary 6 17 TEMPERATURE SENSORS disadvantage to this technique is that linearization can only be achieved within a narrow range LINEARIZATION OF NTC THERMISTOR USING 5 17kQ SHUNT RESISTOR 40 30 RESISTANCE 20 THERMISTOR PARALLEL COMBINATION 10 0 20 40 60 80 100 TEMPERATURE C Figure 6 20 The value of the fixed resistor can be calculated from the following equation _ RT2 RT1 2 RTI RTI RT3 2 RT2 R gt where is the thermistor resistance at T1 the lowest temperature in the measurement range RTS is the thermistor resistance at the highest temperature in the range and RT2 is the thermistor resistance at T2 the midpoint T2 T1 T3 2 For a typical 10kQ NTC thermistor 32 6500 at 0 C RT2 6 5320 at 35 C and 1 7520 at 70 C This
66. input voltage the same or the opposite polarity Consider the basic components of a switcher as stated above The inductor and capacitor are ideally reactive elements which dissipate no power The transistor is effectively ideally a switch in that it is either on thus having no voltage dropped across it while current flows through it or off thus having no current flowing through it while there is voltage across it Since either voltage or current are always zero the power dissipation is zero thus ideally the switch dissipates no power Finally there is the diode which has a finite voltage drop while current flows through it and thus dissipates some power But even that can be substituted with a synchronized switch called a synchronous rectifier so that it ideally dissipates no power either Switchers also offer the advantage that since they inherently require a magnetic element it is often a simple matter to tap an extra winding onto that element and often with just a diode and capacitor generate a reasonably well regulated additional output If more outputs are needed more such taps can be used Since the tap winding requires no electrical connection it can be isolated from other circuitry or made to float atop other voltages Of course nothing is ideal and everything has a price Inductors have resistance and their magnetic cores are not ideal either so they dissipate power Capacitors have
67. is the largest determinant of battery life Although efficiency may indicate power consumption for a specific light load condition it is not the most useful way to address the concern For example if there is no load on the converter output the efficiency will be zero no matter how optimal the converter and one could not distinguish a well power managed converter from a poorly managed one by such a specification The concern of managing power effectively from no load to full load has driven much of the technology which has been and still is emerging from today s switching regulators and controllers Effective power management as well as reliable power conversion is often a substantial factor of quality or noteworthy distinction in a wide variety of equipment The limitations and cost of batteries are such that consumers place a value on not having to replace them more often than necessary and that is certainly a goal for effective power conversion solutions TYPICAL APPLICATION OF A BOOST REGULATOR IN BATTERY OPERATED EQUIPMENT STEP UP Vout gt VBATTERY BOOST SWITCHING REGULATOR VBATTERY Figure 3 6 3 7 SWITCHING REGULATORS INDUCTOR AND CAPACITOR FUNDAMENTALS In order to understand switching regulators the fundamental energy storage capabilities of inductors and capacitors must be fully understood When a voltage is applied to an ideal inductor see Figure 3 7 the current builds up linearly over time at a ra
68. it is common practice to eliminate the additional thermocouple wire and terminate the thermocouple leads in the isothermal block in the arrangement shown in Figure 6 9 The Metal A Copper and the Metal B Copper junctions if at the same temperature are equivalent to the Metal A Metal B thermocouple junction in Figure 6 8 TERMINATING THERMOCOUPLE LEADS DIRECTLY TO AN ISOTHERMAL BLOCK phase E COPPER V OUT V1 V 0 C METAL A COPPER T1 TEMPERATURE COMPENSATION CIRCUIT METAL B COPPER T2 ERES ISOTHERMAL BLOCK Figure 6 9 The circuit in Figure 6 10 conditions the output of a Type K thermocouple while providing cold junction compensation for temperatures between 0 C and 250 C The circuit operates from single 3 3V to 12V supplies and has been designed to produce an output voltage transfer characteristic of 10mV C thermocouple exhibits a Seebeck coefficient of approximately 41 2 therefore at the cold junction the TMP35 voltage output sensor with a temperature coefficient of 10mV C is used with R1 and R2 to introduce an opposing cold junction temperature coefficient of 41yV C This prevents the isothermal cold junction connection between the circuit s printed circuit board traces and the thermocouple s wires from introducing an error in the measured temperature This compensation works extremely well for circuit ambient temperatures in the range o
69. low pass filter made up of a single capacitor and inductor can begin to leak when the applied signal frequency is 100 to 1000 higher than the filter s cutoff frequency For example a 10kHz LPF would not be considered very efficient at filtering frequencies above 1MHz 8 67 HARDWARE DESIGN TECHNIQUES KEEPING RFI AWAY FROM ANALOG CIRCUITS Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA REMOTE LOCAL 77 VNEG B Decouple all voltage supplies to analog chip with high frequency capacitors B Use high frequency filters on all lines that leave the board B Use high frequency filters on the voltage reference if it is not grounded Figure 8 63 A SINGLE LOW PASS FILTER LOSES EFFECTIVENESS 100 1000 Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA TYPICALLY 100 1000 FILTER ATTENUATION N sik f3dB FREQUENCY Figure 8 64 8 68 HARDWARE DESIGN TECHNIQUES Rather than use one LPF stage it is recommended that the interference frequency bands be separated into low band mid band and high band and then use individual filters for each band Kimmel Gerke Associates use the stereo speaker analogy of woofer midrange tweeter for RFI low pass filter design illustrated in Figure 8 65 In this approach low frequencies are grouped from 1
70. make each of them as short as possible This leads to the arrangement shown in Figure 8 13 where each critical ground connection is made directly to the ground plane with the shortest connection length possible By physically locating all critical components associated with the regulator close together and making the ground connections short stray series inductance and resistance are minimized It is true that several small ground loops may occur using this approach but they should not cause significant system problems because they are confined to a very small area of the overall ground plane Refer back to Figure 8 9 ADP1148 switching regulator 8 16 HARDWARE DESIGN TECHNIQUES evaluation board ground plane side and note that this approach to grounding was used GROUNDING AND SIGNAL ROUTING TECHNIQUES FOR SWITCHING REGULATORS METHOD 2 Voc SWITCHING REGULATOR CONTROLLER SGND PGND a SHORT CONNECTION TO GROUND PLANE COMMON SHORT HEAVY TRACES Figure 8 13 8 17 HARDWARE DESIGN TECHNIQUES REFERENCES ON GROUNDING 8 18 High Speed Design Techniques Analog Devices 1996 Chapter 7 Walt Kester A Grounding Philosophy for Mixed Signal Systems Electronic Design Special Analog Issue June 23 1997 p 29 HARDWARE DESIGN TECHNIQUES POWER SUPPLY NOISE REDUCTION AND FILTERING Walt Jung Walt Kester Bill Chestnut Precision analog circuitry has traditionally been powered from well regulated lo
71. may be less than 100kSPS Successful prototyping of these circuits requires that equal attention be given to good high speed and high precision circuit techniques Switching regulators also fall into the high speed catagory Even though their desired output is a DC voltage low output ripple voltage is highly dependent upon the use of proper high speed grounding layout and decoupling techniques The simplest technique for analog prototyping uses a solid copper clad board as a ground plane Reference 5 and 6 The ground pins of the ICs are soldered directly to the plane and the other components are wired together above it This allows HF decoupling paths to be very short All lead lengths should be as short as possible and signal routing should separate high level and low level signals Connection wires should be located close to the surface of the board to minimize the possibility of stray inductive coupling In most cases 18 gauge or larger insulated wire should be used Parallel runs should not be bundled because of possible coupling Ideally the layout at least the relative placement of the components on the board should be similar to the layout to be used on the final PCB This approach is often referred to as deadbug prototyping because the ICs are often mounted upside down with their leads up in the air with the exception of the ground pins which are bent over and soldered directly to the ground plane The upside down ICs look like dece
72. necessary in a system making absolute 16 bit measurements Note that many systems make relative measurements rather than absolute ones and in such cases the absolute accuracy of the reference is not important although noise and short term stability may be Figure 2 1 summarizes some key points of the reference selection process Temperature drift or drift due to aging may be an even greater problem than absolute accuracy The initial error can always be trimmed but compensating for drift is difficult Where possible references should be chosen for temperature coefficient and aging characteristics which preserve adequate accuracy over the operating temperature range and expected lifetime of the system Noise in voltage references is often overlooked but it can be very important in system design It is generally specified on data sheets but system designers frequently ignore the specification and assume that voltage references do not contribute to system noise There are two dynamic issues that must be considered with voltage references their behavior at start up and their behavior with transient loads With regard to the first always bear in mind that voltage references do not power up instantly this is true of references inside ADCs and DACs as well as discrete designs Thus it is rarely possible to turn on an ADC and reference whether internal or external make 2 1 REFERENCES AND LOW DROPOUT LINEAR REGULATORS a reading and tu
73. on the switch for that cycle or not The hysteresis of the comparator tends to give rise to several cycles of switching followed by several cycles of not switching Hence the 3 27 SWITCHING REGULATORS resulting switching signal is characterized by pulses which tend to come in bursts hence the name for the modulation technique There are at least two inherent fundamental drawbacks of the PBM switch modulation technique First the constant variation of the duty cycle between zero and maximum produces high ripple currents and accompanying losses Second there is an inherent generation of subharmonic frequencies with respect to the oscillator frequency This means that the noise spectrum is not well controlled and often audible frequencies can be produced This is often apparent in higher power converters which use pulse skipping to maintain short circuit current control An audible noise can often be heard under such a condition due to the large magnetics acting like speaker coils For these reasons PBM is seldom used at power levels above 10 Watts But for its simplicity it is often preferred below that power level but above a power level or with a power conversion requirement where charge pumps are not well suited CONTROL TECHNIQUES Though often confused with or used in conjunction with discussing the switch modulation technique the control technique refers to what parameters of operation are used and how they are used to cont
74. operation Vout 2Lf IOUT lt boost converter The basic buck and boost converter circuits can work equally well for negative inputs and outputs as shown in Figure 3 19 Note that the only difference is that the polarities of the input voltage and the diode have been reversed In practice however not many IC buck and boost regulators or controllers will work with negative inputs In some cases external circuitry can be added in order to handle negative inputs and outputs Rarely are regulators or controllers designed specifically for negative inputs or outputs In any case data sheets for the specific ICs will indicate the degree of flexibility allowed NEGATIVE IN NEGATIVE OUT BUCK AND BOOST CONVERTERS Vin Vout Vin Vout BUCK BOOST Figure 3 19 3 20 SWITCHING REGULATORS BUCK BOOST TOPOLOGIES The simple buck converter can only produce an output voltage which is less than the input voltage while the simple boost converter can only produce an output voltage greater than the input voltage There are many applications where more flexibility is required This is especially true in battery powered applications where the fully charged battery voltage starts out greater than the desired output the converter must operate in the buck mode but as the battery discharges its voltage becomes less than the desired output the converter must then operate in the boost mode A buck boost converter is capable of produc
75. operation Frequency components may fall into the audio band so proper filtering of the output of such a regulator is mandatory Selection of the inductor value is also more critical in PBM regulators Because the regulation is accomplished with a burst of fixed duty cycle pulses i e higher than needed on average followed by an extended off time the energy stored in the inductor during the burst of pulses must be sufficient to supply the required energy to the load If the inductor value is too large the regulator may never start up or may have poor transient response and inadequate line and load regulation On the other hand if the inductor value is too small the inductor may saturate during the charging time or the peak inductor current may exceed the maximum rated switch 3 33 SWITCHING REGULATORS current However devices such as the ADP3000 incorporate on chip overcurrent protection for the switch An additional feature allows the maximum peak switch current to be set with an external resistor thereby preventing inductor saturation Techniques for selecting the proper inductor value will be discussed in a following section DIODE AND SWITCH CONSIDERATIONS So far we have based our discussions around an ideal lossless switching regulator having ideal circuit elements In practice the diode switch and inductor all dissipate power which leads to less than 100 efficiency Figure 3 31 shows typical buck and boost converte
76. or 5W to the load With a dropout voltage of 1V the input power is 6V times the same 1A or 6W In terms of power efficiency this can be calculated as PRFF 100 PUE where POUT and PrN the total output and input powers respectively In these sample calculations the relatively small portion of power related to Iground will be ignored for simplicity since this power is relatively small In an actual design this simplifying step may not be justified In the case shown the efficiency would be 100 x 5 6 or about 83 But by contrast if an LDO were to be used with a dropout voltage of 0 1V instead of 1V the input voltage can then be allowed to go as low as 5 1V The new efficiency for this condition then becomes 100 x 5 5 1 or 98 It is obvious that an LDO can potentially greatly enhance the power efficiency of linear voltage regulator systems A more detailed look within a typical regulator block diagram reveals a variety of elements as is shown in Figure 2 24 In this diagram virtually all of the elements shown can be considered to be fundamentally necessary the exceptions being the shutdown control and saturation sensor functions shown dotted While these are present on many current regulators the shutdown feature is relatively new as a standard function and certainly isn t part of standard three terminal regulators When present shutdown control is a logic level controllable input whereby a digital HIGH
77. out by the B H curve during one complete operating cycle is the hysteresis loss exhibited by the core during that cycle Hysteresis loss is a function of core material core volume operating frequency and the maximum flux density during each cycle The second major loss within the core is eddy current loss This loss is caused by the flow of circulating magnetic currents 3 57 SWITCHING REGULATORS within the core material caused by rapid transitions in the magnetic flux density It is also dependent on the core material core volume operating frequency and flux density In addition to core loss there is winding loss the power dissipated in the DC resistance of the winding This loss is a function of the wire size core volume and the number of turns In a switching regulator application excessive loss will result in a loss of efficiency and high inductor operating temperatures INDUCTOR POWER LOSSES B Loss FUNCTION OF Magnetic Hysteresis Core Material Core Volume Flux Density Frequency Eddy Currents Core Material Core Volume Flux Density Frequency E Winding Resistance Wire Size Number of Turns Core Volume E Figure of Merit Ec Figure 3 57 Fortunately inductor manufacturers have simplified the design process by specifying maximum peak current maximum continuous current and operating frequency range and temperature for their inductors If the designer derates the maximu
78. pass device appears in series with the input preventing its saturation and thus setting VIN of about 1 or 2V By contrast the inverting mode device connections of both columns C and E do allow the pass device to be effectively saturated which lowers the associated voltage losses to a minimum This single factor makes these two pass device types optimum for LDO use at least in terms of power efficiency 2 29 REFERENCES AND LOW DROPOUT LINEAR REGULATORS PASS DEVICES USEFUL IN VOLTAGE REGULATORS Vin Vin 22V 1V Vout Vout a SINGLE NPN b DARLINGTON NPN O ViN Vin Vin i Vain Vmin Vce sat Va 1 5V mL E 1 Vout 9 me Vout d PNP NPN c SINGLE PNP e PMOS Figure 2 25 PROS AND CONS OF VOLTAGE REGULATOR PASS DEVICES 2 30 A B D E SINGLE DARLINGTON SINGLE PNP NPN PMOS NPN NPN PNP Vmin 1V Vmin 2V Vmin 0 1V Vm 1 5 Vin Roscony IL IL lt 1A I gt 1A 1 1A gt 1A l gt 1A Follower Follower Inverter Inverter Inverter Low Zout Low Zout High Zour High Zour High Zour Wide BW Wide BW Narrow BW Narrow BW Narrow BW C Immune C Immune C Sensitive C Sensitive C Sensitive Figure 2 26 REFERENCES AND LOW DROPOUT LINEAR REGULATORS For currents below 1A either a single PNP or a PMOS pass device is most useful for low dropout and they both can achieve a VMIN of 0 1V or less at currents of 10
79. potential instability it brings is a major deterrent to easily applying LDOs While low dropout goals prevent the use of emitter follower type outputs and so preclude their desirable buffering effect against cap loading there is an alternative technique of providing load immunity One method of providing a measure of insusceptibility against variation in a particular amplifier response pole is called pole splitting see Reference 8 It refers to an amplifier compensation method whereby two response poles are shifted in such a way so as to make one a dominant lower frequency pole In this manner the secondary pole which in this case is the related output pole becomes much less of a major contributor to the net AC response This has the desirable effect of greatly de sensitizing the amplifier to variations in the output pole A Basic Pole Splitting Topology A basic LDO topology with frequency compensation as modified for pole splitting is shown in Figure 2 33 Here the internal compensation capacitor is connected as an integrating capacitor around pass device Q1 C1 is the pass device input capacitance While it is true that this step will help immunize the regulator to the CT related pole it also has a built in fatal flaw With CcoMp connected directly to the Q1 base as shown the line rejection characteristics of this setup will be quite poor In effect when doing it this way one problem sensitivity will be exchange
80. produce a control loop with a fast response time A popular way to circumvent the problem produced by the LC filter phase shift is to use current mode CM control as shown in Figure 3 28 In current mode control it is still desirable of course to regulate the output voltage Thus an error amplifier G1 is still required However the switch modulation is no longer controlled directly by the error amplifier Instead the inductor current is sensed amplified by G2 and used to modulate the switch in accordance with the command signal from the output voltage error amplifier It should be noted that the divider network G1 and G2 are usually part of the IC switching regulator itself rather than external as shown in the simplified diagram 3 29 SWITCHING REGULATORS CURRENT FEEDBACK FOR PWM CONTROL SWITCHING REG IC INDUCTOR DIODE NOTE RESISTORS AMPLIFIERS AND INCLUDED IN SWITCHING REGULATOR IC Figure 3 28 The CM control system uses feedback from both the output voltage and output current Recall that at the beginning of each PWM cycle the switch turns on and the inductor current begins to rise The inductor current develops a voltage across the small sense resistor RgENSE which is amplified by G2 and fed back to the PWM controller to turn off the switch The output voltage sensed by amplifier G1 and also fed back to the PWM controller sets the level at which the peak inductor current will ter
81. regulator solution handles up to 10A Advantages compared to integrated solutions High accuracy 1 5 fixed voltages 2 8 3 3 3 or 5V User flexibility selection of FET for performance Small footprint with anyCAP controller and SMD FET Kelvin output sensing possible Integral low loss current limit sensing for protection Figure 2 41 Regulator Controller Differences An obvious basic difference of the regulator controller versus a stand alone regulator is the removal of the pass device from the regulator chip This design step has both advantages and disadvantages A positive is that the external PMOS pass device can be chosen for the exact size package current rating and power handling which is most useful to the application This approach allows the same basic controller IC to be useful for currents of several hundred mA to more than 10A simply by choice of the FET Also since the regulator controller IC s Iground of 800uUA results is very little power dissipation its thermal drift will be enhanced On the downside there are two packages now used to make up the regulator function And current limiting which can be made completely integral to a standalone IC LDO regulator is now a function which must be split between the regulator controller IC and an external sense resistor This step also increases the dropout voltage of the LDO regulator controller somewhat by about 50mV 2 48 REFERENCES AND LOW DROPOUT LINE
82. regulator was tested in the boost configuration with a 2V input and a 5V 100mA output The prototype board shown previously in Figures 8 4 and 8 5 was used for the tests The second series of experiments were conducted on the ADP3000 configured in the buck mode with a 9V input and a 5V 100mA output Again the prototype board shown previously in Figures 8 4 and 8 5 was used for the tests The third series of experiments involved the ADP1148 synchronous buck regulator with a 9V input and a 3 3V 1A output An evaluation board similar to that shown in Figures 8 8 and 8 9 was used The fourth series of experiments were conducted on the ADP1148 synchronous buck regulator driving an ADP3310 low dropout linear regulator The ADP1148 was configured for a 9V input and a 3 75V 1A output and the ADP3310 for a 3 3V 1A output The fifth series of experiments were made on the ADP3605 5V to 3V switched capacitor voltage converter The ADP3605 output load was set for 100mA FILTERING SWITCHING REGULATOR OUTPUTS SUMMARY OF EXPERIMENTS ADP3000 2V to 5V 100mA Boost Regulator ADP3000 9V to 5V 100mA Buck Regulator ADP1148 9V to 3 3V 1A Buck Regulator ADP1148 9V to 3 75V 1A Buck Regulator with ADP3310 3 3V 1A Linear LDO Post Regulator ADP3605 5V to 3V 100mA Switched Capacitor Voltage Converter Figure 8 19 In addition to observing typical input and output waveforms the objective of these experiments was to reduce the output ripple to
83. resistances associated with each switch as well as the ESRs of the capacitors The quiescent power dissipation Ig VIN must also be included where Ig is the quiescent current drawn by the IC itself VOLTAGE INVERTER POWER LOSSES PLoss loUT VN VouT aVin louT Rout qVin Rout 8Rsw 4ESRc1 a ESRC2 Figure 4 12 4 11 SWITCHED CAPACITOR VOLTAGE CONVERTERS The power dissipated in the switching arm is first calculated When C1 is connected to VIN current of 2I9jyT flows through the switch resistances 2Rgw and the ESR of C1 ESR C When is connected to the output a current of 219pT continues to flow through C1 2RSw and ESR Therefore there is always an rms current of 2IoyT flowing through these resistances resulting in power dissipation in the switching arm of Pow IOUT x 2Rgw ESRC IoyT x 8Rgw 4ESR 1 In addition to these purely resistive losses an rms current of Lout flows through the resistance of the switched capacitor C1 yielding an additional loss of 2 n IouT x Pc1 IOUT The rms current flowing through ESR 2 is IQUT yielding a power dissipation of PESRGs loUT x ESRC2 Adding all the resistive power dissipations to the quiescent power dissipation yields PLOSS x SRsw AESR ESRC zu IqVIN All of the resistive losses can be grouped into an equivalent as shown in the diagram Rout 8Rgw
84. same over the entire length of the cable To protect circuits against low frequency electric field pickup only one end of the shield should be returned to a low impedance point A generalized example of this mechanism is illustrated in Figure 8 78 8 82 HARDWARE DESIGN TECHNIQUES LENGTH OF SHIELDED CABLES DETERMINES AN ELECTRICALLY LONG OR ELECTRICALLY SHORT APPLICATION Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA SHIELDED ENCLOSURE B SHIELDED ENCLOSURE A LENGTH SHIELDED CABLE FULLY SHIELDED ENCLOSURES CONNECTED BY FULLY SHIELDED CABLE KEEP ALL INTERNAL CIRCUITS AND SIGNAL LINES INSIDE THE SHIELD TRANSITION REGION 1 20 WAVELENGTH Figure 8 77 CONNECT THE SHIELD AT ONE POINT AT THE LOAD TO PROTECT AGAINST LOW FREQUENCY 50 60Hz Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA CAPACITIVE COUPLING CABLE SHIELD TO CABLE RECEIVER GROUNDED ATLOAR RECEIVER Crile G IL di a Figure 8 78 8 83 HARDWARE DESIGN TECHNIQUES In this example the shield is grounded at the receiver An exception to this approach which will be highlighted again later is the case where line level gt 1Vrms audio signals are transmitted over l
85. scaling factor of 1 0 is to be used for a Ip the Rg calculation is straightforward and 50 milliohms is the correct Rg value 2 53 REFERENCES AND LOW DROPOUT LINEAR REGULATORS However to account for uncertainties in the threshold voltage and to provide a more conservative output current margin a scaling factor of Kp 1 5 can alternately be used When this approach is used the same 1A I load conditions will result in 33 milliohm Rg value In essence the use of the 1 5 scaling factor takes into account the foldback scheme s reduction in output current allowing higher current in the limit mode The simplest and least expensive sense resistor for high current applications such as Figure 2 43 is a copper PCB trace controlled in both thickness and width Both the temperature dependence of copper and the relative size of the trace must be taken into account in the resistor design The temperature coefficient of resistivity for copper has a positive temperature coefficient of 0 39 C This natural copper in conjunction with the controller s PTAT based current limit threshold voltage can provide for a current limit characteristic which is simple and effective over temperature The table of Figure 2 44 provides resistance data for designing PCB copper traces with various PCB copper thickness or weight in ounces of copper per square foot area To use this information note that the center column contains a resistance
86. side of a switching regulator prototype based on the ADP3000 A CAD system was used in the layout and the board was fabricated using a PC board milling machine board cutter The size of the board is approximately 2 5 by 3 5 The input to the regulator is on the left hand side of the board top view and it is decoupled directly to the ground plane with three 33pF 16V tantalum surface mount capacitors Note that the connections are directly from the input pad to the ground plane for minimum parasitic series resistance and inductance The ADP3000 IC is mounted in a low profile socket on the bottom side of the board near the center The external energy transfer inductor is located in the upper part of the board slightly to the right of the center of the board It is mounted in an encapsulated plastic package suitable for surface mounting The output of the ADP3000 is decoupled with a 8 6 HARDWARE DESIGN TECHNIQUES 33yF 16V surface mount capacitor and is followed by an LC filter before connecting to the output load Notice that all connections are short especially those to the surface mount capacitors This isolates the high speed switching currents to a small area and prevents interference with other circuits which the regulator may be supplying HANDWIRED PROTOTYPE ADP3000 TOP VIEW Figure 8 4 In Figure 8 5 bottom view note that the ADP3000 is mounted in a low profile IC socket for convenience The catch diode is located
87. specific recommendations are given in the data sheet for each device 4 7 SWITCHED CAPACITOR VOLTAGE CONVERTERS UNREGULATED SWITCHED CAPACITOR INVERTER AND DOUBLER IMPLEMENTATIONS An unregulated switched capacitor inverter implementation is shown in Figure 4 8 Notice that the SPDT switches shown in previous diagrams actually comprise two SPST switches The control circuit consists of an oscillator and the switch drive signal generators Most IC switched capacitor inverters and doublers contain all the control circuits as well as the switches and the oscillator The pump capacitor C1 and the load capacitor C2 are external Not shown in the diagram is a capacitor on the input which is generally required to ensure low source impedance at the frequencies contained in the switching transients The switches used in IC switched capacitor voltage converters may be CMOS or bipolar as shown in Figure 4 9 Standard CMOS processes allow low on resistance MOSFET switches to be fabricated along with the oscillator and other necessary control circuits Bipolar processes can also be used but add cost and increase power dissipation SWITCHED CAPACITOR VOLTAGE INVERTER IMPLEMENTATION 793 OSCILLATOR AND SWITCH DRIVE CIRCUITS Figure 4 8 4 8 SWITCHED CAPACITOR VOLTAGE CONVERTERS SWITCHES USED IN VOLTAGE CONVERTERS MOSFET BIPOLAR SWITCHES SWITCHES P CH N CH PNP NPN Figure 4 9 VOLTAGE INVERTER AND DOUBLER DYNAM
88. specified value i e 4 2V per cell The voltage control loop has an accuracy of 1 required by Li Ion batteries At this point the control switches from the current control loop to the voltage control loop VSENSE and the battery is charged with a constant voltage until charging is complete In addition the ADP3810 has an overvoltage comparator which stops the charging process if the battery voltage exceeds 646 of its programmed value This function protects the circuitry should the battery be removed during charging In addition if the supply voltage drops below 2 7V the charging is stopped by the undervoltage lockout UVLO circuit The ADP3811 is identical to the ADP3810 except that the VSENSE input ties directly to the GM2 stage input and R1 R2 are external allowing other voltages to be programmed by the user for battery chemistries other than Li Ion A simplified functional diagram of a battery charger based on the ADP3810 3811 battery charger controller is shown in Figure 5 16 The ADP3810 3811 controls the DC DC converter which can be one of many different types such as a buck flyback or linear regulator The ADP3810 3811 maintains accurate control of the current and voltage loops ADP3810 3811 SIMPLIFIED BATTERY CHARGER O IN ViN DC DC VOLTAGE CONVERTER CURRENT CTRL GND LOOP Rcs INTERNAL FOR ADP3810 Ves CHARGE CURRENT OUT ADP3810 3811 CONTROL CIRCUITS GND Figure 5 16 5 12
89. substrate reaches point B in temperature the external fan will be turned on to create the air stream and lower the temperature If the overall system setup is reasonable in terms of thermal profiling this small IC can thus be used to indirectly control another larger and independent power source with regard to its temperature Note that the dual mode control need not necessarily be used in all applications An unused comparator is simply wired high or low Figure 6 38 shows a circuit diagram using the TMP12 as a general purpose controller The device is connected to a 5V supply which is also used to power a control relay and the TMP12 s internal heater at pin 5 Setpoint programming of the TMP12 is accomplished by the resistor string at pins 4 through 1 R1 R3 These resistors establish a current drain from the internal reference source at pin 4 which sets up a reference current IREF which is set as IREF 5uA C x THYS 74A In this expression THYS is the hysteresis temperature swing desired about the setpoint in C and the 7uA is recommended minimum loading of the reference For a 2 C hysteresis for example IREF is 174A for 5 C it would be 32pA Given a desired setpoint temperature in C the setpoint can be converted to a corresponding voltage Although not available externally the internal temperature dependent voltage of the TMP12 is scaled at 5mV C and is equal to 1 49V at 25 C To convert a setpoint temperature t
90. that the ESL is 20nH ESR and ESL vary widely between manufacturers and are also dependent upon body style through hole vs surface mount but these values will serve to illustrate the point RESPONSE OF CAPACITOR TO CURRENT STEP i INPUT O QO Y RRENT oss oy Ts dt 100ns Eon Stee 0 Equivalent f 3 5MHz ESL 20nH di VPEAK ESL g ESR o IPEAK 400 C 100 OUTPUT Xc 0 00050 YOETAGE 3 5MHz _ ESR Ipeak 200mV lt Figure 3 60 Assume that the actual value of the capacitor is large enough so that its reactance is essentially a short circuit with respect to the step function input For example 100 at 3 5MHz the equivalent frequency of a 100ns risetime pulse has a reactance of 1 2nfC 0 00050 In this case the output voltage ripple is determined exclusively by the ESR and ESL of the capacitor not the actual capacitor value itself These waveforms show the inherent limitations of electrolytic capacitors used to absorb high frequency switching pulses In a practical system the high frequency components must be attenuated by low inductance ceramic capacitors with low ESL or by the addition of an LC filter Figure 3 61 shows the impedance versus frequency for a typical 100yF electrolytic capacitor having an ESR of 0 2Q and an ESL of 20nH At frequencies below about 10kHz the capacitor is nearly ideal Between 10kHz and 1MHz the range of switching frequencies for
91. the LDO Dropout voltage is defined simply as that minimum input output differential where the regulator undergoes a 2 reduction in output voltage For example if a nominal 5 0V LDO output drops to 4 9V 2 under conditions of an input output differential of 0 5V by this definition the LDO s VMIN is 0 5V As will be shown in this section dropout voltage is extremely critical to a linear regulator stage s power efficiency The lower the voltage allowable across a regulator while still maintaining a regulated output the less power the regulator dissipates as a result A low regulator dropout voltage is the key to this as it takes this lower dropout to maintain regulation as the input voltage lowers In performance terms the bottom line for LDOs is simply that more useful power is delivered to the load and less heat is generated in the regulator LDOs are key elements of power systems that must provide stable voltages from batteries such as portable computers cellular phones etc This is simply because they maintain their regulated output down to lower points on the battery s discharge curve Or within classic mains powered raw DC supplies LDOs allow lower transformer secondary voltages reducing system susceptibility to shutdown under brownout conditions as well as allowing cooler operation LINEAR VOLTAGE REGULATOR BASICS A brief review of three terminal linear IC regulator fundamentals is necessary to understanding th
92. the resistor s input sense terminal and the IS connection trace should also connect close to the resistor body or the resistor s output sense terminal Four terminal wiring is increasingly important for output currents of 1A or more Alternately an appropriate selected sense resistor such as surface mount sense devices available from resistor vendors can be used see Reference 13 Sense resistor Rg may not be needed in all applications if a current limiting function is provided by the circuit feeding the regulator For circuits that don t require current limiting the IS and pins of the ADP3310 must be tied together PCB Layout Issues For best voltage regulation place the load as close as possible to the controller device s Vout and GND pins Where the best regulation is required the VOUT trace from the ADP3310 and the pass device s drain connection should connect to the positive load terminal via separate traces This step Kelvin sensing will keep the heavy load currents in the pass device s drain out of the feedback sensing path and thus maximize output accuracy Similarly the unregulated input common should connect to the common side of the load via a separate trace from the ADP3310 GND pin These points are summarized in the Techniques discussion of section 8 around Figures 8 10 and 8 11 specifically A 2 8V 8A LDO Regulator Controller With seemingly minor changes to the basic 1A LDO circuit used in Fig 2 43 an
93. thermal resistance device attached externally to a semiconductor part to aid in heat removal It will have some additional thermal resistance of its own also rated in C W Rather than just a single number 0 in this case will be composed of more than one component i e 01 02 etc Like series resistors thermal impedances add making a net calculation relatively simple For example to compute a net 0JA given a relevant the thermal resistance of the heat sink or case to ambient is added to the as 9JA 9JC OCA and the result is the 0JA for that specific circumstance A real example illustrating these relationships is shown by Figure 8 47 These curves indicate the maximum power dissipation vs temperature characteristic for a device using standard 8 pin SOIC and a thermal coastline 8 pin SOIC Expressed in this fashion the curves are often referred to as derating curves The proprietary Analog Devices thermal coastline package allows additional power to be dissipated with no increase in package size For a TJ max of 150 C the upper curve shows the allowable power in a thermal coastline package This corresponds to a 0 which can be calculated by dividing the AT by P at any point For example 1W of power is 8 48 HARDWARE DESIGN TECHNIQUES allowed at a TA of 60 C so the AT is 150 C 60 C 90 C Dividing by 1W gives the thermal coastline package s 0 of 90 C W Similarly the standard 8 pin SOI
94. they are available in grades with voltage tolerances of 40 1 or 1 of VOUT with corresponding TC s of 50 or 100ppm C Because of stability requirements devices of the AD1582 series must be used with both an output and input bypass capacitor Recommended worst case values for these are shown in the hookup diagram of Figure 2 8 For the electrical values noted it is likely that tantalum chip capacitors will be the smallest in size 2 8 REFERENCES AND LOW DROPOUT LINEAR REGULATORS AD1582 AD1585 2 5 5V SERIES TYPE BANDGAP REFERENCES HAVE TINY SIZE IN SOT 23 FOOTPRINT Figure 2 7 AD1582 AD1585 SERIES CONNECTION DIAGRAM Figure 2 8 2 9 REFERENCES AND LOW DROPOUT LINEAR REGULATORS BURIED ZENER REFERENCES In terms of the design approaches used within the reference core the two most popular basic types of IC references consist of the bandgap and buried zener units Bandgaps have been discussed but zener based references warrant some further discussion In an IC chip surface operated diode junction breakdown is prone to crystal imperfections and other contamination thus zener diodes formed at the surface are more noisy and less stable than are buried or sub surface ones ADI zener based IC references employ the much preferred buried zener This improves substantially upon the noise and drift of surface mode operated zeners see Reference 4 Buried zener references offer very low temperature drift down to the 1
95. uC controls via the SD pin When the battery has been identified the microcontroller can do a pre qualification of the battery to make sure its voltage and temperature are within the charging range Assuming that the battery passes the SD pin is taken high and the charging process begins To program the charge voltage and charge current two digital outputs from the uC can be used in PWM mode with an RC filter on the BATpnG and pins A connection should also be made between the EOC pin of the ADP3801 and a digital input on the uC If the battery has been identified as NiCd NiMH then the must monitor the voltage and temperature to look for AV or d T dt criteria to terminate charging 5 23 BATTERY CHARGERS After this point has been reached the charge current can be set to trickle charge A timer function is needed to terminate charge if the charge time exceeds an upper limit which is usually a sign that the battery is damaged and the normal termination methods will not work The ADP3801 s final battery voltage should be programmed to a higher voltage than the maximum expected charging voltage Doing so prevents interference with the NiCd NiMH charging yet still provides a limited output voltage in case the battery is removed Meanwhile the ADP3801 maintains a tightly regulated charge current If the battery has been identified as a Li Ion battery then the ADP3801 is used to terminate charge The uC should monit
96. understanding of the EMI process is necessary to understand the effects of supply noise on analog circuits and systems Every interference problem has a source a path and a receptor Reference 1 In general there are three methods for dealing with interference First source emissions can be minimized by proper layout pulse edge rise time control reduction filtering and proper grounding Second radiation and conduction paths should be reduced through shielding and physical separation Third receptor immunity to interference can be improved via supply and signal line filtering impedance level control impedance balancing and utilizing differential techniques to reject undesired common mode signals This section focuses on reducing switching power supply noise with external post filters Tools useful for combating high frequency switcher noise are shown by Figure 8 14 They differ in electrical characteristics as well as practicality towards noise reduction and are listed roughly in an order of priorities Of these tools L and C are the most powerful filter elements and are the most cost effective as well as small sized 8 19 HARDWARE DESIGN TECHNIQUES SWITCHING REGULATOR NOISE REDUCTION TOOLS Capacitors Inductors Ferrites Resistors Linear Post Regulation ANALOG CIRCUITS Figure 8 14 Proper Layout and Grounding Techniques PHYSICAL SEPARATION FROM SENSITIVE Capacitors are probably the single most important fil
97. 0 06I N and the rms output current ripple is O 5ITN These boost converter expressions can also be expressed in terms of the output current using the relationship IoyT VOoUT VIN In any case the minimum expected value of input voltage should be used which will result in the largest value of input current In practice a safety factor of 25 should be added to the above approximations for further derating In practical applications especially those using surface mount components it may be impossible to meet the capacitance value ESR and ripple current requirement using a single capacitor Paralleling a number of equal value capacitors is a viable option which will increase the effective capacitance and reduce ESR ESL In addition the ripple current is divided between the individual capacitors 3 65 SWITCHING REGULATORS BOOST CONVERTER INPUT AND OUTPUT CAPACITOR RMS RIPPLE CURRENT APPROXIMATIONS INPUT CURRENT iiy OUTPUT CURRENT INPUT CAPACITOR RMS OUTPUT CAPACITOR RMS RIPPLE CURRENT lp p N12 RIPPLE CURRENT 0 5 lu Figure 3 64 Several electrolytic capacitor manufacturers offer low ESR surface mount devices including the AVX TPS series Reference 14 and the Sprague 595D series Reference 15 Low ESR through hole electrolytic capacitors are the HFQ series from Panasonic Reference 16 and the OS CON series from Sanyo Reference 17 SWITCHING REGULATOR OUTPUT FILTERING In order to minim
98. 0kHz to 1MHz mid band frequencies are grouped from 1MHz to 100MHz and high frequencies grouped from 100MHz to 1GHz In the case of a shielded cable input output the high frequency section should be located close to the shield to prevent high frequency leakage at the shield boundary This is commonly referred to as feed through protection For applications where shields are not required at the inputs outputs then the preferred method is to locate the high frequency filter section as close the analog circuit as possible This is to prevent the possibility of pickup from other parts of the circuit MULTISTAGE FILTERS ARE MORE EFFECTIVE Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA FEEDTHROUGH FERRITE IRON CAPACITOR PA BEAD CORE v oS 2 P 01 ZW OF Ate e e 6 TWEETER MIDRANGE WOOFER STEREO SPEAKER ANALOGY Figure 8 65 Another cause of filter failure is illustrated in Figure 8 66 If there is any impedance in the ground connection for example a long wire or narrow trace connected to the ground plane then the high frequency noise uses this impedance path to bypass the filter completely Filter grounds must be broadband and tied to low impedance points or planes for optimum performance High frequency capacitor leads should be kept as short as possible and low inductance surface mounted ceramic chip capacitors are preferable 8 69
99. 0mA The dropout voltage of a PNP will be highly dependent upon the actual device used and the operating current with vertical PNP devices being superior for saturation losses as well as minimizing the Iground spike when in saturation PMOS pass devices offer the potential for the lowest possible since the actual dropout voltage will be the product of the device Rpg ON and Ij Thus a low RDS ON PMOS device can always be chosen to minimize for a given Ig PMOS pass devices are typically external to the LDO IC making the IC actually a controller as opposed to a complete and integral LDO PMOS pass devices can allow currents up to several amps or more with very low dropout voltages The PNP NPN connection of column D is actually a hybrid hookup intended to boost the current of a single PNP pass device This it does but it also adds the of the NPN in series which cannot be saturated making the net of the connection about 1 5V All of the three connections C D E have the characteristic of high output impedance and require an output capacitor for stability The fact that the output cap is part of the regulator frequency compensation is a most basic application point and one which needs to be clearly understood by the regulator user This factor denoted by Cr sensitive makes regulators using them generally critical as to the exact CT value as well as its ESR equivalent series resistance Typically this type of
100. 1 TYPES OF VOLTAGE REFERENCES In terms of the functionality of their circuit connection standard reference ICs are often only available in series or three terminal form Common VOUT and also in positive polarity only The series types have the potential advantages of lower and more stable quiescent current standard pre trimmed output voltages and relatively high output current without accuracy loss Shunt or two terminal 1 diode like references are more flexible regarding operating polarity but they are also more restrictive as to loading They can in fact eat up excessive power with widely varying resistor fed voltage inputs Also they sometimes come in non standard voltages All of these various factors tend to govern when one functional type is preferred over the other 2 2 REFERENCES AND LOW DROPOUT LINEAR REGULATORS Some simple diode based references are shown in Figure 2 2 In the first of these a current driven forward biased diode or diode connected transistor produces a voltage Vr VREF While the junction drop is somewhat decoupled from the raw supply it has numerous deficiencies as a reference Among them are a strong TC of about 0 3 C some sensitivity to loading and a rather inflexible output voltage it is only available in 600mV jumps By contrast these most simple references as well as all other shunt type regulators have a basic advantage which is the fact that the polarity is readil
101. 1 13 0 61 0 61 0 31 0 15 14 0 31 0 31 0 15 0 08 15 0 15 0 15 0 08 0 04 16 0 08 0 08 0 04 0 02 Figure 2 13 SUPPLY RANGE IC reference supply voltages range from about 3V or less above rated output to as high as 30V or more above rated output Exceptions are devices designed for low dropout such as the REF195 and the AD1582 AD1585 series At low currents the REF 195 can deliver 5V with an input as low as 5 1V 100mV dropout Note that due to process limits some references may have more restrictive maximum voltage input ranges such as the AD1582 AD1585 series 12V or the ADR29X series 18V LOAD SENSITIVITY Load sensitivity or output impedance is usually specified in uV mA of load current mQ While figures of 100 100m or less are quite good AD780 REF 43 REF195 it should be noted that external wiring drops can produce comparable errors at high currents without care in layout Load current dependent errors are minimized with short heavy conductors on the output and on the ground return For the highest precision buffer amplifiers and Kelvin sensing circuits AD588 and AD688 are used to ensure accurate voltages at the load 2 15 REFERENCES AND LOW DROPOUT LINEAR REGULATORS The output of a buffered reference is the output of an op amp and therefore the source impedance is a function of frequency Typical reference output impedance rises at 6dB octave from the DC value and is nominally about 10Q a
102. 10 All these termination methods are generally controlled by a microcontroller After proper signal conditioning the cell 5 7 BATTERY CHARGERS voltage and temperature are converted into digital format using 8 or 10 bit A D converters which may be located inside the microcontroller itself NiCd AND NiMH FAST CHARGE TERMINATION METHODS SUMMARY NiCd NiMH B Primary B Primary dT dt Threshold dT dt Threshold Zero dV dt B Secondary TCO Absolute Temperature Cutoff Timer B Secondary TCO Absolute Temperature Cutoff Timer Figure 5 10 Li Ion cells behave quite differently from the other chemistries in that there is a gradual rise to the final cell voltage when charged from a constant current source see Figure 5 11 The ideal charging source for Li Ion is a current limited constant voltage source sometimes called constant current constant voltage or CC CV A constant current is applied to the cell until the cell voltage reaches the final battery voltage 4 2V 50mV for most Li Ion cells but a few manufacturers cells reach full charge at 4 1V At this point the charger switches from constant current to constant voltage and the charge current gradually drops The gradual drop in charge current is due to the internal cell resistance Charge is terminated when the current falls below a specified minimum value It should be noted that approximately 65 of the total charge is delive
103. 11 HARDWARE DESIGN TECHNIQUES The bottom side of the board ground plane or solder side is shown in Figure 8 9 Note that with the exception of a single crossover and a few vias the entire layer is ground plane This in conjunction with the compact layout ensures that high frequency ground currents generated by the switching action of the regulator are localized to prevent EMI RFI Evaluation boards can range from relatively simple ones linear regulators for example to rather complex ones for mixed signal ICs such as A D converters ADC evaluation boards often have on board memory and DSPs for analyzing the ADC performance Software is often provided with these more complex evaluation boards so that they can interface with a personal computer to perform complex signal analysis such as histogram and FFT testing In summary good analog designers utilize as many tools as possible to ensure that the final system design performs correctly The first step is the intelligent use of IC macromodels where available to simulate the circuit The second step is the construction of a prototype board to further verify the design and the simulation The final PCB layout should be then be based on the prototype layout as much as possible Finally evaluation boards can be extremely useful in evaluating new analog ICs and allow designers to verify the IC performance with a minimum amount of effort The layout of the components on the evaluation boa
104. 12A TO 8 100 30 AD841 Figure 8 44 STANDARD PACKAGE THERMAL RESISTANCES 2 Package ADI Designation JA C W C W Comment 14 pin plastic DIP N 14 150 AD713 14 pin ceramic DIP D 14 110 30 AD585 14 pin SOIC R 14 120 AD813 15 pin SIP Y 15 41 2 AD815 Through Hole 16 pin plastic N 16 120 40 16 pin ceramic DIP D 16 95 22 AD524 16 pin SOIC R 16 85 AD811 18 pin ceramic DIP D 18 120 35 AD7575 Figure 8 45 8 47 HARDWARE DESIGN TECHNIQUES STANDARD PACKAGE THERMAL RESISTANCES 3 Package ADI Designation JA C W ogc C W Comment 20 pin plastic DIP N 20 102 31 20 pin ceramic DIP D 20 70 10 20 pin SOIC R 20 74 24 24 pin plastic DIP N 24 105 35 24 pin ceramic DIP D 24 120 35 AD7547 28 pin plastic DIP N 28 74 24 28 pin ceramic DIP D 28 51 8 28 pin SOIC R 28 71 23 TO 220 53 3 Through Hole TO 263 D2PAK 73 3 Surface Mount Figure 8 46 is the thermal resistance of a given device as measured between its junction and the device case This form is most often used with larger power semiconductors which do dissipate significant amounts of power that is typically more than 1W The reason for this is that a heat sink generally must be used with such devices to maintain a sufficiently low internal junction temperature A heat sink is simply an additional low
105. 19 thermostatic switches and setpoint controller 6 29 32 TMP12 based airflow monitor 32 35 Sensitivity line voltage references 2 16 load voltage references 2 15 16 Sensor temperature change 6 11 15 Sheingold Dan 2 24 6 38 Shielded cable RFI feed through protection 8 69 Shielding 8 78 86 cables 8 82 85 effectiveness calculation 8 82 wave absorption reflection 8 78 gaps high frequency current antenna effect 8 85 grounding 8 83 84 interference conductive enclosures 8 78 impedance 8 78 source 8 78 magnetic fields 8 79 80 materials conductivity permeability 8 81 mechanisms absorption 8 79 80 reflection 8 79 80 openings EMI waveguide 8 81 maximum radiation 8 82 sensors 8 82 Signal processing analog and digital trends 1 4 Simpson Chester 5 25 Simulation prototyping and evaluation boards manufacturer list 8 13 Slattery B 8 77 Slattery Bill 8 87 Snubber 3 14 Socket low profile 8 8 Solder Mount advantages 8 5 components 8 5 Solomon Jim 2 57 SPICE micromodels 8 1 2 Sprague 595D series capacitors 3 66 Step down converter basic scheme 3 10 currents 3 11 discontinuous operation point 3 15 gated oscillator inductance calculation 3 49 gated oscillator control output voltage waveform 3 33 ideal 3 10 15 input output current ripple current rating 3 65 waveforms 3 59 60 input output relationship 3 12 negative in negative out 3 20 power MOSFET switches 3 39
106. 2 18 REFERENCES AND LOW DROPOUT LINEAR REGULATORS OP113 With any amplifier Kelvin sensing can be used at the load point a technique which can eliminate IxR related output voltage errors SCALED REFERENCES A useful approach when a non standard reference voltage is required is to simply buffer and scale a basic low voltage reference diode With this approach a potential difficulty is getting an amplifier to work well at such low voltages as 3V A workhorse solution is the low power reference and scaling buffer shown in Figure 2 17 Here a low current 1 2V two terminal reference diode is used for D1 which can be either a 1 235V AD589 or the 1 225V AD1580 Resistor R1 sets the diode current in either case and is chosen for 50pA at a minimum supply of 2 7V a current suitable for either diode Obviously loading on the unbuffered diode must be minimized at the VREF node RAIL TO RAIL OUTPUT OP AMPS ALLOW GREATEST FLEXIBILITY IN LOW DROPOUT REFERENCES C1 17 Vout VREF OR D1 AD589 1 235V AD1580 1 225V E R3 U1 SEE TEXT VREF O UNBUFFERED Figure 2 17 The amplifier U1 both buffers and optionally scales up the nominal 1 2V reference allowing much higher source sink output currents Of course a higher op amp quiescent current is expended in doing this but this is a basic tradeoff of the approach Quiescent current is amplifier dependent ranging from 45uA channel with the OP196 296 496 series to
107. 2 20 Where possible a reference should be designed to drive large capacitive loads The AD780 is designed to drive unlimited capacitance without oscillation it has excellent drift and an accurate output in addition to relatively low power consumption Other references which are useful with output capacitors are the REF19X and AD1582 AD1585 series As noted above reference bypass capacitors are useful when driving the reference inputs of successive approximation ADCs Figure 2 21 illustrates reference voltage settling behavior immediately following the Start Convert command A small capacitor 0 01 does not provide sufficient charge storage to keep the reference voltage stable during conversion and errors may result As shown by the bottom trace decoupling with a 21 capacitor maintains the reference stability during conversion Where voltage references are required to drive large capacitances it is also critically important to realize that their turn on time will be prolonged Experiment may be needed to determine the delay before the reference output reaches full accuracy but it will certainly be much longer than the time specified on the data sheet for the same reference in a low capacitance loaded state 2 22 REFERENCES AND LOW DROPOUT LINEAR REGULATORS SUCCESSIVE APPROXIMATION ADCs CAN PRESENT A DYNAMIC TRANSIENT LOAD TO THE REFERENCE Vin Q START CONVERT 0 22yF ff 107 fi
108. 20mA of current to power external circuitry such as a microcontroller An Under Voltage Lock Out UVLO circuit is included to safely shut down the charging circuitry when the input voltage drops below its minimum rating A shutdown pin is also provided to turn off the charger when for example the battery has been fully charged The LDO remains active during shutdown and the UVLO circuit consumes only 100 of quiescent current During charging the ADP3801 3802 maintains a constant programmable charge current The high side differential to single ended current sense amplifier has low offset allowing the use of a low voltage drop sense resistor of 100mQ The input common mode range extends from ground to 2V ensuring current control over the full charging voltage of the battery including a short circuit condition The output of the current sense amp is compared to a high impedance DC voltage input ISET lt sets the charge current is as follows VISET 10 RcS For Rcg 100m9 an input voltage of 1 0V gives a charge current of 1 0 Amp ICHARGE When the battery voltage approaches its final limit the device naturally transfers to voltage control mode The charge current then decreases gradually as was shown in Figure 5 11 The pin is used to program one of the six available battery voltages This pin controls a six channel multiplexer that selects the proper tap on a resistor divider as sh
109. 2V reference zener diode based references must of necessity be driven from voltage sources appreciably higher than 6V levels so this precludes operation of zener references from 5V system supplies References based on low TC zener avalanche diodes also tend to be noisy due to the basic noise of the breakdown mechanism This has been improved greatly with monolithic zener types as is described further below At this point we know that a reference circuit can be functionally arranged into either a series or shunt operated form and the technology within may use either bandgap based or zener diode based circuitry In practice there are all permutations of these available as well as a third major technology category The three major reference technologies are now described in more detail BANDGAP REFERENCES The development of low voltage lt 5V references based on the bandgap voltage of silicon led to the introductions of various ICs which could be operated on low voltage supplies with good TC performance The first of these was the LM109 Reference 1 and a basic bandgap reference cell is shown in Figure 2 3 BASIC BANDGAP REFERENCE Vs Figure 2 3 2 4 REFERENCES AND LOW DROPOUT LINEAR REGULATORS This circuit is also called a AVpp reference because the differing current densities between matched transistors Q1 Q2 produces a AVpg across R3 It works by summing of Q3 with the amplified AVpgp of Q1 Q2 develope
110. 3 41 pulse burst modulation inductance calculation 3 49 output voltage waveform 3 33 pulse wave modulation constant frequency inductance calculation 3 52 53 variable frequency inductance calculation 3 53 54 switch and diode voltage effects 3 36 switch duty cycle 3 12 switch duty ratio 3 12 synchronous switch P and N channel MOSFETs 3 44 waveforms 3 10 11 discontinuous mode 3 13 14 Step up converter basic circuit 3 16 discontinuous operation point 3 19 20 gated oscillator inductance calculation 3 50 51 ideal 3 15 20 input output current ripple current rating 3 65 66 waveforms 3 59 60 input output relationship 3 17 negative in negative out 3 20 power MOSFET switches 3 39 3 41 pulse burst modulation inductance calculation 3 50 51 pulse wave modulation constant frequency inductance calculation 3 54 55 waveforms 3 16 discontinuous mode 3 18 Supply range voltage references 2 15 Swager Anne Watson 5 25 Switch modulation techniques 3 26 28 Switched capacitor voltage converter 4 1 21 advantages 4 2 3 CMOS or bipolar switches 4 8 9 diagram 4 2 efficiency 4 2 power losses 4 11 13 regulated output 4 15 21 steady state 4 7 voltage doubler 4 1 3 INDEX voltage inverter 4 1 3 Switched capacitor voltage inverter unregulated 4 8 9 Switched capacitor voltage regulator boost 4 18 Switches voltage converters MOSFET or bipolar 4 9 Switching regulator 3 1 71 advantages 3 2 3 app
111. 51 3 50 SWITCHING REGULATORS We make the same assumptions about the inductor current but note that the output current shown on the diagram is pulsating and not continuous The output current IOUT be expressed in terms of the peak current and the duty cycle D as I Solving for IppgAK yields 2I IPEAK or However IpgAK can also be expressed in terms of VIN Vgw L and ton IPEAK m New Which can be solved for L ta IPEAK Substituting the previous expression for IPRAK yields L ew y D ton L for boost PBM Converter 2IOUT The minimum expected value of VIN should be used in order to ensure sufficient inductor energy storage under all conditions If VIN is expected to vary widely an external resistor can be added to the ADP3000 to limit peak current and prevent inductor saturation at maximum V N The above equations will only yield approximations to the proper inductor value for the PBM type regulators and should be used only as a starting point An exact analysis is difficult and highly dependent on the regulator and input output conditions However there is considerable latitude with this type of regulator and other analyses may yield different results but still fall within the allowable range for proper regulator operation Calculating the proper inductor value for PWM regulators is more straightforward Figure 3 52 shows the output and inductor current wav
112. 56 SERIES ESR 0 60 Figure 8 34 8 36 HARDWARE DESIGN TECHNIQUES ADP1148 BUCK FILTERED OUTPUT 1A HE 3mV ADP1148 BUCK REG CIRCUIT C BED M5 00us Chi 7 200pV C1 220 pF C2 100pF 20V 1 WF VERTICAL SCALE 10 DIV HORIZ SCALE 5ps DIV C1 1yF CERAMIC 220uF 25V GENERAL PURPOSE AL ELECTROLYTIC C2 100pF 20V LEADED TANTALUM KEMET T356 SERIES ESR 0 60 OUTPUT FILTER L COILTRONICS CTX 50 4 Cp 100 20 LEADED TANTALUM T356 SERIES Figure 8 35 ADP1148 9V TO 3 75V BUCK REGULATOR FOLLOWED BY ADP3310 3 3V LINEAR LOW DROPOUT POST REGULATOR Linear regulators are often used following switching regulators for better regulation and lower noise Low dropout LDO regulators such as the ADP3310 are desirable in these applications because they require only a small input to output series voltage to maintain regulation This minimizes power dissipation in the pass device and may eliminate the need for a heat sink Figure 8 36 shows the ADP1148 buck regulator configured for a 9V input and a 3 75V 1A output The output drives an ADP3310 linear LDO regulator configured for 3 75V input and 3 3V 1A output The input and output of the ADP3310 is shown in Figure 8 37 Notice that the regulator reduces the ripple from 25mV to approximately 5mV 8 37 HARDWARE DESIGN TECHNIQUES ADP1148 BUCK REGULATOR DRIVING ADP3310 LOW DROPOUT REGULATOR m 9V
113. 6 26 29 MP04 6 26 29 MP12 6 32 35 MP17 6 21 22 MP35 6 8 9 6 11 6 23 MP36 6 23 MP37 6 23 T T T T T T T T Index 16
114. 6 Nonlinearity Programmable Front End Binary Gains from 1 to 128 Differential Input Capability 3 Wire Serial Interface Ability to Buffer Analog Input 3V or 5V Single Supply Operation Low Power 450pA 3V Programmable Low Pass Digital Filter with Programmable Output Rate 20Hz to 500Hz 16 pin DIP SOIC and TSSOP Figure 7 16 7 12 HARDWARE MONITORING AD7705 BATTERY MONITORING APPLICATION TO LOAD 3V OR 5V AIN1 CIRCUITS MUR AIN1 AD7705 AIN2 AIN2 Figure 7 17 Complex hardware monitoring circuits often interface with a microcontroller such as the 8051 which performs various operations based on the sensor and monitor outputs The ADu810PC is a MicroConverter combination ADC and microcontroller based on the standard 8051 core In addition to the microcontroller core the device has a 10 bit 2us ADC with SHA and a 16 channel analog input multiplexer The chip also contains a temperature sensor and bandgap voltage reference as well as two 8 bit DACs with voltage output buffers MicroConverters such as these allow sophisticated monitoring and control functions such as power supply monitoring and watchdog timeout to be performed in a single chip 7 13 HARDWARE MONITORING 7 14 ADuC810PC MicroConverter Complete Hardware Monitor System with on chip microcontroller Standard 8051 based Core Calibrated 10 bit 2 microsecond ADC with SHA and DMA Mode 16 Channel A
115. 7 4 ADMS86060 4 13 15 ADM8691 7 11 7 13 ADM8693 7 2 3 ADM9240 7 9 11 ADM9261 7 4 6 ADM9264 7 6 8 INDEX ADM9268 7 8 ADP330X 2 38 ADP1073 3 34 ADP1108 3 34 ADP1109 3 34 ADP1110 3 34 ADP1111 3 34 ADP1147 3 26 3 31 3 41 44 3 53 ADP1148 3 44 46 3 53 8 10 11 8 16 8 26 8 34 38 ADP1173 3 34 ADP3000 3 28 3 32 3 34 38 3 49 8 6 7 8 10 8 26 34 ADP3050 3 39 40 ADP3153 3 46 47 ADP3300 2 41 44 2 46 8 10 8 47 ADP3301 2 42 ADP3302 2 42 ADP3308 2 42 ADP3307 2 42 ADP3310 2 48 51 2 53 2 55 56 8 14 15 8 26 8 37 38 8 52 ADP3367 8 47 ADP3608 4 2 4 12 4 16 18 ADP3604 4 2 4 12 4 16 18 ADP3605 4 2 4 12 4 16 18 8 26 8 39 40 ADP3607 4 2 4 12 4 18 21 ADP3801 5 18 24 ADP3801 3802 5 18 24 ADP3810 5 12 ADP3810 3811 5 10 16 ADP3811 5 12 ADP3820 5 17 18 ADR29X 2 15 ADR290 2 11 12 ADR291 2 11 ADR292 2 11 2 18 ADR293 2 11 12 ADTO5 6 29 30 8 47 ADT14 6 32 ADT20 21 22 6 32 ADT41 8 47 ADT45 6 24 25 ADT50 6 24 25 ADT70 6 14 15 O OPO07 8 47 OP27 2 18 OP113 2 18 19 OP176 2 18 OP181 281 481 2 20 OP184 284 484 2 20 OP191 291 491 2 20 Index 15 INDEX OP192 293 493 2 20 198 6 8 9 OP196 296 496 2 19 20 OP279 2 19 20 OP284 2 19 OP295 495 2 20 R REF01 2 5 2 18 REF02 2 5 2 18 REFO5 2 18 REF10 2 18 REFA3 2 14 16 REF19X 2 14 2 16 2 22 REF 195 2 5 2 14 16 T MP01 6 30 31 MP03
116. 8 OP193 single supply op amp 6 8 OP196 296 496 scaled references 2 19 OP279 scaled reference 2 19 OP284 scaled reference 2 19 Op amp low voltage rail rail references specifications 2 20 output as buffered reference 2 16 rail to rail output low dropout references 2 19 Opto isolator 5 10 Optocoupler 5 15 OS CON capacitor 3 63 3 66 8 10 electrolytic 8 20 22 low ESR 2 37 Ott Henry 8 44 8 77 Ott Henry W 8 87 Ott H W 8 86 Output impedance voltage references 2 15 P Pallas Areny Ramon 6 38 Parasitics capacitors 4 3 4 8 23 Pass device tradeoffs 2 29 33 in voltage regulators advantages disadvantages 2 30 Darlington NPN 2 30 design safety margin 2 51 dropout voltage and ground current 2 34 FET drive voltages 2 51 lateral PNP dropout voltage 2 35 PMOS 2 30 disadvantages 2 34 35 PNP NPN 2 30 selection 2 50 51 single NPN 2 30 single PNP 2 30 thermal design 2 51 53 vertical PNP dropout voltage 2 35 Passive component non ideal EMI behavior 8 65 PCB design EMI protection 8 72 76 circuit function partition 8 74 embedding 8 75 line termination 8 76 multilayer 8 74 striplines 8 74 track impedance 8 76 high frequency noise 8 72 73 multilayer EMI protection 8 74 Pease Robert A 8 13 8 87 Permeability 3 56 Personal computers semiconductor sensors 1 7 Pin socket 8 8 9 advantages 8 9 Polyester capacitor stacked film 3 63 64 INDEX Portable equipmen
117. 9 Brownout 7 1 Bryant James 2 1 8 1 8 2 8 86 Buck converter basic scheme 3 10 currents 3 11 discontinuous operation point 3 15 gated oscillator inductance calculation 3 49 gated oscillator control output voltage waveform 3 33 ideal 3 10 15 input output current ripple current rating 3 65 waveforms 3 59 60 input output relationship 3 12 negative in negative out 3 20 power MOSFET switches 3 39 3 41 pulse burst modulation inductance calculation 3 49 output voltage waveform 3 33 pulse wave modulation constant frequency inductance calculation 3 52 53 variable frequency inductance calculation 3 53 54 pulse width modulation variable frequency constant off time 3 26 27 switch and diode voltage effects 3 36 switch duty cycle 3 12 switch duty ratio 3 12 synchronous switch P and N channel MOSFETs 3 44 waveforms 3 10 11 discontinuous mode 3 13 14 Buck boost converter schemes 3 21 22 topologies 3 21 23 Buffer amplifier 2 16 Buried zener reference characteristics 2 13 stability 2 10 Buxton Joe 5 1 5 25 Bypass capacitor 2 21 C Cable shielding electrically long short 8 82 84 grounding 8 83 84 low frequency interference 8 83 84 pigtail connections 8 84 85 Cage jack 8 8 9 advantages 8 9 Capacitor charge redistribution 4 6 charging 3 8 charging from voltage source 4 5 classes 3 62 63 8 20 23 aluminum electrolytic 3 63 8 20 22 ceramic 3 63 8 20 22 polyester 3 63 8
118. ADM8660 54A Shutdown Current 500us Shutdown Recovery Time 8 Pin SOIC Figure 4 15 4 14 SWITCHED CAPACITOR VOLTAGE CONVERTERS ADM660 ADM8660 TYPICAL EFFICIENCY C1 C2 2 2uF 120kHz 10yF 25kHz 100 EFFICIENCY 80 60 3 8 OUTPUT 40 4 2 VOLTAGE Volts 20 4 6 0 5 100 LOAD CURRENT mA Figure 4 16 REGULATED OUTPUT SWITCHED CAPACITOR VOLTAGE CONVERTERS Adding regulation to the simple switched capacitor voltage converter greatly enhances its usefulness in many applications There are three general techniques for adding regulation to a switched capacitor converter The most straightforward is to follow the switched capacitor inverter doubler with a low dropout LDO linear regulator The LDO provides the regulated output and also reduces the ripple of the switched capacitor converter This approach however adds complexity and reduces the available output voltage by the dropout voltage of the LDO Another approach to regulation is to vary the duty cycle of the switch control signal with the output of an error amplifier which compares the output voltage with a reference This technique is similar to that used in inductor based switching regulators and requires the addition of a PWM and appropriate control circuitry However this approach is highly nonlinear and requires long time constants i e lossy components in order to maintain good regulation control 4 15 SWITCHED CAPACITOR VOLTAGE CONVERT
119. AL ANSI JUNCTION MATERIALS USEFUL SENSITIVITY DESIGNATION RANGE C Platinum 6 Rhodium 38 to 1800 7 7 B Platinum 30 Rhodium Tungsten 5 Rhenium 0 to 2300 16 Tungsten 26 Rhenium Chromel Constantan 0 to 982 76 E Iron Constantan 0 to 760 55 J Chromel Alumel 184 to 1260 39 K Platinum 13 Rhodium 0 to 1593 11 7 R Platinum Platinum 10 Rhodium 0 to 1538 10 4 S Platinum Copper Constantan 184 to 400 45 T Figure 6 3 Figure 6 4 shows the voltage temperature curves of three commonly used thermocouples referred to a 0 C fixed temperature reference junction Of the thermocouples shown Type J thermocouples are the most sensitive producing the largest output voltage for a given temperature change On the other hand Type S thermocouples are the least sensitive These characteristics are very important to consider when designing signal conditioning circuitry in that the thermocouples relatively low output signals require low noise low drift high gain amplifiers To understand thermocouple behavior it is necessary to consider the non linearities in their response to temperature differences Figure 6 4 shows the relationships between sensing junction temperature and voltage output for a number of thermocouple types in all cases the reference cold junction is maintained at 0 C It is evident that the responses are not quite linear but the nature of the non linearity is not so obvious Figure 6 5 s
120. AR REGULATORS A functional diagram of the ADP3310 regulator controller is shown in Figure 2 42 The basic error amplifier reference and scaling divider of this circuit are similar to the standalone anyCAP regulator and will not be described in detail The regulator controller version does share the same cap load immunity of the standalone versions and also has a shutdown function similarly controlled by the EN enable pin The main differences in the regulator controller IC architecture is the buffered output of the amplifier which is brought out on the GATE pin to drive the external PMOS FET In addition the current limit sense amplifier has a built in 50mV threshold voltage and is designed to compare the voltage between the VrN and IS pins When this voltage exceeds 50mV the current limit sense amplifier takes over control of the loop by shutting down the error amplifier and limiting output current to the preset level FUNCTIONAL BLOCK DIAGRAM OF anyCAP SERIES LDO REGULATOR CONTROLLER Figure 2 42 A Basic 5V 1A LDO Regulator Controller An LDO regulator controller is easy to use since a PMOS FET a resistor and two relatively small capacitors one at the input one at the output is all that is needed to form an LDO regulator The general configuration is shown by Figure 2 43 an LDO suitable as a regulator operating from a of 6V using the ADP3310 5 controller IC This regulator is stable with virtua
121. ATTERY CHARGERS APPLICATION OF OFF LINE CHARGER IN LAPTOP COMPUTERS AC DC MAY INCLUDE CHARGER BRICK OUTSIDE BRICK INSIDE Figure 5 19 LINEAR BATTERY CHARGER In some applications where efficiency and heat generation is not a prime concern a low cost linear battery charger can be an ideal solution The ADP3820 linear regulator controller is designed to accurately charge single cell Li Ion batteries as shown in Figure 5 20 Its output directly controls the gate of an external p channel MOSFET As the circuit shows a linear implementation of a battery charger is a simple approach In addition to the IC and the MOSFET only an external sense resistor and input and output capacitors are required The charge current is set by choosing the appropriate value of sense resistor Rg The ADP3820 includes all the components needed to guarantee a system level specification of 1 final battery voltage and it is available with either a 4 2V or 4 1V final battery voltage The ADP3820 has an internal precision reference low offset amplifier and trimmed thin film resistor divider to guarantee Li Ion accuracy In addition an enable EN pin is available to place the part in low current shutdown If a linear charger is needed for higher Li Ion battery voltages such as 8 4V 12 6V or 16 8V the ADP3810 with an external MOSFET can also be used Refer to the ADP3810 data sheet for more details 5 17 BATTERY CHARGERS The tradeof
122. BOARD LOW DROPOUT UNREGULATED SWITCHING VOLTAGE eee DC VOLTAGE REGULATORS REFERENCE ANALOG SIGNAL 16 BIT A D INPUTS CONDITIONING CONVERTER DATA AND CONTROL TEMPERATURE MONITORING CONTROL HARDWARE MONITORING Figure 1 4 TRENDS IN DIGITAL AND ANALOG SIGNAL PROCESSING Faster Digital and Analog Signal Processing Higher Power Requires Thermal Management E Distributed Power Systems vs Single Power Supply Implies On Board Regulation Energy Efficient Requires Switching Regulators and Low Dropout Linear Regulators B 16 Bit ADCs Require Precision Voltage References Figure 1 5 Portable electronic equipment such as laptop computers and cell phones require other types of hardware monitoring as well as power and thermal management circuits Today s laptop computers are replacing the traditional desktop systems in many companies see Figure 1 6 Laptops however present a large number of design challenges because of the emphasis on performance light weight low power and long battery life Battery charging circuits are quite complex and battery voltage and temperature must be monitored and controlled during the charging 1 4 INTRODUCTION cycle Redundancy must be built into these circuits in order to prevent damage to the battery or dangerous outgassing Thermal and power management is therefore critical to laptop computers not only relating to the high power micropr
123. C package yields 160 C W Given such data as these derating curves the 0JA for a given device can be readily determined as above MAXIMUM POWER DISSIPATION VS TEMPERATURE FOR STANDARD AND THERMAL COASTLINE 8 PIN SOICs 2 0 8 PIN THERMAL COASTLINE SOIC 1 5 1 0 8 PIN STANDARD SOIC 0 5 5 MAXIMUM POWER DISSIPATION W 50 40 30 20 10 0 10 20 30 40 50 60 70 80 90 AMBIENT TEMPERATURE C Figure 8 47 A physical comparison of the standard 8 pin SOIC leadframe and the Analog Devices thermal coastline leadframe is shown in Figure 8 48 and 8 49 Note that the geometry of the thermal coastline leadframe increases the amount of heat transferred to the pins by decreasing the face to face distance between the leadframe and the paddle as well as increasing the width of the adjoining faces 8 49 HARDWARE DESIGN TECHNIQUES THERMAL COASTLINE PACKAGE 1 8 1 8 2 7 2 7 3 6 3 6 4 5 4 5 STANDARD LEADFRAME SOIC THERMAL COASTLINE SOIC Figure 8 48 DETAILS OF THERMAL COASTLINE PACKAGE STANDARD FRAME THERMAL COASTLINE FRAME Face to face distance from lead to paddle Era d reduced by a factor of 1 5 to 2 Paddle Width of adjoining Center of T faces increased by Package factor of 2to 2 5 __ Center of Package Figure 8 49 8 50 HARDWARE DESIGN TECHNIQUES Heat Sink and Airflow Considerations The fundamental purpose of heat sinks and airflo
124. C to 105 C Typical linearity error is 0 5 C The AD592 is available in a TO 92 package and the TMP17 in an SO 8 package 6 21 TEMPERATURE SENSORS CURRENT OUTPUT SENSORS AD592 TMP17 O V AD592 TO 92 PACKAGE TMP17 SO 8 PACKAGE V 1pA K Scale Factor Nominal Output Current 25 C 298 2 Operation from 4V to 30V 0 5 C Max Error 25 C 1 0 C Error Over Temp 0 1 C Typical Nonlinearity AD592CN 2 5 C Max Error 25 C 3 5 C Error Over Temp 0 5 C Typical Nonlinearity TMP17F 0592 Specified from 25 C to 105 C B TMP17 Specified from 40 C to 105 C Figure 6 24 RATIOMETRIC VOLTAGE OUTPUT SENSORS R T AD22103 AG lit us fem f utp thy Saat f a Vl yf tat nl at Pas a ca ail a et hm lv el e E Figure 6 25 6 22 TEMPERATURE SENSORS In some cases it is desirable for the output of a temperature sensor to be ratiometric with its supply voltage The AD22103 see Figure 6 25 has an output that is ratiometric with its supply voltage nominally 3 3V according to the equation VOUT 28 x 025v x The circuit shown in Figure 6 25 uses the AD22103 power supply as the reference to the ADC thereby eliminating the need for a precision voltage reference The AD22103 is specified over a range of 0 C to 100 C and has an accuracy better than 2 5 C and a linearity better than 0 5 The TMP35 TMP36 TMP37 are low voltage 2 7V to 5 5V SOT 23 5 pin SO 8
125. CAN MAKE AN LDO APPLICATIONS NIGHTMARE 100 UNSTABLE 10 CAPACITOR Eon STABLE 1 UNSTABLE 0 1 0 lour mA 1000 Figure 2 31 A zoned ESR chart such as this is meant to guide the user of an LDO in picking an output capacitor which confines ESR to the stable region i e the central zone for all operating conditions Note that this generic chart is not intended to portray any specific device just the general pattern Unfortunately capacitor facts of life make such data somewhat limited in terms of the real help it provides Bearing in mind the requirements of such a zoned chart it effectively means that general purpose aluminum electrolytic are prohibited from use since they deteriorate in terms of ESR at cold temperatures Very low ESR types such as OS CON or multi layer ceramic units have ESRs which are too low for use While they could in theory be padded up into the stable zone with external resistance this would hardly be a practical solution This leaves tantalum types as the best all around choice for LDO output use Finally since a large capacitor value is likely to be used to maximize stability this effectively means that the solution for an LDO such as Fig 2 29 must use a more expensive and physically large tantalum capacitor This is not desirable if small size is a major design criteria 2 37 REFERENCES AND LOW DROPOUT LINEAR REGULATORS THE anyCAP Low DROPOUT REGULATOR FAMILY Some novel modifications to the ba
126. E CONVERTERS Walt Kester Brian Erisman Gurjit Thandi INTRODUCTION In the previous section we saw how inductors can be used to transfer energy and perform voltage conversions This section examines switched capacitor voltage converters which accomplish energy transfer and voltage conversion using capacitors The two most common switched capacitor voltage converters are the voltage inverter and the voltage doubler circuit shown in Figure 4 1 In the voltage inverter the charge pump capacitor C1 is charged to the input voltage during the first half of the switching cycle During the second half of the switching cycle its voltage is inverted and applied to capacitor C2 and the load The output voltage is the negative of the input voltage and the average input current is approximately equal to the output current The switching frequency impacts the size of the external capacitors required and higher switching frequencies allow the use of smaller capacitors The duty cycle defined as the ratio of charging time for C1 to the entire switching cycle time is usually 50 because that generally yields the optimal charge transfer efficiency After initial start up transient conditions and when a steady state condition is reached the charge pump capacitor only has to supply a small amount of charge to the output capacitor on each switching cycle The amount of charge transferred depends upon the load current and the switching frequency During
127. E standard C62 41 Military Equipment The defining EMC specification for military equipment is MIL STD 461 which applies to radiated equipment emissions and equipment susceptibility to interference Radiated emission limits are very typically 10 to 100 times more stringent than the levels shown in Figure 8 58 Required limits on immunity to RF fields are typically 200 times more stringent RF field strengths of 5 50mV m than the limits for commercial equipment Medical Equipment Although not yet mandatory as of December 1997 EMC regulations for medical equipment are presently being defined by the FDA Food and Drug Administration in the USA and the European Community The primary focus of these EMC regulations will be on immunity to RF fields electrostatic discharge and power line disturbances and may very well be more stringent than the limits spelled out in MIL STD 461 The primary objective of the medical EMC regulations is to guarantee safety to humans Industrial and Process Control Equipment Presently equipment designed and marketed for industrial and process control applications are not required to meet pre existing mandatory EMC regulations In fact manufacturers are exempt from complying to any standard in the USA However since industrial environments are very much electrically hostile all equipment manufacturers are required to comply with all European Community EMC regulations as of 1996 8 60 HARDWARE DESIG
128. E DESIGN TECHNIQUES RFI CAN CAUSE RECTIFICATION IN SENSITIVE ANALOG CIRCUITS Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA Ed IA M INPUTS PICK UP HIGH FREQUENCY ENERGY ON SIGNAL LINE WHICH IS DETECTED BY THE AMPLIFIER NY Vcc xe dir a OUTPUT DRIVERS CAN BE JAMMED TOO ENERGY COUPLES BACK TO INPUT VIA Vcc OR SIGNAL LINE AND THEN IS DETECTED OR AMPLIFIED Figure 8 62 There are techniques that can be used to protect analog circuits against interference from RF fields see Figure 8 63 The three general points of RFI coupling are signal inputs signal outputs and power supplies At a minimum all power supply pin connections on analog and digital ICs should be decoupled with 0 1 ceramic capacitors As was shown in Reference 3 low pass filters whose cutoff frequencies are set no higher than 10 to 100 times the signal bandwidth can be used at the inputs and the outputs of signal conditioning circuitry to filter noise Care must be taken to ensure that the low pass filters LPFs are effective at the highest RF interference frequency expected As illustrated in Figure 8 64 real low pass filters may exhibit leakage at high frequencies Their inductors can lose their effectiveness due to parasitic capacitance and capacitors can lose their effectiveness due to parasitic inductance A rule of thumb is that a conventional
129. EGULATOR IC E Input Voltage Range 0 8V to 30V E Output Voltage Standard 12V 5V 3 3V 3V Specialized VID Programmable for Microprocessors Some are Adjustable Output Current Up to 1 5A Using Internal Switches of a Regulator No Inherent Limitations Using External Switches with a Controller Output Line Load Regulation 50mV typical Output Voltage Ripple peak peak 20mV 100mV Switching Frequency B Switching Frequency 20kHz 1MHz B Efficiency Up to 95 Figure 3 3 POPULAR APPLICATIONS OF SWITCHING REGULATORS For equipment which is powered by an AC source the conversion from AC to DC is generally accomplished with a switcher except for low power applications where size and efficiency concerns are outweighed by cost Then the power conversion may be done with just an AC transformer some diodes a capacitor and a linear regulator The size issue quickly brings switchers back into the picture as the preferable conversion method as power levels rise up to 10 watts and beyond Off line power conversion is heavily dominated by switchers in most modern electronic equipment Many modern high power off line power supply systems use the distributed approach by employing a switcher to generate an intermediate DC voltage which is then distributed to any number of DC DC converters which can be located near to their respective loads see Figure 3 4 Although there is the obvious redundancy of converting th
130. EGULATORS current decreases with a slope equal to VoyyTq L Note that the inductor current is equal to the output current in a buck converter The diode and switch currents are shown in Figures 3 10C and 3 10D respectively and the inductor current is the sum of these waveforms Also note by inspection that the instantaneous input current equals the switch current Note however that the average input current is less than the average output current In a practical regulator both the switch and the diode have voltage drops across them during their conduction which creates internal power dissipation and a loss of efficiency but these voltages will be neglected for now It is also assumed that the output capacitor C is large enough so that the output voltage does not change significantly during the switch on or off times BASIC STEP DOWN BUCK CONVERTER WAVEFORMS m ViN lin sw IL zlour Vp A ton loft ton y lur 0 Lower Case Instantaneous Value Upper Case Average Value iin 0 Figure 3 10 There are several important things to note about these waveforms The most important is that ideal components have been assumed i e the input voltage source has zero impedance the switch has zero on resistance and zero turn on and turn off times It is also assumed that the inductor does not saturate and that the diode is ideal with no forward drop Also note that the output current is continuo
131. EMC Regulations Impact on Design In all these applications and many more complying with mandatory EMC regulations will require careful design of individual circuits modules and systems using established techniques for cable shielding signal and power line filtering against both small and large scale disturbances and sound multi layer PCB layouts The key to success is to incorporate sound EMC principles early in the design phase to avoid time consuming and expensive redesign efforts A DIAGNOSTIC FRAMEWORK FOR EMI RFI PROBLEM SOLVING With any problem a strategy should be developed before any effort is expended trying to solve it This approach is similar to the scientific method initial circuit misbehavior is noted theories are postulated experiments designed to test the theories are conducted and results are again noted This process continues until all theories have been tested and expected results achieved and recorded With respect to EMI a problem solving framework has been developed As shown in Figure 8 59 the model suggested by Kimmel Gerke in Reference 1 illustrates that all three elements a source a receptor or victim and a path between the two must exist in order to be considered an EMI problem The sources of electromagnetic interference can take on many forms and the ever increasing number of portable instrumentation and personal communications computation equipment only adds the number of possible sources and recept
132. ERALIZED BATTERY CHARGING CIRCUIT CHARGING CURRENT CONTROL CURRENT SENSE BATTERY ABEIENE TEMP VOLTAGE CIRCUITS AND BATTERY TEMP Figure 5 6 This type of circuit represents a high level of sophistication and is primarily used in fast charging applications where the charge time is less than 3 hours Voltage and sometimes temperature monitoring is required to accurately determine the state of the battery and the end of charge Slow charging charge time greater than 12 hours requires much less sophistication and can be accomplished using a simple current source Typical characteristics for slow charging are shown in Figure 5 7 Charge termination is not critical but a timer is sometimes used to end the slow charging of NiMH batteries If no charge termination is indicated in the table then it is safe to trickle charge the battery at the slow charging current for indefinite periods of time Trickle charge is the charging current a cell can accept continually without affecting its service life A safe trickle charge current for NiMH batteries is typically 0 03C For example for an NiMH battery with C 1A hr 30mA would be safe Battery manufacturers can recommend safe trickle charge current limits for specific battery types and sizes 5 5 BATTERY CHARGERS BATTERY CHARGING CHARACTERISTICS FOR SLOW CHARGING SLA NiCd NiMH Li lon Current 0 25C 0 1C 0 1C 0 1C Voltage
133. ERS By far the simplest and most effective method for achieving regulation in a switched capacitor voltage converter is to use an error amplifier to control the on resistance of one of the switches as shown in Figure 4 17 a block diagram of the ADP3603 3604 3605 voltage inverters These devices offer a regulated 3V output for an input voltage of 4 5V to 6V The output is sensed and fed back into the device via the VSENSE pin Output regulation is accomplished by varying the on resistance of one of the MOSFET switches as shown by control signal labeled RON CONTROL in the diagram This signal accomplishes the switching of the MOSFET as well as controlling the on resistance Key features of the ADP3603 3604 3605 series are shown in Figure 4 18 Note that the output regulation of the ADP3605 is 2 and the switching frequency is 250kHz All three devices have a shutdown feature and a turn on turn off time of about 5ms A typical application circuit for the ADP3603 3604 3605 series is shown in Figure 4 19 In the normal mode of operation the SHUTDOWN pin should be connected to ground The 10 capacitors should have ESRs of less than 150mQ and values of 4 can be used at the expense of slightly higher output ripple voltage The equations for ripple voltage shown in Figure 4 10 also apply to the ADP3603 3604 3605 Using the values shown typical ripple voltage ranges from 25mV to 60mV as the output current varies over its allowable
134. ES S TO 220 OJA 73 C W 3 C W 7 S 0 4 10 16mm OJA 53 C W 3 C W Figure 8 52 We will first select a suitable heat sink for the TO 220 package Heat sink manufacturers such as AAVID Thermal Technologies have a variety of heat sinks suitable for a wide range of power dissipation levels Selection tables provide nominal power dissipation thermal resistance and physical size for each heat sink available for a given package style 8 53 HARDWARE DESIGN TECHNIQUES A heat sink suitable for the TO 220 package is shown in Figure 8 53 It is a finned heat sink manufactured by AAVID Thermal Technologies AAVID part number 582002B12500 The width of the heat sink is approximately 1 9 The entire assembly is bolted to the PC board and the through hole pins of the FET are then soldered to pads on the PC board The sink to ambient thermal resistance of this heat sink as a function of airflow is shown in Figure 8 54 Notice that even with no airflow the thermal resistance is approximately 5 C W which is much less than the calculated maximum allowable value of 11 7 C W This heat sink will therefore provide more than adequate design margin under the specified operating conditions AAVID 582002B12500 HEAT SINK FOR TO 220 Courtesy AAVID Thermal Technologies Inc Figure 8 53 8 54 HARDWARE DESIGN TECHNIQUES THERMAL RESISTANCE VS AIRFLOW FOR AAVID 582002B12500 HEAT SINK Courtesy AAVID
135. ESIGN TECHNIQUES ADP3605 INPUT AND OUTPUT WAVEFORMS INPUT OUTPUT 100mVp p 120mVp p BM oma Ms 00us Chi 7 OV mE oim Ms 00us Chi VERTICAL SCALE 100mV DIV VERTICAL SCALE 100mV DIV HORIZ SCALE 5ps DIV HORIZ SCALE 5ps DIV Figure 8 39 ADP3605 FILTERED OUTPUT Vin Lp 3V 4 5V TO 6V 100H 100mA 5mV p p TO OmV AA W3 00gs Chi 7 200pV Lp 10H COILTRONICS CTX10 3 VERTICAL SCALE 10mV DIV C1 C2 Cp 10pF 16V T491C SERIES HORIZ SCALE 5ps DIV Figure 8 40 SUMMARY OF RESULTS OF EXPERIMENTS The preceding experiments serve to illustrate the large number of tradeoffs which can be made when filtering switching regulator outputs The success of any combination is highly dependent upon a compact layout and the use of a large area 8 40 HARDWARE DESIGN TECHNIQUES ground plane As has been stated earlier all connections to the ground plane should be made as short as possible to minimize parasitic resistance and inductance Output ripple can be reduced by the addition of low ESL ESR capacitors to the output However it may be more efficient to use an LC filter to accomplish the ripple reduction In any case proper component selection is critical The inductor should not saturate under the maximum load current and its DC resistance should be low enough as not to induce significant voltage dr
136. Eout ADM8691 ADM8693 ADM800L ADM800M 0 15 pP VOLINE POWER uP LOW LINE WDO Y SYSTEM Y STATUS INDICATORS Figure 7 3 Several other actions occur when falls below its threshold value The battery backup VpBATT is connected to the CMOS RAM power supply input via the VOUT pin Under normal operation Vcc is connected to VOUT and the CMOS RAM receives its power from the Vcc input of the chip The switch resistance from Voc to VoUT is 0 80 and 120 from VpgATT to VoyT ON goes high when VOUT is internally switched to the input It goes low when Voy is internally switched to Vcc The BAT ON output may also be used to drive the base via a resistor of an external PNP transistor to increase the output current above the 250mA rating of VOUT The Chip Enable output CEQUT goes low only when CEIN is low and Vcc is above the reset threshold If CEIN is low when reset is asserted CEQUT will remain low for 15 or until CEIN goes high whichever occurs first The watchdog input WDI is a three level input If WDI remains either high or low for longer than the watchdog timeout period RESET pulses low and WDO goes low The internal timer resets with each transition on the WDI line The Watchdog Timer is disabled when WDI is left floating or driven to midsupply With OSC SEL high or floating the internal oscillator is enabled and sets the reset delay and the watchdog timeout period Connec
137. F C2 440 BE jo mvs soaps Chi Fav 1 uF 220uF 10V x 2 e 220 2 VERTICAL SCALE 100mV DIV HORIZ SCALE 5ps DIV C1 1pF CERAMIC 220pF 25V GENERAL PURPOSE AL ELECTROLYTIC C2 220 uF 10V OSCON x 2 Figure 8 32 In order to evaluate the effects of an output filter the ripple was increased by replacing the two OS CON output capacitors with a single 100 leaded tantalum The resulting output ripple was increased to 40mV and is shown in Figure 3 34 Now the effects of a filter could be evaluated It consisted of a 50nH inductor followed by a 100 leaded tantalum Ripple was reduced from 40mV to 3mV as shown in Figure 3 35 8 35 HARDWARE DESIGN TECHNIQUES ADP1148 BUCK OUTPUT WAVEFORM CONDITION 1 Von ee 3 3V bs 6mV 1A ADP1148 BUCK REG CIRCUIT C1 220 pF C2 440yF 1 pF 220uF 10V x 2 BM i 0mv 5 0005 Chi 3 8mv VERTICAL SCALE 10mV DIV HORIZ SCALE 5ys DIV C1 1yuF CERAMIC 220uF 25V GENERAL PURPOSE AL ELECTROLYTIC C2 220 10 OSCON x 2 Figure 8 33 ADP1148 BUCK OUTPUT CONDITION 2 Vour Vin 3 3V 1A 40mV p p ADP1148 REG ee e e e a a a CIRCUIT 100 e nid 0071207 BEND 10 espa Chi Fram VERTICAL SCALE 10mV DIV HORIZ SCALE 5ps DIV CERAMIC 220uF 25V GENERAL PURPOSE AL ELECTROLYTIC C2 100pF 20V LEADED TANTALUM KEMET T3
138. Fax 49 8152 40525 Ralph Morrison Grounding and Shielding Techniques in Instrumentation Third Edition John Wiley Inc 1986 Henry W Ott Noise Reduction Techniques in Electronic Systems Second Edition John Wiley Inc 1988 Robert A Pease Troubleshooting Analog Circuits Butterworth Heinemann 1991 8 87 HARDWARE DESIGN TECHNIQUES 13 14 15 16 17 18 19 20 21 22 23 8 88 Jim Williams Editor Analog Circuit Design Art Science and Personalities Butterworth Heinemann 1991 Doug Grant and Scott Wurcer Avoiding Passive Component Pitfalls The Best of Analog Dialogue pp 143 148 Analog Devices Inc 1991 Walt Jung and Richard Marsh Picking Capacitors Part I Audio February 1980 Walt Jung and Richard Marsh Picking Capacitors Part II Audio March 1980 Daryl Gerke and Bill Kimmel The Designer s Guide to Electromagnetic Compatibility EDN Supplement January 20 1994 Walt Kester Basic Characteristics Distinguish Sampling A D Converters EDN September 3 1992 pp 135 144 Walt Kester Peripheral Circuits Can Make or Break Sampling ADC System EDN October 1 1992 pp 97 105 Walt Kester Layout Grounding and Filtering Complete Sampling ADC System EDN October 15 1992 pp 127 134 Howard W Johnson and Martin Graham High Speed Digital Design PTR Prentice Hall 1993 High Speed Design Techniques Analog Devices 1996 Walt Kester A Gr
139. GE EOC DETECTION IN THE ADP3801 ADP3802 EOC t SD CHARGING SHUTDOWN t SD FROM SYSTEM LOGIC I lt 30min gt 5 24 5 21 BATTERY CHARGERS The internal EOC comparator actually monitors the voltage across CS and CS Vcg When drops to 8mV the EOC comparator trips Thus the actual current level for detecting the end of charge can be adjusted by changing the value of Res This may be useful when more than one cell is charged in parallel For example two parallel cells may use an end of charge current of 160mA so Rog should be 0 05Q This results in a total charging current of 2A 1A cell for VISET It should be noted however that changing the value of Rcg in order to change IMIN also requires a change VyggT in order to maintain the same charging current To prevent false triggering of EOC during start up the internal comparator is gated by a second comparator that monitors the battery voltage The EOC comparator is only enabled when VpAT is at least 95 of its final value Because of the soft start the charge current is initially zero when the power is applied If the EOC comparator was not gated by the battery voltage it would initially signal the EOC until the charge current rose above 80mA which could cause incorrect battery charging Typically system operation is to continue charging for 30 minutes after the EOC signal and then shutdown the charger using t
140. HARDWARE DESIGN TECHNIQUES NON ZERO INDUCTIVE AND OR RESISTIVE FILTER GROUND REDUCES EFFECTIVENESS Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA FILTER See HF HF ENERGY enn ENERGY BOND IMPEDANCE Figure 8 66 SOLUTIONS FOR POWER LINE DISTURBANCES The goal of this next section is not to describe in detail all the circuit system failure mechanisms which can result from power line disturbances or faults Nor is it the intent of this section to describe methods by which power line disturbances can be prevented Instead this section will describe techniques that allow circuits and systems to accommodate transient power line disturbances Figure 8 67 is an example of a hybrid power transient protection network commonly used in many applications where lightning transients or other power line disturbances are prevalent These networks can be designed to provide protection against transients as high as 10kV and as fast as 10ns Gas discharge tubes crowbars and large geometry zener diodes clamps are used to provide both differential and common mode protection Metal oxide varistors MOVs can be substituted for the zener diodes in less critical or in more compact designs Chokes are used to limit the surge current until the gas discharge tubes fire Commercial EMI filters as illustrated in Figure 8 68 can be used to
141. Hz switching frequency 10 is recommended and for 120kHz operation 2 2 provides comparable performance If frequencies less than the selected output frequency are desired an external capacitor can be placed between the OSC input and ground The internal oscillator can also be overridden by driving the OSC input with an external logic signal in which case the internal charge pump frequency is one half the external clock frequency The ADM8660 is similar to the ADM660 however it is optimized for inverter operation and includes a shutdown feature which reduces the quiescent current to 5uA Shutdown recovery time is 500us Key specifications for the ADM660 ADM8660 series are given in Figure 4 15 Efficiency for the ADM660 ADM8660 is greater than 90 for output currents up to 50mA and greater than 80 for output currents to 100mA see Figure 4 16 4 13 SWITCHED CAPACITOR VOLTAGE CONVERTERS ADM660 SWITCHED CAPACITOR VOLTAGE CONVERTER INVERTER DOUBLER 2Vin Vin 1 5 TO 7V FC CAP ADM660 LV CAP GND OUT 2 5 TO 7V Figure 4 14 ADM660 ADM8660 KEY SPECIFICATIONS ADM660 Inverts or Doubles Input Supply Voltage ADM8660 Inverts Input Supply Voltage Input Range Inverting 1 5V to 7V Input Range Doubling 2 5 to 7V ADM660 100mA Output Current Selectable Switching Frequency 120kHz or 25kHz 2 2uF or 10pF External Capacitors 120kHz 25kHz 600pA Quiescent Current Shutdown Function
142. IC OPERATION The steady state current and voltage waveforms for a switched capacitor voltage inverter are shown in Figure 4 10 The average value of the input current waveform must be equal to When the pump capacitor is connected to the input charging current flows The initial value of this charging current depends on the initial voltage across C1 the ESR of C1 and the resistance of the switches The switching frequency switch resistance and the capacitor ESRs generally limit the peak amplitude of the charging current to less than 2 5IOUT The charging current then decays exponentially as C1 is charged The waveforms in Figure 4 10 assume that the time constant due to capacitor C1 the switch resistance and the ESR of C1 is several times greater than the switching period 1 f Smaller time constants will cause the peak currents to increase as well as increase the slopes of the charge discharge waveforms Long time constants cause longer start up times and require larger and more costly capacitors For the conditions shown in Figure 4 10 A the peak value of the input current is only slightly greater than 210177 The output current waveform of C1 is shown in Figure 4 10 B When C1 is connected to the output capacitor the step change in the output capacitor current is approximately 21017 This current step therefore creates an output voltage step equal to 2I9yrT x ESR C2 as shown in Figure 4 10 C After the step change
143. In order to understand power management better we will consider a few typical applications Consider the traditional desktop PC power supply shown in Figure 1 2 This approach suffers from a number of disadvantages including inefficiency all the voltages are on all the time which is probably not necessary multiple high current distribution busses etc CLASSICAL POWER SUPPLY SYSTEM e g TRADITIONAL DESKTOP PCs 110 220 Vac 50 60 Hz gt 12 15V STANDARD 12 15V POWER SUPPLY 45V 5V Major Disadvantages of the Traditional PS System Include Inefficiency Output Voltages Are Always Turned Cable Lengths Inductance 1 2 Figure 1 2 INTRODUCTION The trend in today s systems is to make use of the distributed power approach as shown in Figure 1 3 The AC input is rectified filtered and converted into an unregulated intermediate voltage which is distributed throughout the system Each subsystem uses localized voltage regulators usually switching types for high efficiency for generating required voltages This simplifies the power distribution problem and also allows individual voltages to be shutdown if they are not in use DISTRIBUTED POWER SUPPLY SYSTEM Regulator 24V Intermediate Voltage e g 5V 12V or 48V PHBIDO 110 220 Vac Regulator 12V 50 60 Hz AC DC POWER SUPPLY SHUTDOWN Regulator 5V Distributed Supply Advantages SHUTDOWN Regulation of
144. Intermediate Voltage not Critical Regulator 3 3V Lower Current in Intermediate Voltage Distribution Bus Flexibility Selective Shutdown SHUTDOWN Figure 1 3 To see how this concept is extended to the PC board level design Figure 1 4 shows a simplified block diagram of a data acquisition board The unregulated intermediate voltage enters the board and drives the switching regulators In the example shown one switching regulator is dedicated to the processor and the other drives a low dropout linear regulator The critical analog circuits on the board including the signal conditioning and A D converter are supplied from the output of the linear regulator This ensures that the analog circuits operate with a well regulated and low noise supply voltage A separate low noise voltage reference is used in conjunction with the 16 bit A D converter for even lower noise and higher accuracy The hardware monitoring circuits monitor the processor power supply voltages to ensure the processor functions properly Airflow and heat sinking is often required with modern high speed DSPs or microprocessors because of their high power dissipation Therefore a temperature sensor monitors the processor temperature and works in conjunction with the temperature monitoring and control circuit to regulate the airflow Figure 1 5 summarizes some of the trends in digital and analog signal processing 1 3 INTRODUCTION SIMPLIFIED DATA ACQUISITION
145. LATING L FOR BUCK CONVERTER GATED OSCILLATOR PBM TYPE OUTPUT AND INDUCTOR CURRENT VOUT Vp EE IPEAK Vin VoUT Vsw Vin VoUT V5w IPEAK L Jton bs 9 Vout Vsw 2louT USE MINIMUM Vin Figure 3 50 3 49 SWITCHING REGULATORS The peak current is easily calculated from the slope of the positive going portion of the ramp VIN VOUT VSW IPEAK This equation then be solved for L s um Vout VSW IPEAK However the average output current IOUT is equal to 2 and therefore IpERAK 2IQUT Substituting this value for into the previous equation yields L UN VOUT VSW Yos L for buck PBM Converter 2IOUT The minimum expected value of should be used in order to minimize the inductor value and maximize its stored energy If VIN is expected to vary widely an external resistor can be added to the ADP3000 to limit peak current and prevent inductor saturation at maximum V N A similar analysis can be carried out for a boost PBM regulator as shown in Figure 3 51 CALCULATING L FOR BOOST CONVERTER GATED OSCILLATOR PBM TYPE OUTPUT CURRENT MN Ysw E ee VoUT VN VD N n IPEAK ton 2loUT OUT 1_D PEAK 2 1 D IpEAK VIN VSW L on IPEAK n Ysw wee L VSW jion VIN VSW IPEAK 2loUT 17 D ton USE MINIMUM Viy Figure 3
146. LDED CABLE CAN CARRY HIGH FREQUENCY CURRENT AND BEHAVES AS AN ANTENNA Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA EQUIVALENT CIRCUIT j 2 0 Icom Icm COMMON MODE CURRENT Figure 8 79 8 85 HARDWARE DESIGN TECHNIQUES REFERENCES CABLE SHIELDING 10 11 8 86 H W Ott Noise Reduction Techniques in Electronic Systems Second Edition John Wiley amp Sons Inc New York 1988 Ralph Morrison Grounding and Shielding Techniques in Instrumentation Third Edition John Wiley amp Sons Inc New York 1988 Systems Application Guide Section 1 Analog Devices Inc Norwood MA 1993 AD620 Instrumentation Amplifier Data Sheet Analog Devices Inc A Rich Understanding Interference Type Noise Analog Dialogue 16 3 1982 pp 16 19 A Rich Shielding and Guarding Analog Dialogue 17 1 1983 pp 8 13 EDN s Designer s Guide to Electromagnetic Compatibility EDN January 20 1994 material reprinted by permission of Cahners Publishing Company 1995 Designing for EMC Workshop Notes Kimmel Gerke Associates Ltd 1994 James Bryant and Herman Gelbach High Frequency Signal Contamination Analog Dialogue Vol 27 2 1993 Walt Jung System RF Interference Prevention Analog Dialogue Vol 28 2 1994 Neil Muncy Noise Susceptibility in Analog and Digital Signal Pr
147. N TECHNIQUES Automotive Equipment Perhaps the most difficult and hostile environment in which electrical circuits and systems must operate is that found in the automobile All of the key EMI threats to electrical systems exist here In addition operating temperature extremes moisture dirt and toxic chemicals further exacerbate the problem To complicate matters further standard techniques ferrite beads feed through capacitors inductors resistors shielded cables wires and connectors used in other systems are not generally used in automotive applications because of the cost of the additional components Presently automotive EMC regulations defined by the very comprehensive SAE Standards J551 and J1113 are not yet mandatory They are however very rigorous SAE standard J551 applies to vehicle level EMC specifications and standard J1113 functionally similar to MIL STD 461 applies to all automotive electronic modules For example the J1113 specification requires that electronic modules cannot radiate electric fields greater than 300nV m at a distance of 3 meters This is roughly 1000 times more stringent than the FCC Part 15 Class A specification In many applications automotive manufacturers are imposing J1113 RF field immunity limits on each of the active components used in these modules Thus in the very near future automotive manufacturers will require that IC products comply with existing EMC standards and regulations
148. NCY f ATTENUATION 2741 ESR Example ESR 0 20 L 10uH f 100kHz ATTENUATION 32 Figure 3 65 SWITCHING REGULATOR INPUT FILTERING The input ripple current in a buck converter is pulsating while that of a boost converter is a sawtooth Additional filtering may be required to prevent the switching frequency and the other higher frequency components from affecting the main supply ripple current This is easily accomplished by the addition of a small inductor in series with the main input capacitor of the regulator as shown in Figure 3 66 The reactance of the inductor at the switching frequency forms a divider with the ESR of the input capacitor The inductor will block both low and high frequency components from the main input voltage source The attenuation of the ripple current at the switching frequency f is approximately 21fL ESR 3 67 SWITCHING REGULATORS SWITCHING REGULATOR INPUT FILTERING ATTENUATION 3 68 SWITCHING REGULATOR SWITCHING FREQUENCY f 1 MAIN INPUT CAPACITOR 2nfL ESR Figure 3 66 SWITCHING REGULATORS REFERENCES 10 11 12 13 Irving M Gottlieb Power Supplies Switching Regulators Inverters and Converters Second Edition McGraw Hill TAB Books 1994 Marty Brown Practical Switching Power Supply Design Academic Press 1990 Marty Brown Power Supply Cookbook Butterworth Heinemann 1994 John D Lenk Simplified Design of
149. NT TANTALUM Figure 8 24 The inductor selected Lp 12 5nH was the same value and type used as the energy transfer inductor in the regulator circuit thereby ensuring the inductor has adequate current carrying capability The capacitor Cp 47 was a surface mount tantalum Peak to peak ripple was reduced from 32mV to 14 mV consisting mostly of high frequency spikes In order to reduce the high frequency spikes a second capacitor Cpo 10 was added in parallel with the 47pF This output filter combination reduced the ripple to approximately 3mV as shown in Figure 8 25 The ESL of the 10 capacitor was approximately 2 2nH Kemet T491C series From this experiment we concluded that the 47 surface mount tantalum filter capacitor had an ESL ESR combination which was not sufficiently low to remove the high frequency ripple components The capacitor was removed and replaced with a 33yF tantalum Sprague 293D series In addition the regulator output capacitor C2 was reduced to a single tantalum also Sprague 293D series The resulting output waveform is shown in Figure 8 26 Note that the high frequency components have been removed but a small amount of ripple remains at the frequency of the gated oscillator bursts approximately 40kHz 8 30 HARDWARE DESIGN TECHNIQUES ADP3000 BOOST FILTERED OUTPUT CONDITION 2 3mV p p C1 100pF C2 100 T0 0mV v amp EN 5 00us Chi 7 3 8mV S3pF 16
150. ODUCTION HARDWARE DESIGN TECHNIQUES Verifying the Design Simulation Prototyping B Minimizing Noise Layout Grounding Decoupling and Filtering Shielding B Thermal Management Temperature Sensing Airflow Control Heat Sinks B EMI RFI Qualification Figure 1 10 1 8 REFERENCES AND LOW DROPOUT LINEAR REGULATORS SECTION 2 REFERENCES AND LOW DROPOUT LINEAR REGULATORS Walt Jung Reference circuits and linear regulators actually have much in common In fact the latter could be functionally described as a reference circuit but with greater current or power output Accordingly almost all of the specifications of the two circuit types have great commonality even though the performance of references is usually tighter with regard to drift accuracy etc This chapter is broadly divided into an initial discussion on voltage references followed by a concluding discussion on linear regulators with emphasis on their low dropout operation for highest power efficiency PRECISION VOLTAGE REFERENCES Walt Jung Walt Kester James Bryant Voltage references have a major impact on the performance and accuracy of analog systems A 5mV tolerance on a 5V reference corresponds to 0 1 absolute accuracy only 10 bits For a 12 bit system choosing a reference that has a 1mV tolerance may be far more cost effective than performing manual calibration while both high initial accuracy and calibration will be
151. PERATURE SENSORS HARDWARE MONITORING HARDWARE DESIGN TECHNIQUES INDEX INTRODUCTION SECTION 1 INTRODUCTION Walt Kester This book focuses on three rather broad and inter related topics power management thermal management and hardware monitoring We will discuss them in terms of the various design and application issues associated with each and show how modern ICs allow cost effective and efficient solutions Power management broadly refers to the generation and control of regulated voltages required to operate an electronic system It encompasses much more than just power supply design Today s systems require that power supply design be integrated with the system design in order to maintain high efficiency In addition distributed power supply systems require localized regulators at the PC board level thereby requiring the design engineer to master at least the basics of switching and linear regulators Integrated circuit components such as switching regulators linear regulators switched capacitor voltage converters and voltage references are typical elements of power management Battery charging is also an important portion of power management Closely related to power management is thermal management In addition to traditional applications of temperature sensors in industrial process control today s systems require accurate control of monitoring and control of temperature airflow etc Today s computers
152. PRACTICAL DESIGN TECHNIQUES FOR POWER AND THERMAL MANAGEMENT INTRODUCTION REFERENCES AND LOW DROPOUT LINEAR REGULATORS SWITCHING REGULATORS SWITCHED CAPACITOR VOLTAGE CONVERTERS BATTERY CHARGERS TEMPERATURE SENSORS HARDWARE MONITORING HARDWARE DESIGN TECHNIQUES INDEX ANALOG DEVICES TECHNICAL REFERENCE BOOKS PUBLISHED BY PRENTICE HALL Analog Digital Conversion Handbook Digital Signal Processing Applications Using the ADSP 2100 Family Volume 1 1992 Volume 2 1994 Digital Signal Processing in VLSI DSP Laboratory Experiments Using the ADSP 2101 ADSP 2100 Family User s Manual PUBLISHED BY ANALOG DEVICES High Speed Design Techniques Practical Analog Design Techniques Linear Design Seminar ADSP 21000 Family Applications Handbook System Applications Guide Applications Reference Manual Amplifier Applications Guide Mixed Signal Design Seminar Notes High Speed Design Seminar Notes Nonlinear Circuits Handbook Transducer Interfacing Handbook Synchro amp Resolver Conversion THE BEST OF Analog Dialogue 1967 1991 HOW TO GET INFORMATION FROM ANALOG DEVICES Analog Devices publishes data sheets and a host of other technical literature supporting our products and technologies Follow the instructions below for worldwide access to this information FOR DATA SHEETS U S A and Canada Fax Retrieval Telephone number 800 446 6212 Call this number and use a faxcode corresponding to the data sheet of your choic
153. R KEY SPECIFICATIONS Input Voltages from 2V to 12V Step Up 2V to 30V Step Down Fixed 3 3V 5V 12V and Adjustable Output Voltage Step Up or Step Down Mode PBM Gated Oscillator Control Simplifies Design 50mV Typical Output Ripple Voltage 5V Output 400kHz Switching Frequency Allows Low Value Inductors 80 Duty Cycle 500pA Quiescent Current Output Drive Capability 100mA from 5V Input in Step Down Mode 180mA 3 3V from 2V Input in Step Up Mode 8 Pin DIP or SOIC Package Figure 3 34 The device uses the gated oscillator or pulse burst modulation PBM feedback control scheme The internal oscillator operates at a frequency of 400kHz allowing the use of small value inductors and capacitors The internal resistors R1 and R2 set the output voltage to 3 3V 5V or 12V depending upon the option selected A completely adjustable version is also available where the comparator input is brought out directly to the SENSE pin and the user provides the external divider resistors Total quiescent current is only 500pA The uncommitted gain block A1 can be used as a low battery detector or to reduce output hysteretic ripple limits by adding gain in the feedback loop A current limit pin I JM allows switch current to be limited with an external resistor Limiting the switch current on a cycle by cycle basis allows the use of small inductors with low saturation current It also allows physically small tantalum capacitors with
154. RS THERMOCOUPLE RTD THERMISTOR SEMICONDUCTOR Widest Range Range Range Range 184 to 2300 C 200 C to 850 0 C to 100 C 55 to 150 C High Accuracy and Fair Linearity Poor Linearity Linearity 1 Repeatability Accuracy 1 C Needs Cold Junction Requires Requires Requires Excitation Compensation Excitation Excitation Low Voltage Output Low Cost High Sensitivity 10mV K 20mV K or 1 Typical Output Figure 6 2 THERMOCOUPLE PRINCIPLES AND COLD JUNCTION COMPENSATION Thermocouples are small rugged relatively inexpensive and operate over the widest range of all temperature sensors They are especially useful for making measurements at extremely high temperatures up to 2300 C in hostile environments They produce only millivolts of output however and require precision amplification for further processing They also require cold junction compensation CJC techniques which will be discussed shortly They are more linear than many other sensors and their non linearity has been well characterized Some common thermocouples are shown in Figure 6 3 The most common metals used are Iron Platinum Rhodium Rhenium Tungsten Copper Alumel composed 6 2 TEMPERATURE SENSORS of Nickel and Aluminum Chromel composed of Nickel and Chromium and Constantan composed of Copper and Nickel COMMON THERMOCOUPLES TYPICAL NOMIN
155. START CONVERT SCOPE TOP TRACE VERTICAL SCALE 5V div ALL OTHER VERTICAL SCALES 5mV div HORIZONTAL SCALE 1ys div Figure 2 21 Low NOISE REFERENCES FOR HiGH RESOLUTION CONVERTERS High resolution converters both sigma delta and high speed ones can benefit from recent improvements in IC references such as lower noise and the ability to drive capacitive loads Even though many data converters have internal references the performance of these references is often compromised because of the limitations of the converter process In such cases using an external reference rather than the internal one often yields better overall performance For example the AD7710 series of 22 bit ADCs has a 2 5V internal reference with a 0 1 to 10Hz noise of 8 3nV rms 2600nV VHz while the AD780 reference noise is only 0 67 rms 200nV VHz The internal noise of the AD7710 series in this bandwidth is about 1 7uV rms The use of the AD780 increases the effective resolution of the AD7710 from about 20 5 bits to 21 5 bits Figure 2 22 shows the AD780 used as the reference for the AD7710 series ADCs The use of the AD780 s optional 3V scaling enhances the dynamic range of the ADC while lowering overall system noise as described above In addition the AD780 allows a large decoupling capacitor on its output thereby minimizing conversion errors due to transients There is one possible but yet quite real problem when replacing the internal reference of a conver
156. Secaucus NJ 07094 201 348 7000 OS CON Aluminum Electrolytic Capacitor 93 94 Technical Book Sanyo 3333 Sanyo Road Forrest City AK 72335 501 633 6634 Ian Clelland Metalized Polyester Film Capacitor Fills High Frequency Switcher Needs PCIM June 1992 Type 5MC Metallized Polycarbonate Capacitor Electronic Concepts Inc Box 1278 Eatontown NJ 07724 908 542 7880 Walt Jung Regulators for High Performance Audio Parts 1 and 2 The Audio Amateur issues 1 and 2 1995 Henry Ott Noise Reduction Techniques in Electronic Systems 2d Ed 1988 Wiley Fair Rite Linear Ferrites Catalog Fair Rite Products Box J Wallkill NY 12886 914 895 2055 Type EXCEL leaded ferrite bead EMI filter and type EXC L leadless ferrite bead Panasonic 2 Panasonic Way Secaucus NJ 07094 201 348 7000 Steve Hageman Use Ferrite Bead Models to Analyze EMI Suppression The Design Center Source MicroSim Newsletter January 1995 Type 5250 and 6000 101K chokes J W Miller 306 E Alondra Blvd Gardena CA 90247 310 515 1720 DIGI KEY PO Box 677 Thief River Falls MN 56701 0677 800 344 4539 Tantalum Electrolytic Capacitor SPICE Models Kemet Electronics Box 5928 Greenville SC 29606 803 963 6300 Eichhoff Electronics Inc 205 Hallene Road Warwick RI 02886 401 738 1440 HARDWARE DESIGN TECHNIQUES THERMAL MANAGEMENT Walt Jung Walt Kester For reliability reasons modern semiconductor
157. TY B er AIR CORE INDUCTOR CURRENT MAGNETIC FIELD STRENGTH H Figure 3 55 The addition of a ferromagnetic core increases the slope of the curve and increases the effective inductance but at some current level the inductor core will saturate i e the inductance is drastically reduced It is obvious that inductor saturation can wreck havoc in a switching regulator and can even burn out the switch if it is not current limited This effect can be reduced somewhat while still maintaining higher inductance than an air core by the addition of an air gap in the ferromagnetic core The air gap reduces the slope of the curve but provides a wider linear operating range of inductor current Air gaps do have their problems however and one of them is the tendency of the air gapped inductor to radiate high frequency energy more than a 3 56 SWITCHING REGULATORS non gapped inductor Proper design and manufacturing techniques however can be used to minimize this EMI problem so air gapped cores are popular in many applications The effects of inductor core saturation in a switcher can be disastrous to the switching elements as well as lowering efficiency and increasing noise Figure 3 56 shows a normal inductor current waveform in a switching regulator as well as a superimposed waveform showing the effects of core saturation Under normal conditions the slope is linear for both the charge and discharge cycle If saturation occur
158. The output of the sigma delta modulator is encoded using a proprietary technique which results in a serial digital output signal with a mark space ratio format see Figure 6 30 that is easily decoded by any microprocessor into either degrees centigrade or degrees Fahrenheit and readily transmitted over a single wire Most importantly this encoding method avoids major error sources common to other modulation techniques as it is clock independent The nominal output frequency is 35Hz at 25 C and the device operates with a fixed high level pulse width T1 of 10ms 6 26 TEMPERATURE SENSORS DIGITAL OUTPUT SENSORS TMP03 04 REFERENCE VOLTAGE SIGMA DELTA OUTPUT 4 OUTPUT 4 JZ AS Lo GND V Figure 6 29 TMP03 TMP04 OUTPUT FORMAT lt T1 lt 12 5 400 x T2 TEMPERATURE C 235 4 TEMPERATURE F 455 7 m T2 T1 Nominal Pulse Width z 10ms 1 5 C Error Over Temp 0 5 C Non Linearity Typical Specified 40 C to 100 C Nominal T1 T2 0 C 60 Nominal Frequency 25 C 35Hz 6 5mW Power Consumption 5V TO 92 SO 8 or TSSOP Packages Figure 6 30 6 27 TEMPERATURE SENSORS The TMP03 TMP04 output is a stream of digital pulses and the temperature information is contained in the mark space rat
159. This sensitivity is brought about by the fundamental requirement for an output capacitor for the IC s frequency compensation which is a differentiation from the original LM317 Low DROPOUT REGULATOR ARCHITECTURES As has been shown thus far all LDO pass devices have the fundamental characteristics of operating in an inverting mode This allows the regulator circuit to 2 33 REFERENCES AND LOW DROPOUT LINEAR REGULATORS achieve pass device saturation and thus low dropout A by product of this mode of operation is that this type of topology will necessarily be more susceptible to stability issues These basic points give rise to some of the more difficult issues with regard to LDO performance In fact these points influence both the design and the application of LDOs to a very large degree and in the end determine how they are differentiated in the performance arena A traditional LDO architecture is shown in Figure 2 29 and is generally representative of actual parts employing either a PNP pass device as shown or alternately a PMOS device There are both DC and AC design and application issues to be resolved with this architecture which are now discussed TRADITIONAL LDO ARCHITECTURE Vout PNP OR PMOS PASS DEVICE Figure 2 29 In DC terms perhaps the major issue is the type of pass device used which influences dropout voltage and ground current If a lateral PNP device is used for Q1 the will be low sometimes
160. To illustrate this point let us assume that a 100Q platinum RTD with 30 gauge copper leads is located about 100 feet from a controller s display console The resistance of 30 gauge copper wire is 0 105Q ft and the two leads of the RTD will contribute a total 210 to the network which is shown in Figure 6 14 This additional resistance will produce a 55 C error in the measurement The leads temperature coefficient can contribute an additional and possibly significant error to the measurement To eliminate the effect of the lead resistance a 4 wire technique is used 1000 Pt RTD WITH 100 FEET OF 30 GAUGE LEAD WIRES R 10 50 1000 Pt RTD R 10 50 COPPER RESISTANCE TC OF COPPER 0 40 20 C RESISTANCE TC OF Pt RTD 0 385 20 C Figure 6 14 In Figure 6 15 a 4 wire or Kelvin connection is made to the RTD A constant current is applied though the FORCE leads of the RTD and the voltage across the RTD itself is measured remotely via the SENSE leads The measuring device can be a DVM or an instrumentation amplifier and high accuracy can be achieved provided that the measuring device exhibits high input impedance and or low input bias current Since the SENSE leads do not carry appreciable current this technique is insensitive to lead wire length Sources of errors are the stability of the constant current source and the input impedance and or bias currents in the amplifier or DVM RTDs are
161. URE CAN ACT AS AN EMI WAVEGUIDE BY COMPROMISING SHIELDING EFFECTIVENESS Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA VENTILATORS Figure 8 76 8 81 HARDWARE DESIGN TECHNIQUES The longest dimension not the total area of an opening is used to evaluate the ability of external fields to enter the enclosure because the openings behave as slot antennas Equation 8 10 can be used to calculate the shielding effectiveness or the susceptibility to EMI leakage or penetration of an opening in an enclosure Shielding Effectiveness dB 201og10 zx Eq 8 10 where A wavelength of the interference and L 2 maximum dimension of the opening Maximum radiation of EMI through an opening occurs when the longest dimension of the opening is equal to one half wavelength of the interference frequency 0dB shielding effectiveness A rule of thumb is to keep the longest dimension less than 1 20 wavelength of the interference signal as this provides 20dB shielding effectiveness Furthermore a few small openings on each side of an enclosure is preferred over many openings on one side This is because the openings on different sides radiate energy in different directions and as a result shielding effectiveness is not compromised If openings and seams cannot be avoided then conductive gaskets screens and paints alone or in combination should be used judiciously to limit the l
162. UTRALS FOR DIFFERENTIAL MODE Figure 8 69 PRINTED CIRCUIT BOARD DESIGN FOR EMI PROTECTION This section will summarize general points regarding the most critical portion of the design phase the printed circuit board layout It is at this stage where the performance of the system is most often compromised This is not only true for signal path performance but also for the system s susceptibility to electromagnetic interference and the amount of electromagnetic energy radiated by the system Failure to implement sound PCB layout techniques will very likely lead to system instrument EMC failures Figure 8 70 is a real world printed circuit board layout which shows all the paths through which high frequency noise can couple radiate into out of the circuit Although the diagram shows digital circuitry the same points are applicable to precision analog high speed analog or mixed analog digital circuits Identifying critical circuits and paths helps in designing the PCB layout for both low emissions and susceptibility to radiated and conducted external and internal noise sources 8 72 HARDWARE DESIGN TECHNIQUES METHODS BY WHICH HIGH FREQUENCY ENERGY COUPLE AND RADIATE INTO CIRCUITRY VIA PLACEMENT Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA COUPLING TO I O VIA COUPLING VIA COMMON CROSSTALK OR RADIATION POWER IMPEDANCE 108
163. V 90 oko JOR 95 3kQ Ince REF ni R2 R3 Figure 6 38 6 35 TEMPERATURE SENSORS ADCs With On Chip Temperature Sensors The AD7416 7417 7418 series digital temperature sensors have on board temperature sensors whose outputs are digitized by a 10 bit ADC The output interface is IC compatible for convenience The device family offers a variety of input options for further flexibility The AD7816 7817 7818 are similar but have standard serial interfaces AD7416 DIGITAL TEMPERATURE SENSOR WITH I2C COMPATIBLE INTERFACE 2 7V 5 5V 10 BIT CHARGE DATA OUTPUT TEMP REDISTRIBUTION AND SENSOR SAR ADC 2 INTERFACE GND A0 A1 A2 Figure 6 39 6 36 TEMPERATURE SENSORS AD7416 7417 7418 SERIES TEMP SENSOR 10 BIT ADCs WITH 2 INTERFACE 10 Bit ADC with 20us Conversion Time I C Compatible Interface On Chip Temperature Sensor 55 C to 125 C On Chip Voltage Reference 2 5V 0 1 2 7V to 5 5V Power Supply 4uW Power Dissipation at 10Hz Sampling Rate Auto Power Down after Conversion Over Temp Interrupt Output Four Single Ended Analog Input Channels AD7417 One Single Ended Analog Input Channel AD7418 AD7816 7817 7818 Similar but have Serial Interface Figure 6 40 6 37 TEMPERATURE SENSORS REFERENCES 1 10 6 38 Ramon Pallas Areny and John G Webster Sensors and Signal Conditioning John Wiley New York 1991 Dan Sheingold Editor
164. V x 3 S3pF 16V x 3 VERTICAL SCALE 10mV DIV HORIZ SCALE 5ps DIV OUTPUT FILTER Lp 12 5pH COILTRONICS CTX25 4 Cr 7 10 SURFACE MOUNT TANTALUM Cro 100F 16V SURFACE MOUNT TANTALUM KEMET T491C SERIES Figure 8 25 ADP3000 BOOST FILTERED OUTPUT CONDITION 3 Lr V 12 5uH 100mA p p Cro 10 10 0mVv amp M5 00us Chi 7 3 8mV C1 100 C2 33pF 16V S3uF 16V 3 VERTICAL SCALE 10mV DIV HORIZ SCALE 5us DIV CHANGED C2 TO SINGLE 33yF 16V SURFACE MOUNT TANTALUM OUTPUT FILTER CHANGED Cg Lr 12 5pH COILTRONICS CTX25 4 Cr 33yF 16V SURFACE MOUNT TANTALUM SPRAGUE 293D SERIES Cro 100F 16V SURFACE MOUNT TANTALUM T491C SERIES Figure 8 26 8 31 HARDWARE DESIGN TECHNIQUES ADP3000 9V To 5V 100MA BUCK REGULATOR The circuit for the ADP3000 9V to 5V 100mA buck regulator is shown in Figure 8 27 The input waveform is shown in Figure 8 28 Note that the pulsating waveform followed by an increasing ramp voltage is characteristic of the gated oscillator buck input The corresponding output waveform is shown in Figure 8 29 The output filter chosen see Figure 8 30 consisted of a 12 5yH inductor followed by a capacitor in parallel with 10pF capacitor identical to the filter used in the circuit shown in Figure 8 26 The ripple was reduced from 30mV to approximately 6mV peak to peak ADP3000 9V TO 5V BUCK APPLICATION
165. Versions Available Choice Depends Upon Source and Frequency of Interference Impedance Required at Interference Frequency Environmental Temperature AC DC Field Strength Size Space Available Always Test the Design Figure 8 18 HARDWARE DESIGN TECHNIQUES Several ferrite manufacturers offer a wide selection of ferrite materials from which to choose as well as a variety of packaging styles for the finished network see References 10 and 11 A simple form is the bead of ferrite material a cylinder of the ferrite which is simply slipped over the power supply lead to the decoupled stage Alternately the leaded ferrite bead is the same bead pre mounted on a length of wire and used as a component see Reference 11 More complex beads offer multiple holes through the cylinder for increased decoupling plus other variations Surface mount beads are also available PSpice ferrite models for Fair Rite materials are available and allow ferrite impedance to be estimated see Reference 12 These models have been designed to match measured impedances rather than theoretical impedances A ferrite s impedance is dependent upon a number of inter dependent variables and is difficult to quantify analytically thus selecting the proper ferrite is not straightforward However knowing the following system characteristics will make selection easier First determine the frequency range of the noise to be filtered Second the
166. a junction case thermal resistance of 2 C W The required external heatsink s thermal resistance is determined as follows OCA 9JC gt 2 52 REFERENCES AND LOW DROPOUT LINEAR REGULATORS where is the required heat sink case to ambient thermal resistance is the calculated overall junction to ambient thermal resistance and 0j is the pass device junction to case thermal resistance which in this case is 2 C W typical for TO 220 devices and NDP6020P OCA 29 4 C W 2 C W 27 4 C W For a safety margin select a heatsink with a less than the results of this calculation For example the Aavid TO 220 style clip on heat sink 576802 has a of 18 8 C W and in fact many others have performance of 25 C W or less As an alternative the NDB6020P D PAK FET pass device could be used in this same design with an SMD style heat sink such as the Aavid 573300 series used in conjunction with an internal PCB heat spreader Note that many LDO applications like the above will calculate out with very modest heat sink requirements This is fine as long as the output never gets shorted With a shorted output the current goes to the limit level as much as 1 5A in this case while the voltage across the pass device goes to which could also be at a maximum In this case the new pass device dissipation for short circuit conditions becomes 1 5A x 6 7V or 10W Supporting this level of
167. a range of input output conditions it is generally more costly in non isolated systems to accommodate a requirement for both voltage step up and step down So generally it is preferable to limit the input output ranges such that one or the other case can exist but not both and then a simpler converter design can be chosen The concerns of minimizing power dissipation and noise as well as the design complexity and power converter versatility set forth the limitations and challenges for designing switchers whether with regulators or controllers The ideal switching regulator shown in Figure 3 2 performs a voltage conversion and input output energy transfer without loss of power by the use of purely reactive components Although an actual switching regulator does have internal losses efficiencies can be quite high generally greater than 80 to 90 Conservation of energy applies so the input power equals the output power This says that in step down buck designs the input current is lower than the output current On the other hand in step up boost designs the input current is greater than the output current Input currents can therefore be quite high in boost applications and this should be kept in mind especially when generating high output voltages from batteries 3 3 SWITCHING REGULATORS THE IDEAL SWITCHING REGULATOR LOSSLESS SWITCHING REGULATOR P in out Efficiency Pout Pin 100 B vinti
168. a typical ESR of 0 1Q to achieve an output ripple voltage as low as 40 to 80mV as well as low input ripple current A typical ADP3000 boost application circuit is shown in Figure 3 35 The input voltage can range from 42V to 3 2V The output is 5V and supplies a load current of 100mA Typical efficiency for the circuit is 80 All components are available in surface mount The ADP3000 can also be used in the buck configuration as shown in Figure 3 36 The input voltage to the regulator is between 5V and 6V and the output is 3V at 100mA Note that in this case the adjustable version of the ADP3000 is used The external divider resistors R1 and R2 are chosen to set the nominal output voltage to 3V All components are available in surface mount and the efficiency of the circuit is approximately 75 3 37 SWITCHING REGULATORS ADP3000 2V TO 5V BOOST APPLICATION L1 6 8 IN 2V TO 3 2V 1N5817 yout O 100mA C1 100p 10V swi ADP3000 5 C2 GND SW2 100pF 10V L1 SUMIDA CD43 6R8 C1 C2 AVX TPS D107 M010R0100 TYPICAL EFFICIENCY 80 Figure 3 35 ADP3000 5V TO 3V BUCK APPLICATION Vin 5V TO 6V 1 5817 L1 SUMIDA CD43 100 C1 C2 AVX TPS D107 M010R0100 TYPICAL EFFICIENCY 75 Figure 3 36 3 38 SWITCHING REGULATORS The ADP3050 is a 1 5A buck converter with an internal saturable NPN switch It utilizes PWM current mode control and operates at a fixed 250kHz switching frequency An a
169. ability Excellent Drift and Long Term Stability Fair Hysteresis Fair Hysteresis Figure 2 11 Low Hysteresis STANDARD POSITIVE OUTPUT THREE TERMINAL REFERENCE HOOKUP 8 PIN DIP PINOUT O POWER COMMON Figure 2 12 OUTPUT LEAD Y SHORT HEAVY TRACE 6 2 13 REFERENCES AND LOW DROPOUT LINEAR REGULATORS Figure 2 12 shows the standard footprint for such a series type IC positive reference in an 8 pin package Note that x numbers refer to the standard pin for that function There are several details which are important Many references allow optional trimming by connecting an external trim circuit to drive the references trim input pin 5 Some bandgap references also have a high impedance PTAT output for temperature sensing 3 The intent here is that no appreciable current be drawn from this pin but it can be useful for such non loading types of connections as comparator inputs to sense temperature thresholds etc All references should use decoupling capacitors on the input pin 2 but the amount of decoupling if any placed on the output 6 depends upon the stability of the reference s output op amp with capacitive load Simply put there is no hard and fast rule for capacitive loads here For example some three terminal types require the output capacitor 1 REF19X and AD1582 85 series while with others it is optional for performance improvement AD780 REF43 The saf
170. ally disappears if the cell is almost fully discharged and then recharged a time or two In practical applications memory is not often a problem because NiCd battery packs are rarely discharged to the same level before recharging Environmental concerns exist regarding the proper disposal of sealed lead acid and NiCd batteries because of hazardous metal content NiMH and Li Ion batteries do not contain significant amounts of pollutant but nevertheless some caution should be used in their disposal The discharge profiles of these four popular type of batteries are shown in Figure 5 5 A discharge current of 0 2C was used in each case Note that NiCd NiMH and SLA batteries have a relatively flat profile while Li Ion batteries have a nearly linear discharge profile BATTERY DISCHARGE PROFILES AT 0 2C RATE 5 TERMINAL VOLTAGE V 4 Li lon 3 2 SLA 1 NiCd and NiMH 0 0 1 2 3 4 5 DISCHARGE TIME HOURS Figure 5 5 5 4 BATTERY CHARGERS BATTERY CHARGING A generalized battery charging circuit is shown in Figure 5 6 The battery is charged with a constant current until fully charged The voltage developed across the RSENSE resistor is used to maintain the constant current The voltage is continuously monitored and the entire operation is under the control of a microcontroller which may even have an on chip A D converter Temperature sensors are used to monitor battery temperature and sometimes ambient temperature GEN
171. and design of the primary side PWM is widely described in the technical literature and is not detailed here However the following explanation should make clear the reasons for the primary side component choices The PWM frequency is set to around 100kHz as a reasonable compromise between inductive and capacitive component sizes switching losses and cost The primary side PWM IC derives its starting Vcc through 100kQ resistor directly from the rectified AC input After start up a simple rectifier circuit driven from a third winding on the transformer charges a 13V zener diode which supplies the Vcc to the 3845 PWM While the signal from the ADP3810 3811 controls the average charge current the primary side should have cycle by cycle limit of the switching current This current limit has to be designed so that with a failed or malfunctioning secondary circuit or optocoupler the primary power circuit components the MOSFET and the transformer won t be overstressed In addition during start up or for a shorted battery to the ADP3810 3811 will not be present Thus the primary side current limit is the only control of the charge current As the secondary side rises above 2 7V the ADP3810 3811 takes over and controls the average current The primary side current limit is set by the Ry TM resistor The current drive of the ADP3810 3811 s output stage directly connects to the photodiode of an optocoupler with no additional circuitr
172. ane whereas the ground plane side of a double sided board is often disrupted with signal crossovers etc If the system has separate analog and digital ground and power planes the analog ground plane should be underneath the analog power plane and similarly the digital ground plane should be underneath the digital power plane There should be no overlap between analog and digital ground planes nor analog and digital power planes The preferred multi layer board arrangement is to embed the signal traces between the power and ground planes as shown in Figure 8 72 These low impedance planes form very high frequency stripline transmission lines with the signal traces The return current path for a high frequency signal on a trace is located directly above and below the trace on the ground power planes The high frequency signal is thus contained inside the PCB thereby minimizing emissions The embedded signal trace approach has an obvious disadvantage debugging circuit traces that are hidden from plain view is difficult 8 74 HARDWARE DESIGN TECHNIQUES TO EMBED OR NOT TO EMBED THAT IS THE QUESTION Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA BEFORE AFTER Route Power EEE Power Route 2 Ground E Route EE Route pem Ground S B Advantages of Embedding Lower impedances therefore lower emissions and crosstalk Reduction in emissions and crossta
173. ansfer assuming ideal circuit elements This fundamental method of energy transfer forms the basis for all switching regulators IDEAL STEP DOWN BUCK CONVERTER The fundamental circuit for an ideal step down buck converter is shown in Figure 9 9 The actual integrated circuit switching regulator contains the switch control circuit and may or may not include the switch depending upon the output current requirement The inductor diode and load bypass capacitor are external BASIC STEP DOWN BUCK CONVERTER ERROR AMPLIFIER AND SWITCH CONTROL CIRCUIT 1 ton toff SW ON SW OFF on off Figure 3 9 The output voltage is sensed and then regulated by the switch control circuit There are several methods for controlling the switch but for now assume that the switch is controlled by a pulse width modulator PWM operating at a fixed frequency f The actual waveforms associated with the buck converter are shown in Figure 3 10 When the switch is on the voltage VIN VOUT appears across the inductor and the inductor current increases with a slope equal to VTN VQUuT L see Figure 3 10B When the switch turns off current continues to flow through the inductor and into the load remember that the current cannot change instantaneously in an inductor with the ideal diode providing the return current path The voltage across the inductor is now VOUT but the polarity has reversed Therefore the inductor 3 10 SWITCHING R
174. approved filter It should be installed in such a manner that it is the first thing the AC line sees upon entering the equipment Standard three wire IEC style line cords are designed to mate with three terminal male connectors integral to many line filters This is the best way to achieve this function as it automatically grounds the third wire to the shell of the filter and equipment chassis via a low inductance path Commercial power line filters can be quite effective in reducing AC power line noise This noise generally has both common mode and differential mode components Common mode noise is noise that is found on any two of the three power connections black white or green with the same amplitude and polarity In contrast differential mode noise is noise found only between two lines By design most commercially available filters address both noise modes see Reference 16 8 43 HARDWARE DESIGN TECHNIQUES REFERENCES NOISE REDUCTION AND FILTERING 1 10 11 12 13 14 15 16 8 44 EMC Design Workshop Notes Kimmel Gerke Associates Ltd St Paul MN 55108 612 330 3728 Walt Jung Dick Marsh Picking Capacitors Parts 1 amp 2 Audio February March 1980 Tantalum Electrolytic and Ceramic Capacitor Families Kemet Electronics Box 5928 Greenville SC 29606 803 963 6300 Type HFQ Aluminum Electrolytic Capacitor and type V Stacked Polyester Film Capacitor Panasonic 2 Panasonic Way
175. ar regulators switching regulators subject capacitors to large AC currents AC currents can cause heating in the dielectric material and change the temperature dependent characteristics of the capacitor Also the capacitor is more likely to fail at the higher temperatures produced by the 3 64 SWITCHING REGULATORS ripple current Fortunately most manufacturers provide ripple current ratings and this problem can be averted if understood Calculating the exact ripple current can be tedious especially with complex switching regulator waveforms Simple approximations can be made however which are sufficiently accurate Consider first the buck converter input and output currents refer to Figure 3 63 The rms input capacitor ripple current can be approximated by a square wave having a peak to peak amplitude equal to IOUT The rms value of this square wave is therefore 7 2 The output capacitor current waveform can be approximated by a sawtooth waveform having a peak to peak amplitude of 0 2IQUT The rms value of this sawtooth is therefore approximately 0 2IQUT 12 or 0 0619pT BUCK CONVERTER INPUT AND OUTPUT CAPACITOR RMS RIPPLE CURRENT APPROXIMATIONS INPUT CURRENT iy OUTPUT CURRENT INPUT CAPACITOR RMS OUTPUT CAPACITOR RMS RIPPLE CURRENT 0 5 RIPPLE CURRENT Ip p N12 0 06 lout Figure 3 63 Similarly for a boost converter see waveforms shown in Figure 3 64 the input capacitor rms ripple current is
176. are increased versus the standard leadframe on the left DETAILS OF THERMAL COASTLINE PACKAGE STANDARD FRAME THERMAL COASTLINE FRAME Face to face distance from lead to paddle ene reduced by a factor of 1 5 to 2 Width of adjoining Center of T faces increased by Package factor of 2 to 2 5 Center of Package Figure 2 40 2 47 REFERENCES AND LOW DROPOUT LINEAR REGULATORS LDO REGULATOR CONTROLLERS To complement the anyCAP series of standalone LDO regulators there is also the LDO regulator controller The regulator controller IC picks up where the standalone regulator IC is no longer useful in either load current or power dissipation terms and uses an external PMOS FET for the pass device The ADP3310 is a basic LDO regulator controller device designed for fixed output voltage applications while operating from sources over a range of 3 8 to 15V and a temperature range of 40 to 85 C The actual ADP3310 device ordered would be specified as ADP3310AR YY where the YY is a voltage designator suffix such as 2 8 3 3 3 or 5 for those respective voltages The AR portion of the part number designates the SO 8 Thermal Coastline 8 lead package A summary of the main features of the ADP3310 device is listed in Figure 2 41 anyCAP ADP3310 LDO REGULATOR CONTROLLER FEATURES Controller drives external PMOS power FETs User FET choice determines and Vy performance Small 2 chip
177. are wired point to point in the air a type of construction strongly advocated by Robert A Pease of National Semiconductor Reference 6 and sometimes known as bird s nest construction there is always the risk of the circuitry being crushed and resulting short circuits Also if the circuitry rises high above the ground plane the screening effect of the ground plane is diminished and interaction between different parts of the circuit is more likely Nevertheless the technique is very practical and widely used because the circuit may easily be modified assuming the person doing the modifications is adept at using a soldering iron solder wick and a solder sucker Copper clad boards are available with pre drilled holes on 0 1 centers Reference 7 Because of the loss of copper area due to the pre drilled holes this technique does not provide as low a ground impedance as a completely covered copper clad board However this type of board is convenient if the ICs in the prototype have the proper pin spacing In a variation of this technique the ICs and other components are mounted on the non copper clad side of the board The holes are used as vias and the point to point wiring is done on the copper clad side of the board The copper surrounding each 8 4 HARDWARE DESIGN TECHNIQUES hole used for a via must be drilled out to prevent shorting This approach requires that all IC pins be on 0 1 centers Low profile sockets can be used
178. ary between near and far field If a circuit is within 5 inches of a 350MHz interference source then the circuit operates in the near field of the interference If the distance is greater than 5 inches the circuit operates in the far field of the interference Regardless of the type of interference there is a characteristic impedance associated with it The characteristic or wave impedance of a field is determined by the ratio of its electric or E field to its magnetic or H field In the far field the ratio of the electric field to the magnetic field is the characteristic wave impedance of free space given by Zo 3770 In the near field the wave impedance is determined by the nature of the interference and its distance from the source If the interference source is high current and low voltage for example a loop antenna or a power line transformer the field is predominately magnetic and exhibits a wave impedance which is less than 3770 If the source is low current and high voltage for example rod antenna or a high speed digital switching circuit then the field is predominately electric and exhibits a wave impedance which is greater than 3770 Conductive enclosures can be used to shield sensitive circuits from the effects of these external fields These materials present an impedance mismatch to the incident interference because the impedance of the shield is lower than the wave impedance of the incident field The effectivene
179. ased insects hence the name Figure 8 2 shows a hand wired breadboard using two high speed op amps which gives excellent performance in spite of its lack of esthetic appeal The IC op amps are mounted upside down on the copper board with the leads bent over The signals are connected with short point to point wiring The characteristic impedance of a wire over a ground plane is about 1200 although this may vary as much as 40 depending on the distance from the plane The decoupling capacitors are connected directly from the op amp power pins to the copper clad ground plane When working at frequencies of several hundred MHz it is a good idea to use only one side of the board for ground Many people drill holes in the board and connect both sides together with short pieces of wire soldered to both sides of the board If care is not taken however this may result in unexpected ground loops between the two sides of the board especially at RF frequencies 8 3 HARDWARE DESIGN TECHNIQUES HANDWIRED DEADBUG PROTOTYPE Figure 8 2 Pieces of copper clad board may be soldered at right angles to the main ground plane to provide screening or circuitry may be constructed on both sides of the board with connections through holes with the board itself providing screening In this case the board will need standoffs at the corners to protect the components on the underside from being crushed When the components of a breadboard of this type
180. aspects of the regulator application 2 41 REFERENCES AND LOW DROPOUT LINEAR REGULATORS BENEFITS OF anyCAP LDO TOPOLOGY Internal Dominates Response Rolloff C Can Range from 0 47pF min to Infinity Low and Ultra Low C ESR is OK MLCC Types for C Work is Physically Smallest Solution No ESR Exclusion Zones Fast Load Transient Response and Good Line Rejection Figure 2 34 The anyCAP DO series devices The major specifications of the anyCAPTM series of LDO regulators are summarized in Figure 2 35 The devices include both single and dual output parts with current capabilities ranging from 50 to 200mA Rather than separate individual specifications for output tolerance line and load regulation plus temperature the anyCAP series devices are rated simply for a combined total accuracy figure This accuracy is either 0 8 at 25 C or 1 4 over the temperature range with the device operating over an input range 0 3 or 0 5V up to 12V With total accuracy being covered by one clear specification the designer can then achieve a higher degree of confidence It is important to note that this method of specification also includes operation within the regulator dropout range unlike some LDO parts specified for higher input output voltage difference conditions anyCAP SERIES LDO REGULATOR DEVICES 2 42 Part Vm IL I Accuracy Package Comment Number
181. ate C RATE DEFINITION Battery Charge and Discharge Currents are Expressed Normalized in Terms of C Rate C Rate 1 hour Where C is the Battery Capacity Expressed in A hour or mA hour B Example A 1000 mA h Battery has a C Rate of 1000mA The Current Corresponding to 1C is 1000mA The Current Corresponding to 0 1C is 100mA The Current Corresponding to 2C is 2000mA E For a Given Cell Type the Behavior of Cells with Varying Capacity is Similar at the same C rate Figure 5 2 There are a number of other figures of merit used to characterize batteries which are summarized in Figure 5 3 These figures of merit are used to characterize various battery chemistries as shown in Figure 5 4 Note that in Figure 5 4 the approximate chronology of battery technology is from left to right A few terms relating to batteries deserve further clarification Self discharge is the rate at which a battery discharges with no load Li Ion batteries are a factor of two better than NiCd or NiMH in this regard The discharge rate is the maximum allowable load or discharge current expressed in units of C Rate Note that all chemistries can be discharged at currents higher than the battery C Rate The number of charge and discharge cycles is the average number of times a battery can be discharged and then recharged and is a measure of the battery s service life 5 2 BATTERY CHARGERS RECHARGEABLE BATTERY FIGURES OF MERIT Cel
182. aware that if the reference source impedance is too high dynamic loading can cause the reference input to shift by more than 5mV A bypass capacitor on the output of a reference may help it to cope with load transients but many references are unstable with large capacitive loads Therefore it is quite important to verify that the device chosen will satisfactorily drive the output capacitance required In any case the input to references should always be decoupled with at least 0 1uF and with an additional 5 50pF if there is LF ripple on its supply See Figure 2 12 again Since some references misbehave with transient loads either by oscillating or by losing accuracy for comparatively long periods it is advisable to test the pulse response of voltage references which may encounter transient loads A suitable circuit is shown in Figure 2 20 In a typical voltage reference a step change of 1mA produces the transients shown Both the duration of the transient and the amplitude of the ringing increase when a 0 01pF capacitor is connected to the reference output 2 21 REFERENCES AND LOW DROPOUT LINEAR REGULATORS MAKE SURE REFERENCE IS STABLE WITH LARGE CAPACITIVE LOADS TOP TRACE NO LOAD C 0 50mV div 1mA to 2 STEP SCOPE t UNDER Pe I BOTTOM TRACE C 0 01pF 200mV div PULSE BOTH TRACES 5us div Figure
183. based bipolar references operating at an equivalent current the temperature drift is low and linear at 3 8 ppm C allowing easier compensation when required and the series has lower hysteresis than bandgaps Thermal hysteresis is a low 50ppm over a 40 to 125 C range less that half that of a typical bandgap device Finally the long term stability is excellent typically only 0 2ppm 1000 hours Figure 2 11 summarizes the pro and con characteristics of the three reference architectures bandgap buried zener and Modern IC references come in a variety of styles but series operating fixed output positive types do tend to dominate These devices can use bandgap based bipolars JFETs or buried zeners at the device core all of which has an impact on the part s ultimate performance and application suitability They may or may not also be low power low noise and or low dropout and be available within a certain package Of course in a given application any single one of these differentiating factors can drive a choice thus it behooves the designer to be aware of all the different devices available 2 12 REFERENCES AND LOW DROPOUT LINEAR REGULATORS CHARACTERISTICS OF REFERENCE ARCHITECTURES BANDGAP BURIED ZENER XFETTM lt 5V Supplies gt 5V Supplies lt 5V Supplies High Noise Low Noise Low Noise High Power High Power Low Power Fair Drift and Long Term Stability Good Drift and Long Term St
184. based systems are increasingly called upon to observe some form of thermal management All semiconductors have some specified safe upper limit for junction temperature TJ usually on the order of 150 C but sometimes 175 Like maximum power supply potentials maximum junction temperature is a worst case limitation which shouldn t be exceeded In conservative designs it won t be approached by less than an ample safety margin This is a critical point since the lifetime of all semiconductors is inversely related to their operating junction temperature The cooler semiconductors can be kept during operation the more closely they will approach maximum useful life Thermal basics The general symbol 0 is used for thermal resistance that is 0 thermal resistance in units of C watt or C W 0JA and 0JC are two more specific terms used in dealing with semiconductor thermal issues which are explained below In general a device with a thermal resistance 0 equal to 100 C W will exhibit a temperature differential of 100 C for a power dissipation of 1W as measured between two reference points Note that this is a linear relation so a 500mW dissipation in the same part will produce a 50 C differential and so forth For any power P in watts calculate the effective temperature differential AT in C as AT P x 6 where is the total applicable thermal resistance Figure 8 43 summarizes these thermal relationships As th
185. be reduced to a mA or less using bipolar technology and to only a few in CMOS parts In regulators which offer a shutdown mode the shutdown state standby current will be reduced to a uA or less Nearly all regulators will have some means of current limiting and over temperature sensing to protect the pass device against failure Current limiting is usually by a series sensing resistor for high current parts or alternately by a more simple drive current limit to a controlled B pass device which achieves the same end For higher voltage circuits this current limiting may also be combined with voltage limiting to 2 28 REFERENCES AND LOW DROPOUT LINEAR REGULATORS provide complete load line control for the pass device All power regulator devices will also have some means of sensing over temperature usually by means of a fixed reference voltage and a Vpg based sensor monitoring chip temperature When the die temperature exceeds a dangerous level above 150 C this can be used to shutdown the chip by removing the drive to the pass device In some cases an error flag output may be provided to warn of this shutdown and also loss of regulation from other sources PASS DEVICES AND THEIR ASSOCIATED TRADEOFFS The discussion thus far has not treated the pass device in any detail In practice this major part of the regulator can actually take on quite a number of alternate forms Precisely which type of pass device is chosen has a ma
186. be wide heavy traces and are indicated by the wide interconnection lines on the diagram The low current ground GND and sense pins of the ADP3310 are connected directly to the load so that the regulator regulates the voltage at the load rather than at its own output The IS and VIN connections to the Rg current sense resistor should be made directly to the resistor terminals to minimize parasitic resistance since the current limit resistor is typically a very low value milliohms In fact for very low values it may actually consist of a PC board trace of the proper width length and thickness to yield the desired resistance GROUNDING AND SIGNAL ROUTING TECHNIQUES FOR LOW DROPOUT REGULATORS METHOD 1 Rs Is GATE ADP3310 LDO LINEAR REG GND POWER COMMON m SHORT CONNECTION gt SHORT TO GROUND PLANE HEAVY TRACES Figure 8 10 8 14 HARDWARE DESIGN TECHNIQUES The input decoupling capacitor C1 should be connected with short leads at the regulator input in order to absorb any transients which may couple onto the input voltage line Similarly the load capacitor C2 should have minimum lead length in order to absorb transients at that point and prevent them from coupling back into the regulator The single point connection to the low impedance ground plane is made directly at the load Figure 8 11 shows a grounding arrangement which is similar to that of Figure 8 10 with the excep
187. between 2 2Vr and 3Viw 2Vp Choosing R 33 2kO yields an output voltage VoyyT9 of 12V for a nominal input voltage of 5V Regulation is maintained for output currents up to approximately 20mA 4 20 SWITCHED CAPACITOR VOLTAGE CONVERTERS REGULATED 12V FROM A 5V INPUT USING THE ADP3607 5 1N5817 V ADP3607 5 Gs Figure 4 23 4 21 BATTERY CHARGERS SECTION 5 BATTERY CHARGERS Walt Kester Joe Buxton INTRODUCTION Rechargeable batteries are vital to portable electronic equipment such as laptop computers and cell phones Fast charging circuits must be carefully designed and are highly dependent on the particular battery s chemistry The most popular types of rechargeable batteries in use today are the Sealed Lead Acid SLA Nickel Cadmium NiCd Nickel Metal Hydride NiMH and Lithium Ion Li Ion Li Ion is fast becoming the chemistry of choice for many portable applications because it offers a high capacity to size weight ratio and a low self discharge characteristic RECHARGEABLE BATTERY CONSIDERATIONS IN PORTABLE EQUIPMENT Amp Hour Capacity C and Cell Voltage B Multiple Cell Configurations Series Parallel Combinations Matching Requirements Weight and Volume Cost of Battery Pack Battery Chemistry Sealed Lead Acid SLA Nickel Cadmium NiCd Nickel Metal Hydride NiMH Lithium lon Li lon Lithium Metal Relatively New Discharge Characteristics Charge Character
188. buffer amplifier which also provides convenient voltage scaling to standard levels An improved three terminal bandgap reference the AD580 is shown in Figure 2 4 Popularly called the Brokaw Cell see References 2 and 3 this circuit provides on chip output buffering which allows good drive capability and standard output voltage scaling The AD580 was the first precision bandgap based IC reference and variants of the topology have influenced further generations of both industry standard references such as the REF01 and REFO2 series as well as more recent ADI parts such as the REF195 series the AD680 AD780 and the AD1582 85 series The AD580 has two 8 1 emitter scaled transistors Q1 Q2 operating at identical collector currents and thus 1 8 current densities by virtue of equal load resistors and a closed loop around the buffer op amp Due to the resultant smaller of the 8x area Q2 R2 in series with Q2 drops the AVpg voltage while R1 due to the current relationships drops a PTAT voltage V1 R1 V1 2 x x AV 7 1 R2 BE 2 5 REFERENCES AND LOW DROPOUT LINEAR REGULATORS AD580 PRECISION BANDGAP REFERENCE USES BROKAW CELL Vin Vout 2 5V Vz 1 205V A TRANSISTOR AREA Figure 2 4 The bandgap cell reference voltage Vz appears at the base of Q1 and is the sum of VpE Q1 and or 1 205V the bandgap voltage VZ VBE Q1 V1 R1 VBE Q1 2 R2 X AVBE R1 kT J1 V 2x x xl BE QD
189. c isolation as well as allowing the buck boost function to be easily performed However adding a transformer to the circuit creates a more complicated and expensive design as well as increasing the physical size The basic flyback buck boost converter circuit is shown in Figure 3 24 It is derived from the buck boost converter topology When the switch is on the current builds up in the primary of the transformer When the switch is opened the current reverts to the secondary winding and flows through the diode and into the load The relationship between the input and output voltage is determined by the turns ratio N and the duty cycle D per the following equation VIN D 1 D VOUT 3 24 SWITCHING REGULATORS A disadvantage of the flyback converter is the high energy which must be stored in the transformer in the form of DC current in the windings This requires larger cores than would be necessary with pure AC in the windings ISOLATED TOPOLOGY FLYBACK CONVERTER BUCK BOOST DERIVED MN D N VOUT I D D Duty Cycle Figure 3 24 The basic forward converter topology is shown in Figure 3 25 It is derived from the buck converter This topology avoids the problem of large stored energy in the transformer core However the circuit is more complex and requires an additional magnetic element a transformer an inductor an additional transformer winding plus three diodes When the switch is on cur
190. cations the exact value is not very critical so approximations can be used with a high degree of confidence The heart of a switching regulator analysis involves a thorough understanding of the inductor current waveform Figure 3 50 shows an assumed inductor current waveform which is also the output current for a buck converter such as the ADP3000 which uses the gated oscillator PBM switch modulation technique Note that this waveform represents a worst case condition from the standpoint of storing energy in the inductor where the inductor current starts from zero on each cycle In high output current applications the inductor current does not return to zero but ramps up until the output voltage comparator senses that the oscillator should be turned off at which time the current ramps down until the comparator turns the oscillator on again This assumption about the worst case waveform is necessary because in a simple PBM regulator the oscillator duty cycle remains constant regardless of input voltage or output load current Selecting the inductor value using this assumption will always ensure that there is enough energy stored in the inductor to maintain regulation It should be emphasized that the following inductance calculations for the PBM buck and boost regulators should be used only as a starting point and larger or smaller values may actually be required depending on the specific regulator and the input output conditions CALCU
191. ces are highly package dependent as different materials have differing degrees of thermal conductivity As a general rule of thumb thermal resistance for the conductors within packaging materials is closely analogous to electrical resistances that is copper is the best followed by aluminum steel and so on Thus copper lead frame packages offer the highest performance lowest 0 A summary of the thermal resistances of various IC packages is shown in Figures 8 44 8 45 and 8 46 In general most of these packages do not lend themselves to easy heat sink attachment with notable exceptions such as the older round metal can types or the TO 220 package Devices which are amenable to heat sink attachment will often be noted by dramatically lower than the See for example the 15 pin SIP package used by the AD815 the TO 220 package and the TO 263 package 8 46 HARDWARE DESIGN TECHNIQUES STANDARD PACKAGE THERMAL RESISTANCES 1 Package ADI Designation JA C W JC C W Comment 3 pin SOT 23 SOT 23 3 300 180 ADT45 ADT50 5 pin SOT 23 SOT 23 5 190 ADTO5 6 pin SOT 23 SOT 23 6 165 92 ADP3300 8 pin plastic DIP N 8 90 AD823 8 pin ceramic D 8 110 22 AD712 8 pin SOIC R 8 160 60 8 pin SOIC R 8 90 60 ADP3367 Thermal Coastline 8 pin metal can H 08A TO 99 150 45 OP07 10 pin metal can H 10A TO 100 150 25 AD582 12 pin metal can H
192. child Semiconductor application note AN1029 April 1996 http www fairchildsemi com Rob Blattner Wharton McDaniel Thermal Management in On Board DC to DC Power Conversion Temic application note http www temic com 9 series surface mount current sensing resistors KRL Bantry Components 160 Bouchard Street Manchester NH 03103 3399 603 668 3210 2 57 SWITCHING REGULATORS SECTION 3 SWITCHING REGULATORS Walt Kester Brian Erisman INTRODUCTION Virtually all of today s electronic systems require some form of power conversion The trend toward lower power portable equipment has driven the technology and the requirement for converting power efficiently Switchmode power converters often referred to simply as switchers offer a versatile way of achieving this goal Modern IC switching regulators are small flexible and allow either step up boost or step down buck operation When switcher functions are integrated and include a switch which is part of the basic power converter topology these ICs are called switching regulators When no switches are included in the IC but the signal for driving an external switch is provided it is called a switching regulator controller Sometimes usually for higher power levels the control is not entirely integrated but other functions to enhance the flexibility of the IC are included instead In this case the device might be called a controller of sorts
193. citor for example is actually a handful Metalized as opposed to foil electrodes does help to reduce size but even the highest dielectric constant units among film types polyester polycarbonate are still larger than any electrolytic even using the thinnest films with the lowest voltage ratings 50V Where film types excel is in their low dielectric losses a factor which may not necessarily be a practical advantage for filtering switchers For example ESR in film capacitors can be as low as 10m2 or less and the behavior of films generally is very high in terms of Q In fact this can cause problems of spurious resonance in filters requiring damping components Typically using a wound layer type construction film capacitors can be inductive which can limit their effectiveness for high frequency filtering Obviously only non inductively made film caps are useful for switching regulator filters One specific style which is non inductive is the stacked film type where the capacitor plates are cut as small overlapping linear sheet sections from a much larger wound drum of dielectric plate material This technique offers the low inductance attractiveness of a plate sheet style capacitor with conventional leads see References 9 10 11 Obviously minimal lead length should be used for best high frequency effectiveness Very high current polycarbonate film types are also available specifically designed for switching power supplies with a var
194. coefficient which is the conductor resistance in milliohms inch divided by the trace width W For example the first entry for 1 2 ounce copper is 0 983 milliohms inch W So for a reference trace width of 0 1 the resistance would be 9 83 milliohms inch Since these are all linear relationships everything scales for wider skinnier traces or for differing copper weights As an example to design a 50 milliohm Rg for the circuit of Fig 2 43 using 1 2 ounce copper a 2 54 length of a 0 05 wide PCB trace could be used PRINTED CIRCUIT COPPER RESISTANCE DESIGN FOR LDO CONTROLLERS Copper Thickness Resistance Coefficient Reference 0 1 Milliohms inch W Inch wide trace trace width W in Milliohms inch inches 1 2 oz ft 0 983 W 9 83 1 2 ft 0 491 W 4 91 2 oz ft 0 246 W 2 46 3 oz ft 0 163 W 1 63 Figure 2 44 To minimize current limit sense voltage errors the two connections to Rg should be made four terminal style as is noted in Figure 2 43 again It is not absolutely 2 54 REFERENCES AND LOW DROPOUT LINEAR REGULATORS necessary to actually use four terminal style resistors except for the highest current levels However as a minimum the heavy currents flowing in the source circuit of the pass device should not be allowed to flow in the ADP3310 sense pin traces To minimize such errors the VIN connection trace to the ADP3310 should connect close to the body of Rg or
195. ction to case is 180 C W The thermal resistance case to ambient is the difference between OJA and 8jc and is determined by the characteristics of the thermal connection With no air flow and the device soldered on a PC board is 300 C W The temperature sensor s power dissipation Pp is the product of the total voltage across the device and its total supply current including any current delivered to the load The rise in die temperature above the medium s ambient temperature is given by Tj Ppx 9jc TA Thus the die temperature rise of an unloaded ADT45 ADT50 SOT 23 3 package soldered on a board in still air at 25 C and driven from a 5V supply quiescent current 60pA Pp 300pW is less than 0 09 C In order to prevent further temperature rise it is important to minimize the load current always keeping it less than 100pA The transient response of the ADT45 ADT50 sensors to a step change in temperature is determined by the thermal resistances and the thermal mass of the die and the case The thermal mass of the case varies with the measurement medium since it includes anything that is in direct contact with the package In all practical cases the thermal mass of the case is the limiting factor in the thermal response time of the sensor and can be represented by a single pole RC time constant Thermal mass is often considered the thermal equivalent of electrical capacitance The thermal time cons
196. d for another poor line rejection 2 40 REFERENCES AND LOW DROPOUT LINEAR REGULATORS THE SOLUTION TO C SENSITIVITY POLE SPLIT COMPENSATION WRONG WAY EXAMPLE Vout Figure 2 33 The anyCAP Pole Splitting Topology Returning to the anyCAP series topology Fig 2 32 again it can be noted that in this case 15 isolated from the pass device s base and thus input ripple variations by the wideband non inverting driver But insofar as frequency compensation is concerned because of this buffer s isolation Ccomp still functions as a modified pole splitting capacitor see Reference 9 and it does provide the benefits of a buffered Cy independent single pole response The regulator s frequency response is dominated by the internal compensation and becomes relatively immune to the value and ESR of load capacitor Thus the name anyCAP for the series is apt as the design is tolerant of virtually any output capacitor type The benefits of the anyCAP topology are summarized by Figure 2 34 As can be noted be as low as 0 47uF and it can also be a multi layer ceramic capacitor MLCO type This allows a very small physical size for the entire regulation function such as when a SOT 23 packaged anyCAP LDO is used for example the ADP3300 device Because of the in sensitivity to the designer needn t worry about such things as ESR zones and can better concentrate on the system
197. d across R2 The AVpg components have opposite polarity TCs is proportional to absolute temperature PTAT while is complementary to absolute temperature CTAT The summed output is Vg and when it is equal to 1 205V silicon bandgap voltage the TC is a minimum The bandgap reference technique is attractive in IC designs because of several reasons among these are the relative simplicity and the avoidance of zeners and their noise However very important in these days of ever decreasing system supplies is the fundamental fact that bandgap devices operate at low voltages i e lt 5V Not only are they used for stand alone IC references but they are also used within the designs of many other linear ICs such as ADCs DACs and op amps Buffered forms of 1 2V two terminal bandgap references such as the AD589 IC remain stable under varying load currents The H 02A metal can AD589 a 1 235V reference handles 50 to 5mA with an output impedance of 0 60 and TCs ranging between 10 and 100ppm C The more recent and functionally similar AD1580 a 1 225V reference is in the tiny SOT 23 package and handles the same nominal currents as the AD589 with TCs of 50 and 100ppm C However the basic designs of Figure 2 3 suffer from load and current drive sensitivity plus the fact that the output needs accurate scaling to more useful levels i e 2 5V 5V etc The load drive issue is best addressed with the use of a
198. d to the basic form which was described with Figure 2 3 Resistor R8 drops a PTAT voltage which drives the Darlington connected error amplifier Q9 Q10 The negative TC Vpgps of Q9 Q10 Q12 Q13 are summed with this PTAT voltage and this sum produces a temperature stable 5V output voltage Current buffering of the error amplifier Q10 is provided by PNP Q11 which drives the NPN pass devices 2 31 REFERENCES AND LOW DROPOUT LINEAR REGULATORS SIMPLIFIED SCHEMATIC OF LM309 FIXED 5V 1A THREE TERMINAL REGULATOR n E ad Q19 C Q2 Vour V a13 R9 Q3 V R1 V a12 Q11 Du Q4 c1 R3 R8 a9 a7 Q5 R4 R7 R5 R6 V as COMMON O Figure 2 27 Later developments in references and three terminal regulation techniques led to the development of the voltage adjustable regulator The original IC to employ this concept was the LM317 see Reference 2 which is shown in simplified schematic form in Figure 2 28 Note that this design does not use the same form of reference as in the LM309 Instead Q17 Q19 etc are employed as a form of a Brokaw bandgap reference cell see Figure 2 4 again and Reference 3 This adjustable regulator bootstraps the reference cell transistors Q17 Q19 and the error amplifier transistors Q16 Q18 The output of the error amplifier drives Darlington pass transistors Q25 Q26 through buffer Q12 The basic reference cell produces a fixed voltage of 1 25V which appears between the Vout and ADJ pin
199. dbook of Batteries 2 4 Edition McGraw Hill 1995 Chester Simpson Rechargeable Lithium Cells Power to Burn for Portables ED Analog Applications Issue June 27 1994 p 39 5 25 TEMPERATURE SENSORS SECTION 6 TEMPERATURE SENSORS Walt Kester INTRODUCTION Measurement of temperature is critical in modern electronic devices especially expensive laptop computers and other portable devices with densely packed circuits which dissipate considerable power in the form of heat Knowledge of system temperature can also be used to control battery charging as well as prevent damage to expensive microprocessors Compact high power portable equipment often has fan cooling to maintain junction temperatures at proper levels In order to conserve battery life the fan should only operate when necessary Accurate control of the fan requires a knowledge of critical temperatures from the appropriate temperature sensor APPLICATIONS OF TEMPERATURE SENSORS Monitoring Portable Equipment CPU Temperature Battery Temperature Ambient Temperature B Compensation Oscillator Drift in Cellular Phones Thermocouple Cold Junction Compensation B Control Battery Charging Process Control Figure 6 1 Accurate temperature measurements are required in many other measurement systems such as process control and instrumentation applications In most cases because of low level nonlinear outputs the sensor output must be properly condit
200. directly beneath the ADP3000 The two resistors to the right of the ADP3000 set the output voltage and the resistor above and to the left of the ADP3000 is the current limiting resistor The loop of wire allows the inductor current to be monitored with a current probe Note that the resistors are in individual pin sockets to allow easy modifications 8 7 HARDWARE DESIGN TECHNIQUES HANDWIRED PROTOTYPE ADP3000 BOTTOM VIEW Figure 8 5 IC sockets however can degrade the performance of high speed or high precision analog ICs Although they make prototyping easier even low profile sockets often introduce enough parasitic capacitance and inductance to degrade the performance of the circuit If sockets must be used in high speed circuits an IC socket made of individual pin sockets sometimes called cage jacks mounted in the ground plane board may be acceptable clear the copper on both sides of the board for about 0 5mm around each ungrounded pin socket and solder the grounded ones to ground on both sides of the board Both capped and uncapped versions of these pin sockets are available AMP part numbers 5 330808 3 and 5 330808 6 respectively The pin sockets protrude through the board far enough to allow point to point wiring interconnections between them see Figure 8 6 The spring loaded gold plated contacts within the pin socket makes good electrical and mechanical connection to the IC pins Multiple insertions however may d
201. duct the maximum short circuit current both instantaneously and longer term Thermal Design The maximum allowable thermal resistance between the FET junction and the highest expected ambient temperature must be taken into account to determine the type of FET package and heat sink used if any 2 51 REFERENCES AND LOW DROPOUT LINEAR REGULATORS Whenever possible to do so reliably the FET pass device can be directly mounted to the PCB and the available PCB copper lands used as an effective heat sink This heat sink philosophy will likely be adequate when the power to be dissipated in the FET is on the order of 1 2W or less Note that the very nature of an LDO helps this type of design immensely as the lower voltage drop across the pass device reduces the power to be dissipated Under normal conditions for example Q1 of Figure 2 43 dissipates less than 1W at a current of 1A since the drop across the FET is less than 1V To use PCB lands as effective heat sinks with SO 8 and other SMD packages the pass device manufacturer s recommendations for the lowest mounting should be followed see References 11 and 12 In general these suggestions will likely parallel the 5 rules noted above under LDO regulator thermal considerations for SO 8 and SOT 23 packaged anyCAP LDOs For lowest possible thermal resistance also connect multiple FET pins together as follows Electrically connect multiple FET source and drain pins in paral
202. e PTAT and given by 6 20 TEMPERATURE SENSORS 2R1 VBE VN ip RE kT In N R2 R2 q VPTAT CLASSIC BANDGAP TEMPERATURE SENSOR Vin BROKAW CELL VBE VN ain R1 kT Figure 6 23 The bandgap cell reference voltage VBANDGAP appears at the base of Q1 and is the sum of Vgg Q1 and VBE Q1 is complementary to absolute temperature and summing it with causes the bandgap voltage to be constant with respect to temperature assuming proper choice of R1 R2 ratio and N to make the bandgap voltage equal to1 205V This circuit is the basic band gap temperature sensor and is widely used in semiconductor temperature sensors Current and Voltage Output Temperature Sensors The concepts used in the bandgap temperature sensor discussion above can be used as the basis for a variety of IC temperature sensors to generate either current or voltage outputs AD592 and TMP17 see Figure 6 24 are current output sensors which have scale factors of 1nA K The sensors do not require external calibration and are available in several accuracy grades The AD592 is available in three accuracy grades The highest grade version AD592CN has a maximum error 25 C of 0 5 C and 1 0 C error from 25 C to 105 C Linearity error is 0 35 The TMP17 is available in two accuracy grades The highest grade version TMP17F has a maximum error 25 C of 2 5 C and 3 5 C error from 40
203. e coefficient of the XFET core such that the overall net temperature drift of the reference is typically in a range of 3 to 8ppm C During manufacture the R1 R3 scaling resistance values are adjusted to produce the different voltage output options of 2 048 2 5 4 096 and 5 0V for the ADR290 ADR291 ADR292 and ADR293 family ADR29X This ADR29X family of series mode references is available in 8 pin packages with a standard footprint as well as a TO 92 lead format They operate from supplies of Voit plus 200mV to 15V with a typical quiescent current of 12 uA and output currents of up to 5 mA summary of specifications for the family appears in Figure 2 10 2 11 REFERENCES AND LOW DROPOUT LINEAR REGULATORS ADR290 ADR293 XFET SERIES SPECIFICATIONS Vout 2 048 2 500 4 096 amp 5 000V 2 7V to 15V Supply Range Supply Current 120A max Initial Accuracy 2 mV max Temperature Coefficient 8 ppm C max Low Noise 6uVp p 0 1 10Hz Wideband Noise 420nV VHz 1kHz Long Term Drift 0 2ppm 1khrs High Output Current 5mA min Temperature Range 40 C to 125 C Standard REFO02 Pinout 8 Lead Narrow Body SOIC 8 Lead TSSOP and 3 Lead TO 92 Figure 2 10 The XFET architecture offers performance improvements over bandgap and buried zener references particularly for systems where operating current is critical yet drift and noise performance must still be excellent XFET noise levels are lower than bandgap
204. e for a fax on demand through our automated AnalogFax system Data sheets are available 7 days a week 24 hours a day Product faxcode cross reference listings are available by calling the above number and following the prompts There is a short index with just part numbers faxcodes page count and revision for each data sheet There is also a longer index sorted by product type with short descriptions World Wide Web and Internet Our address is http www analog com Use the browser of your choice and follow the prompts We also provide extensive DSP literature support on an Internet FTP site Type ftp ftp analog com or ftp 137 71 23 11 Log in as anonymous using your e mail address for your password Analog Devices Literature Distribution Center Call 800 262 5643 and select option two from the voice prompts or call 781 461 4700 for direct access or fax your request to 781 821 4273 DSP Bulletin Board Service For the latest updates call 781 461 4258 8 data bits 1 stop bit no parity 300bps to 14 4kbps Europe and Israel Fax Retrieval Telephone number 49 8765 9300 xxxx where xxxx is the faxcode For a list of faxcodes dial 49 8765 9300 1000 1000 is the code for the faxcode cross reference listing World Wide Web Our address is http www analog com use the browser of your choice and follow the prompts Analog Devices Sales Offices Call your local sales office and request a data sheet A Worldwide Sales Directory inc
205. e watchdog input has not been toggled within a specified time 4 A 1 25V threshold detector for power fail warning low battery detection or to monitor a supply other than 5V An application of the ADM8691 is shown in Figure 7 3 Resistors R1 and R2 divide the regulator input voltage down and provide a Power Fail Indication when the voltage at the POWER FAIL INPUT falls below 1 25V 7 1 HARDWARE MONITORING The nominal low supply voltage threshold is set internally to 4 65V ADM8691 and ADM800L or 4 4V ADM8693 and ADM800M If Vcc falls below these values RESET will be asserted MICROPROCESSOR SUPERVISORY FUNCTIONS ADM8691 SERIES Low Microprocessor Supply Voltage 4 65V or 4 4V Battery Backup Steering Switch Power Failure Monitor Low Line Voltage at Regulator Input Power On Power Down Brownout Reset Watchdog Timer E Inhibit Chip Enable to CMOS Memory Prevents out of tolerance Microprocessor Addressing Memory Figure 7 1 ADM8691 SERIES BLOCK DIAGRAM 0 LOW LINE O Vout CHIP ENABLE gt OUTPUT LEM RESET amp WATCHDOG TIMEBASE O RESET O RESET WATCHDOG WATCHDOG WATCHDOG WATCHDOG INPUT WDI TRANSITION DETECTOR TIMER O OUT WDO ponet gt gt POWER OUT PFO 1 25V 125v GND Figure 7 2 7 2 HARDWARE MONITORING APPLICATION OF ADM8691 SERIES LINE INPUT VOLTAGE Voc BAT ON Vout C
206. e LDO variety As it turns out almost all LDOs available today as well as many of the more general three terminal regulator types are positive leg series style regulators This simply means that they control the regulated voltage 2 25 REFERENCES AND LOW DROPOUT LINEAR REGULATORS output by means of a pass element which is in series with the positive side of unregulated input This is shown more clearly in Figure 2 23 which is a hookup diagram for a hypothetical three terminal style regulator To re iterate what was said earlier in the chapter about reference ICs in terms of their basic functionality many standard voltage regulator ICs are available in the series three terminal form as is shown here GND or Common Voy7 A BASIC THREE TERMINAL VOLTAGE REGULATOR O 6V Vn Vour 1V Q Vour SV THREE TERMINAL REGULATOR R 50 IGROUND imA Figure 2 23 This diagram also allows some statements to be made about power losses in the regulator There are two components to power which are dissipated in the regulator one a function of VIN Voy and IT plus a second which is a function of VIN and Iground If we call the total power Pp this then becomes Pp VIN VoUT IL VIN Iground Obviously the magnitude of the load current and the regulator dropout voltage both greatly influence the power dissipated However it is also easy to see that for a given Ig as the dr
207. e power twice distribution offers some advantages Since such systems require isolation from the line voltage only the first converter requires the isolation all cascaded converters need not be isolated or at least not to the degree of isolation that the first converter requires The intermediate DC voltage is usually regulated to less than 60 volts in order to minimize the isolation requirement for the cascaded converters Its regulation is not critical since it is not a direct output Since it is typically higher than any of the switching regulator output voltages the distribution current is substantially less than the sum of the output currents thereby reducing I R losses in the system power distribution wiring This also allows the use of a smaller energy storage capacitor on the intermediate DC supply output Recall that the energy stored in a capacitor is YCV2 3 5 SWITCHING REGULATORS Power management can be realized by selectively turning on or off the individual DC DC converters as needed POWER DISTRIBUTION USING LINEAR AND SWITCHING REGULATORS TRADITIONAL USING DISTRIBUTED USING LINEAR REGULATORS SWITCHING REGULATORS RECTIFIER FILTER E OFF LINE SWREG Vpc lt 60V S RECTIFIER LINEAR VN VN AND REG FILTER Figure 3 4 ADVANTAGES OF DISTRIBUTED POWER SYSTEMS USING SWITCHING REGULATORS B Higher Efficiency with Switching Regulators than Linear Regulators Use of High Intermediate DC
208. e relationships signify to maintain a low TJ either 0 or the power dissipated or both must be kept low A low AT is the key to extending semiconductor lifetimes as it leads to low maximum junction temperatures In semiconductors one temperature reference point is always the device junction taken to mean the hottest spot inside the chip operating within a given package The other relevant reference point will be either the case of the device or the ambient temperature T A that of the surrounding air This then leads in turn to the above mentioned individual thermal resistances and 8 45 HARDWARE DESIGN TECHNIQUES THERMAL DESIGN BASICS 6 Thermal Resistance C W AT P x0 TA 9 0jA Junction to Ambient Thermal Resistance Junction to Case Thermal Resistance Case to Ambient Thermal Resistance 9JA Ty TA P x Total Device Power Dissipation TJ Max 150 C Sometimes 175 C Ty Figure 8 43 Taking the more simple case first is the thermal resistance of a given device measured between its junction and the ambient air This thermal resistance is most often used with small relatively low power ICs which do not dissipate serious amounts of power that is 1W or less 0JA figures typical of op amps and other small devices are on the order of 90 100 C W for a plastic 8 pin DIP package It must be understood that thermal resistan
209. easurement and limit comparison of up to four power supplies and two processor core voltages plus temperature fan speed and chassis intrusion Measured values can be read out via an I2C compatible serial interface and values for limit comparisons can be programmed over the same serial bus The high speed 10 bit ADC allows frequent sampling of all analog channels to ensure a fast interrupt response to any out of limit measurement Key specifications for the ADM9240 are summarized in Figure 7 13 ADM9240 ADC BASED HARDWARE MONITOR NTEST OUT A0 O 1 SERIAL BUS O INTERFACE VID lt lt ________ __ gt REGISTER O 2 sce O FAN SPEED COUNTER FAN2 O LIMIT REGISTERS ANDCOMPARATORS BS O INTERRUPT STATUS REGISTERS 10 BIT O 2 5V INT AND MASK O 43 3V vcc O REGISTERS lt VALUE RAM CONFIGURATION 1 REGISTER lt gt Uu Q 12V NTEST_IN AOUT O 8 BIT DAC O VCCP2 BANDGAP O GNDA RESET O CHASSIS INT TEMP CLEAR REGISTER SENSOR WY Figure 7 12 7 9 HARDWARE MONITORING ADM9240 KEY SPECIFICATIONS 6 Direct Voltage Measurement Inputs Including 2 Processor Core Voltages with On Chip Attenuators On Chip 10 bit ADC and 8 bit DAC 5 Digital Voltage Identification VID Inputs 2 Fan Speed Monitoring Inputs I2C Compatible System Management Bus Chassis Intrusion Detect On Chip Temperature Sensor Vec 2
210. ector 5 21 lithium metal 5 3 5 6 lithium ion 5 3 4 5 6 5 8 10 charge termination 5 24 charger end of charge detect 5 20 22 linear 5 17 18 switch mode dual 5 18 22 universal 5 22 24 memory 5 4 NiCd 5 3 4 5 6 charger switch mode dual 5 18 22 universal 5 22 24 nickel metal hydride 5 3 4 5 6 charger switch mode dual 5 18 22 universal 5 22 24 overcharging 5 1 rechargeable 5 1 figures of merit 5 2 3 technologies 5 3 sealed lead acid 5 3 4 5 6 self discharge 5 2 switching regulator 3 6 temperature charge termination method 5 7 voltage charge termination method 5 7 Billings Keith 3 69 Bird s nest prototyping 8 4 Blattner Rob 2 57 8 58 Bleaney B 8 87 Bleaney B I 8 87 Blood William R Jr 8 87 Boltzmann s constant 6 19 Boost converter basic circuit 3 16 discontinuous operation point 3 19 20 gated oscillator inductance calculation 3 50 51 ideal 3 15 20 input output current ripple current rating 3 65 66 waveforms 3 59 60 Index 4 input output relationship 3 17 negative in negative out 3 20 power MOSFET switches 3 39 3 41 pulse burst modulation inductance calculation 3 50 51 pulse wave modulation constant frequency inductance calculation 3 54 55 waveforms 3 16 discontinuous mode 3 18 Boost switched capacitor voltage regulator 4 18 Boyle 8 13 Brokaw cell 2 6 2 9 6 20 21 bandgap reference 2 32 Brokaw Paul 2 24 2 57 6 38 8 77 8 87 Brown Marty 3 6
211. ectronics Box 5828 Greenville SC 29606 803 963 6300 Inductor Manufacturers Coiltronics 6000 Park of Commerce Blvd Boca Raton FL 33487 407 241 7876 Sumida 5999 New Wilke Rd Suite 110 Rolling Meadow IL 60008 847 956 0666 Pulse Engineering 12220 World Trade Drive San Diego CA 92128 619 674 8100 Gowanda Electronics 1 Industrial Place Gowanda NY 14070 716 532 2234 Coilcraft 1102 Silver Lake Rd Cary IL 60013 847 639 2361 Dale Electronics Inc E Highway 50 P O Box 180 Yankton SD 57078 605 665 9301 Hurricane Electronics Lab 331 N 2260 West P O Box 1280 Hurricane UT 84737 801 635 2003 Core Manufacturers Magnetics P O Box 391 Butler PA 16003 412 282 8282 MOSFET Manufacturers International Rectifier 233 Kansas Street El Segundo CA 90245 310 322 3331 Motorola Semiconductor 3102 North 56 Street MS56 126 Phoenix AZ 85018 800 521 6274 Siliconix Inc 2201 Laurelwood Road P O Box 54951 Santa Clara CA 95056 408 988 8000 30 31 32 SWITCHING REGULATORS Schottky Diode Manufacturers General Instrument Power Semiconductor Division 10 Melville Park Road Melville NY 11747 516 847 3000 International Rectifier 233 Kansas Street El Segundo CA 90245 310 322 3331 Motorola Semiconductor 3102 North 56 Street MS56 126 Phoenix AZ 85018 800 521 6274 3 71 SWITCHED CAPACITOR VOLTAGE CONVERTERS SECTION 4 SWITCHED CAPACITOR VOLTAG
212. ed schematic of a flyback battery charger with isolation provided by the flyback transformer and the optocoupler For details of the schematic refer to the ADP3810 3811 data sheet Reference 7 Caution This circuit contains lethal AC and DC voltages and appropriate precautions must be observed Please refer to the data sheet text and schematic if building this circuit The operation of the circuit is similar to that of Figure 5 16 The DC DC converter block is comprised of a primary side PWM circuit and flyback transformer and the control signal passes through the optocoupler to the PWM ADP3810 OFF LINE FLYBACK BATTERY CHARGER Voc VSENSE CHARGE CURRENT our ADP3810 8 4 VOLTAGE dmm CONTROL GND kk WARNING LETHAL VOLTAGES PRESENT USE EXTREME CAUTION Figure 5 17 A typical current mode flyback PWM controller 3845 series was chosen for the primary control for several reasons First and most importantly it is capable of operating from very small duty cycles to near the maximum desired duty cycle This makes it a good choice for a wide input AC supply voltage variation requirement which is usually between 70V and 270V for world wide applications Add to that the additional requirement of 0 to 100 current control and the PWM duty cycle must 5 14 BATTERY CHARGERS have a wide range This charger achieves these ranges while maintaining stable feedback loops The detailed operation
213. ed voltage VSEgNSE is pulled high causing GM2 to source more current thereby increasing IOUT As with the current loop the higher IQ jt reduces the duty cycle of the DC DC converter and causes the battery voltage to fall balancing the feedback loop Notice that because of the low side sensing scheme the ground of the circuits in the system must be isolated from the ground of the DC DC converter Further design details for specific applications are given in the ADP3810 3811 data sheet Reference 7 including detailed analysis and computations for compensating the feedback loops with resistor and capacitor Cc The ADP3810 3811 does not include circuitry to detect charge termination criteria such as AV or dT dt which are common for NiCd and NiMH batteries If such charge termination schemes are required a low cost microcontroller can be added to the system to monitor the battery voltage and temperature A PWM output from the microcontroller can subsequently program the V vTRI input to set the charge current The high impedance of enables the addition of an RC filter to integrate the PWM output into a DC control voltage 5 13 BATTERY CHARGERS OFF LINE ISOLATED FLYBACK BATTERY CHARGER The ADP3810 3811 are ideal for use in isolated off line chargers Because the output stage can directly drive an optocoupler feedback of the control signal across an isolation barrier is a simple task Figure 5 17 shows a simplifi
214. eform for a buck PWM regulator operating in the continuous mode It is accepted design practice to design for a peak to peak ripple current which is between 10 and 30 of the output current IOUT We will assume that Ip 0 2IQUT 3 51 SWITCHING REGULATORS CALCULATING L FOR BUCK CONVERTER CONSTANT FREQUENCY PWM TYPE OUTPUT AND INDUCTOR CURRENT CONTINUOUS MODE VOUT Vp saco otio E L Vin Vour Vsw L ton toft 7 VIN Vout Vsw Y VOUT fA VIN Vsw Vp Ipp NOMINALLY MAKE I pp 0 2 loy Figure 3 52 By inspection we can write m Vout Vsw Jon E our Vp L L Jor or tore JIN VOUT VSW VouT Vp Jj However the switching frequency f is given by 1 E ms ton am Bose Substituting this expression for tofr in the previous equation for and solving for ton yields t 1 VOUT VD f Vin Vsw Vp J However VIN VOUT VW pp L on 3 52 SWITCHING REGULATORS Combining the last two equations and solving for L yields L Gl VIN VoUT VSW VOUT VD Lfor buck PWM converter constant frequency As indicated earlier choose Ip to be nominally 0 2IQUT and solve the equation for L Calculate L for the minimum and maximum expected value of VIN and choose value halfway between System requirements may dictate a larger or smaller value of Ipp which will inv
215. egative supply lines To help reduce the emissions generated by extremely fast moving digital signals at DAC inputs or ADC outputs a small resistor or ferrite bead may be required at each digital input output 8 73 HARDWARE DESIGN TECHNIQUES POWER SUPPLY FILTERING AND SIGNAL LINE SNUBBING GREATLY REDUCES EMI EMISSIONS Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA FERRITE BEAD tee BEAD a s 10 330 RESISTOR ME 7 1 dis MICROPROCESSOR OR OTHER HIGH SPEED CLOCKED CIRCUIT Figure 8 71 Once the system s critical paths and circuits have been identified the next step in implementing sound PCB layout is to partition the printed circuit board according to circuit function This involves the appropriate use of power ground and signal planes Good PCB layouts also isolate critical analog paths from sources of high interference I O lines and connectors for example High frequency circuits analog and digital should be separated from low frequency ones Furthermore automatic signal routing CAD layout software should be used with extreme caution and critical paths routed by hand Properly designed multilayer printed circuit boards can reduce EMI emissions and increase immunity to RF fields by a factor of 10 or more compared to double sided boards A multilayer board allows a complete layer to be used for the ground pl
216. egrade the performance of the pin socket and pin sockets should never be used in the high current paths associated with linear or switching regulators The uncapped versions allow the IC pins to extend out the bottom of the socket After the prototype is functional and no further changes are to be made the IC pins can be soldered directly to the bottom of the socket thereby making a permanent and rugged connection 8 8 HARDWARE DESIGN TECHNIQUES PIN SOCKETS CAGE JACKS HAVE MINIMUM PARASITIC RESISTANCE INDUCTANCE AND CAPACITANCE SPRING SOLDER CONTACTS COPPER SOLDER PCB DIELECTRIC PCB DIELECTRIC 7 SOLDER SOLDER core OR UNCAPPED VERSIONS AVAILABLE Figure 8 6 The prototyping techniques discussed so far have been limited to single or double sided PC boards Multilayer PC boards do not easily lend themselves to standard prototyping techniques If multilayer board prototyping is required one side of a double sided board can be used for ground and the other side for power and signals Point to point wiring can be used for additional runs which would normally be placed on the additional layers provided by a multi layer board However it is difficult to control the impedance of the point to point wiring runs and the high frequency performance of a circuit prototyped in this manner may differ significantly from the final multilayer board
217. ence Figure 5 25 shows a simplified block diagram for a universal charger using a microcontroller with the ADP3801 5 22 BATTERY CHARGERS UNIVERSAL BATTERY CHARGER SIMPLIFIED O Vin BATA MICRO ADP3801 3802 VL CONTROLLER CHARGER CIRCUIT ah TN BATpnc SD EOC GND T BATTERY THERMISTOR SEE DATA SHEET FOR DETAILS NOTE PWM COMPONENTS AND CONNECTIONS NOT SHOWN Figure 5 25 The microcontroller is used to monitor the battery voltage and temperature via its internal 8 bit ADC and multiplexer input It also keeps track of the overall charge time It may also monitor the ambient temperature via a thermistor or an analog temp sensor The ADP3801 s LDO makes an ideal supply for the microcontroller and the RESET pin generates the necessary power on reset signal The LDO can also be used as a 1 reference for the ADC The first step when a battery is inserted into the charger is to identify the type of battery placed in the charger The most common method of doing this is reading the value of the in pack thermistor Different values of thermistors are used to identify if the battery is Li Ion or if it is NICd NiMH This thermistor is also used to monitor the temperature of the battery A resistor from the ADP3801 s LDO to the battery s thermistor terminal forms a resistor divider and generates a voltage across the thermistor for the microcontroller to read During this time the ADP3801 should be in shutdown which the
218. ency interference to ground The best shield can be compromised by poor connection techniques Shields often use pig tail connections to make the connection to ground A pig tail connection is a single wire connection from shield to either chassis or circuit ground This type of connection is inexpensive but at high frequency it does not provide low impedance Quality shields do not leave large gaps in the cable instrument shielding system Shield gaps provide paths for high frequency EMI to enter the system The cable shielding system should include the cable end connectors Ideally cable shield connectors should make 360 contact with the chassis ground As shown in Figure 8 79 pigtail terminations on cables very often cause systems to fail radiated emissions tests because high frequency noise has coupled into the cable shield generally through stray capacitance If the length of the cable is considered electrically long at the interference frequency then it can behave as a very efficient quarter wave antenna The cable pigtail forms a matching network as shown in the s to radiate the noise which coupled into the shield In general pigtails are only recommended for applications below 10kHz such as 50 60Hz interference protection For applications where the interference is greater than 10kHz shielded connectors electrically and physically connected to the chassis should be used 8 84 HARDWARE DESIGN TECHNIQUES SHIE
219. ens of uA and strongly dependent upon design specifics A subset of the general electrolytic family includes tantalum types generally limited to voltages of 100V or less with capacitance of 500uF or less Reference 8 In a given size tantalums exhibit a higher capacitance to volume ratio than do general purpose electrolytics and have both a higher frequency range and lower ESR They are generally more expensive than standard electrolytics and must be carefully applied with respect to surge and ripple currents A subset of aluminum electrolytic capacitors is the switching type designed for handling high pulse currents at frequencies up to several hundred kHz with low losses Reference 9 This capacitor type competes directly with tantalums in high frequency filtering applications with the advantage of a broader range of values A more specialized high performance aluminum electrolytic capacitor type uses an organic semiconductor electrolyte Reference 10 The OS CON capacitors feature appreciably lower ESR and higher frequency range than do other electrolytic types with an additional feature of low low temperature ESR degradation 3 63 SWITCHING REGULATORS Film capacitors are available in very broad value ranges and different dielectrics including polyester polycarbonate polypropylene and polystyrene Because of the low dielectric constant of these films their volumetric efficiency is quite low and a 10uF 50V polyester capa
220. ere turns per meter or oersteds and is proportional to the current flowing in the wire The magnetic field strength produces a magnetic flux density B measured in webers per square meter or gauss 3 55 SWITCHING REGULATORS Using a number of turns of wire to form a coil increases the magnetic flux density for a given current The effective inductance of the coil is proportional to the ratio of the magnetic flux density to the field strength This simple air core inductor is not very practical for the values of inductance required in switching regulators because of wiring resistance interwinding capacitance sheer physical size and other factors Therefore in order to make a reasonable inductor the wire is wound around some type of ferromagnetic core having a high permeability Core permeability is often specified as a relative permeability which is basically the increase in inductance which is obtained when the inductor is wound on a core instead of just air A relative permeability of 1000 for instance will increase inductance by 1000 1 above that of an equivalent air core Figure 3 55 shows magnetic flux density B versus inductor current for the air core and also ferromagnetic cores Note that B is linear with respect to H for the air core inductor i e the inductance remains constant regardless of current MAGNETIC FLUX DENSITY VERSUS INDUCTOR CURRENT a 2 NO AIRGAP LARGE AIRGAP MAGNETIC FLUX DENSI
221. ere will be a net e m f in the circuit and a current will flow determined by the e m f and the total resistance in the circuit Figure 6 6B If we break one of the wires the voltage across the break will be 6 5 TEMPERATURE SENSORS equal to the net thermoelectric e m f of the circuit and if we measure this voltage we can use it to calculate the temperature difference between the two junctions Figure 6 6C We must always remember that a thermocouple measures the temperature difference between two junctions not the absolute temperature at one junction We can only measure the temperature at the measuring junction if we know the temperature of the other junction often called the reference junction or the cold junction But it is not so easy to measure the voltage generated by a thermocouple Suppose that we attach a voltmeter to the circuit in Figure 6 6C Figure 6 6D The wires attached to the voltmeter will form further thermojunctions where they are attached If both these additional junctions are at the same temperature it does not matter what temperature then the Law of Intermediate Metals states that they will make no net contribution to the total e m f of the system If they are at different temperatures they will introduce errors Since every pair of dissimilar metals in contact generates a thermoelectric e m f including copper solder kovar copper kovar is the alloy used for IC leadframes and aluminum kovar at
222. ersely affect the inductor value A variation of the buck PWM constant frequency regulator is the buck PWM regulator with variable frequency and constant off time e g ADP1147 ADP1148 A diagram of the output and inductor current waveform is shown in Figure 3 53 for the continuous mode CALCULATING L FOR BUCK CONVERTER CONSTANT OFF TIME VARIABLE FREQUENCY PWM TYPE OUTPUT AND INDUCTOR CURRENT CONTINUOUS MODE VOUT Vp OPEP S Vin VoUT Vsw ER 2 A L our Vp L V V L lt OUT D hott Ipp NOMINALLY MAKE I pp 0 2 lout Figure 3 53 The calculations are very straightforward since the peak to peak amplitude of the ripple current is constant _ Voum VD Ipp Sour toff 3 53 SWITCHING REGULATORS Solving for L L Sour Jae L for buck PWM constant off time lpp variable frequency converter Again choose Ipp 0 2IQUT or whatever the system requires The final example showing the inductance calculation is for the boost PWM constant frequency regulator The inductor and input current waveform is shown in Figure 3 54 CALCULATING L FOR BOOST CONVERTER CONSTANT FREQUENCY PWM TYPE INPUT AND INDUCTOR CURRENT CONTINUOUS MODE VOUT VD gt D a VOUT VIN as fA VouT Vsw Vp NOMINALLY MAKE I pp 0 2 IN Figure 3 54 The analysis is similar to that of the constant frequency buck PWM
223. es around 100nV VHz so additional filtering is obviously required in most high resolution systems especially those with low values of VREF REFERENCE NOISE REQUIREMENTS FOR VARIOUS SYSTEM ACCURACIES 1 2 LSB 100kHZ CRITERIA NOISE DENSITY nV 4Hz FOR 10 5 AND 2 5V FULLSCALE RANGES BITS 10V 5V 2 5V 12 643 322 161 13 322 161 80 14 161 80 40 15 80 40 20 16 40 20 10 Figure 2 15 2 17 REFERENCES AND LOW DROPOUT LINEAR REGULATORS Some references for example the AD587 buried zener type have a pin designated as the noise reduction pin see data sheet This pin is connected to a high impedance node preceding the on chip buffer amplifier Thus an externally connected capacitor Cy will form a low pass filter with an internal resistor to limits the effective noise bandwidth seen at the output 1pF capacitor gives a dB bandwidth of 40 Hz Note that this method of noise reduction is by no means universal and other devices may implement noise reduction differently if at all There are also general purpose methods of noise reduction which can be used to reduce the noise of any reference IC at any standard voltage level The reference circuit of Figure 2 16 References 5 and 6 is one such example This circuit uses external filtering and a precision low noise op amp to provide both very low noise and high DC accuracy Reference U1 is a 2 5 3 0 5 or 10V reference with a low noise bu
224. es discussed above 2 56 REFERENCES AND LOW DROPOUT LINEAR REGULATORS REFERENCES Low Dropout Regulators 1 10 11 12 13 Bob Widlar New Developments in IC Voltage Regulators IEEE Journal of Solid State Circuits Vol SC 6 February 1971 Robert C Dobkin 3 Terminal Regulator is Adjustable National Semiconductor AN 181 March 1977 Paul Brokaw A Simple Three Terminal IC Bandgap Voltage Reference IEEE Journal of Solid State Circuits Vol SC 9 December 1974 Frank Goodenough Linear Regulator Cuts Dropout Voltage Electronic Design April 16 1987 Chester Simpson LDO Regulators Require Proper Compensation Electronic Design November 4 1996 Frank Goodenough Vertical PNP Based Monolithic LDO Regulator Sports Advanced Features Electronic Design May 13 1996 Frank Goodenough Low Dropout Regulators Get Application Specific Electronic Design May 13 1996 Jim Solomon The Monolithic Op Amp A Tutorial Study IEEE Journal of Solid State Circuits Vol SC 9 No 6 December 1974 Richard J Reay Gregory T A Kovacs An Unconditionally Stable Two Stage CMOS Amplifier IEEE Journal of Solid State Circuits Vol SC 30 No 5 May 1995 NDP6020P NDB6020P P Channel Logic Level Enhancement Mode Field Effect Transistor Fairchild Semiconductor data sheet September 1997 http www fairchildsemi com Alan Li et all Maximum Power Enhancement Techniques for SO 8 Power MOSFETs Fair
225. es is 10 2 Typical accuracies are 1 C at 25 C and 2 C over the 40 C to 125 C range The ADT45 provides a 250mV output at 25 C and is specified for temperature from 0 C to 100 C The ADT50 provides a 750mV output at 25 C and is specified for temperature from 40 C to 125 C ADT45 ADT50 ABSOLUTE VOLTAGE OUTPUT SENSORS 2 7V TO 12V Vout 0 1 SOT 23 B Vout ADT45 250mV 25 C 10mV C Scale Factor ADT50 750mV 25 C 10mV C Scale Factor B 2 C Error Over Temp Typical 0 5 C Non Linearity Typical E Specified 40 to 125 C 60pA Quiescent Current Figure 6 27 If the ADT45 ADT50 sensors are thermally attached and protected they can be used in any temperature measurement application where the maximum temperature range of the medium is between 40 C to 125 C Properly cemented or glued to the surface of the medium these sensors will be within 0 01 C of the surface temperature Caution should be exercised as any wiring to the device can act as heat pipes introducing errors if the surrounding air surface interface is not isothermal Avoiding this condition is easily achieved by dabbing the leads of the sensor and the hookup wires with a bead of thermally conductive epoxy This will ensure that the ADT45 ADT50 die temperature is not affected by the surrounding air temperature 6 24 TEMPERATURE SENSORS In the SOT 23 3 package the thermal resistance jun
226. est rule then is that you should verify what are the specific capacitive loading ground rules for the reference you intend to use for the load conditions your circuit presents VOLTAGE REFERENCE SPECIFICATIONS TOLERANCE It is usually better to select a reference with the required value and accuracy and to avoid external trimming and scaling if possible This allows the best TCs to be realized as tight tolerances and low TCs usually go hand in hand Tolerances as low as 0 04 can be achieved with the AD586 AD780 REF 195 while the 0588 is 0 01 If and when trimming must be used be sure to use the recommended trim network with no more range than is absolutely necessary When if additional external scaling is required a precision op amp should be used along with ratio accurate low TC tracking thin film resistors DRIFT The XFET and buried zener reference families have the best long term drift and TC performance TCs as low as 1 2ppm C are available with the AD586 and AD588 and the AD780 bandgap reference is almost as good at 3ppm C The XFET series achieve long terms drifts of 0 2 ppm 1000 hours while the buried zener types come in at 25ppm 1000 hours Note that where a figure is given for long term drift it is usually drift expressed in ppm 1000 hours There are 8766 hours in a year and many engineers multiply the 1000 hour figure by 8 77 to find the annual drift this is not correct and can in fact be quite pessimistic
227. example a pulse having a 1ns rise time is equivalent to an EMI frequency of over 300MHz This time frequency relationship can also be applied to high speed analog circuits where slew rates in excess of 1000V us and gain bandwidth products greater than 500MHz are not uncommon When this concept is applied to instruments and systems EMI emissions are again functions of signal rise time and pulse repetition rates Spectrum analyzers and high speed oscilloscopes used with voltage and current probes are very useful tools in quantifying the effects of EMI on circuits and systems Another important parameter in the analysis of EMI problems is the physical dimensions of cables wires and enclosures Cables can behave as either passive antennas receptors or very efficient transmitters sources of interference Their physical length and their shield must be carefully examined where EMI is a concern As previously mentioned the behavior of simple conductors is a function of length cross sectional area and frequency Openings in equipment enclosures can behave as slot antennas thereby allowing EMI energy to affect the internal electronics PASSIVE COMPONENTS YOUR ARSENAL AGAINST EMI Minimizing the effects of EMI requires that the circuit system designer be completely aware of the primary arsenal in the battle against interference passive components To use successfully these components the designer must understand their non ideal behavior For exa
228. f 20 C to 50 C Over a 250 C measurement temperature range the thermocouple produces an output voltage change of 10 151mV Since the required circuit s output full scale voltage change is 2 5V the gain of the circuit is set to 246 3 Choosing R4 equal to 4 99kQ sets R5 equal to 1 22MQ Since the closest 1 value for R5 is 1 21MQ a 50kQ potentiometer is used with R5 for fine trim of the full scale output voltage Although the OP193 is a single supply amp its output stage is not rail to rail and will only go down to about 0 1V above ground For this reason R3 is added to the circuit to supply an output offset voltage of about 0 1V for a nominal supply voltage of 5V This offset 10 C must be subtracted when making measurements 6 8 TEMPERATURE SENSORS referenced to the OP193 output R3 also provides an open thermocouple detection forcing the output voltage to greater than 3V should the thermocouple open Resistor R7 balances the DC input impedance of the OP193 and the 0 1 film capacitor reduces noise coupling into its non inverting input USING A TEMPERATURE SENSOR FOR COLD JUNCTION COMPENSATION TMP35 O 3 3V TO 5 5V TYPEK THERMO COUPLE CHROMEL COLD JUNCTION 0 1uF FILM R2 essc NEC 1020 USE 1 RESISTORS 0 lt T lt 250 ALUMEL ISOTHERMAL BLOCK Figure 6 10 The AD594 AD595 is a complete instrumentation amplifier and thermocouple cold junction compensator o
229. f between using a linear regulator as shown versus using a flyback or buck type of charger is efficiency versus simplicity The linear charger of Figure 5 20 is very simple and it uses a minimal amount of external components However the efficiency is poor especially when there is a large difference between the input and output voltages The power loss in the power MOSFET is equal to VIN VBAT ICHARGE Since the circuit is powered from a wall adapter efficiency may not be a big concern but the heat dissipated in the pass transistor could be excessive ADP3820 LINEAR REGULATOR CONTROLLER FOR Li lon BATTERY CHARGING ViN Rs IFR9014 VBAT G OUT Li lon Battery 1pF ADP3820 4 2 10 GND 1 Accuracy over 20 C 85 C 4 2V 4 1V Final Battery Voltage Options Low Quiescent Current Shutdown Current lt 1pA Externally Programmable Current Limit Figure 5 20 SWITCH MODE DUAL CHARGER FOR LI ION NICD AND NIMH BATTERIES The ADP3801 and ADP3802 are complete battery charging ICs with on chip buck regulator control circuits The devices combine a high accuracy final battery voltage control with a constant charge current control and on chip 3 3V Low Drop Out Regulator The accuracy of the final battery voltage control is 0 75 to safely charge Li Ion batteries An internal multiplexer allows the alternate charging of two separate battery stacks The final voltage is pin programmable to one of six options 4 2V one Li I
230. fer lower Roy choice options they also demand a boosted voltage supply to turn on making them impractical for a simple LDO PMOS pass devices are widely available in low both Ron and low threshold forms with current levels up to several amperes They offer the potential of the lowest dropout of any device since dropout can always be lowered by picking a lower Ron part The dropout voltage of lateral PNP pass devices is reasonably good typically around 300mV at 150mA with a maximum of 600mV These levels are however considerably bettered in regulators using vertical PNPs which have a typical of 150 at currents of 200mA This leads directly to an Iground of 1 5mA at the 200mA output current The dropout voltage of vertical PNPs is also an improvement vis vis that of the lateral PNP regulator and is typically 180mV at 200mA with maximum of 400mV There are also major AC performance issues to be dealt with in the LDO architecture of Fig 2 29 This topology has an inherently high output impedance due to the operation of the PNP pass device in a common emitter or common source with a PMOS device mode In either case this factor causes the regulator to appear as a high source impedance to the load The internal compensation capacitor of the regulator Ccomp forms a fixed frequency pole in conjunction with the gm of the error amplifier In addition load capacitance forms an output pole in conjunction with RT This particula
231. ffered output The output of U1 is applied to the R1 C1 C2 noise filter to produce a corner frequency of about 1 7 Hz Electrolytic capacitors usually imply DC leakage errors but the bootstrap connection of C1 causes its applied bias voltage to be only the relatively small drop across R2 This lowers the leakage current through R1 to acceptable levels Since the filter attenuation is modest below a few Hertz the reference noise still affects overall performance at low frequencies i e lt 10 Hz COMBINING LOW NOISE AMPLIFIER WITH EXTENSIVE FILTERING YIELDS EXCEPTIONAL REFERENCE NOISE PERFORMANCE 1 5 TO 5nV VHz 1kHZ 15V 15V C1 100 25 DIODES 1N4148 C2 25V 100yF 25V 10pF U1 AD586 0587 REFO1 U2 OP113 OP27 L xni REF02 REF05 REF10 OP176 AD797 Figure 2 16 The output of the filter is then buffered by a precision low noise unity gain follower such as the OP113EP With less than 150yV of offset error and under 1yV C drift the buffer amplifier s DC performance will not seriously affect the accuracy drift of most references For example an ADR292E for U1 will have a typical drift of 3ppm C equivalent to 7 5uV C higher than the buffer amplifier Almost any op amp will have a current limit higher than a typical IC reference Further even lower noise op amps are available for 5 10V use The AD797 offers 1kHz noise performance less than 2nV VHz in this circuit compared to about 5nV VHz for the
232. filter less catastrophic transients or high frequency interference These EMI filters provide both common mode and differential mode filtering An optional choke in the safety ground can provide additional protection against common mode noise The value of this choke cannot be too large however because its resistance may affect power line fault clearing These filters work in both directions they are not only protect the equipment from surges on the power line but also prevent transients from the internal switching power supplies from corrupting the power line 8 70 HARDWARE DESIGN TECHNIQUES POWER LINE DISTURBANCES CAN GENERATE EMI Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA TRANSIENT GAS DISCHARGE SUPPRESSORS TUBES BIG ZENERS CROWBARS CHOKES OR MOV V e e p O e LINE N S 2 Y YN e o LOAD e TEC X G e e O COMMON MODE AND DIFFERENTIAL MODE PROTECTION Figure 8 67 SCHEMATIC FOR A COMMERCIAL POWER LINE FILTER Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA li LOAD HOT FE ra UU ROT 2 OPTIONAL NOTE OPTIONAL CHOKE ADDED FOR COMMON MODE PROTECTION Figure 8 68 Transformers provide the best common mode power line isolation They provide good protec
233. ge 5 5 6 V Valley control 3 31 Vargha Doug 5 25 Voltage converter high resolution low noise references 2 23 24 switched capacitor 4 1 21 advantages 4 2 3 diagram 4 2 efficiency 4 2 regulated output 4 15 21 steady state 4 7 voltage doubler 4 1 3 voltage inverter 4 1 3 Voltage doubler power losses 4 12 13 waveforms 4 9 11 Voltage inverter power losses diagram 4 11 switched capacitor unregulated 4 8 9 waveforms 4 9 11 Voltage inverter doubler bandgap characteristics 2 13 unregulated 4 13 15 Voltage reference architectures characteristics 2 13 bandgap 2 4 9 characteristics 2 13 buried zener characteristics 2 13 stability 2 10 decoupling 2 14 dynamic load response 2 20 noise 2 1 2 16 19 system requirements 17 trimming 2 23 precision 2 1 2 pulse response circuit 2 21 22 pulse loading conditions 2 2 INDEX selection criteria 2 2 simple diode circuit forward biased 2 3 zener avalanche diode 2 3 specifications 2 14 24 drift 2 14 15 line sensitivity 2 16 load sensitivity 2 15 16 low noise references for high resolution converters 2 23 24 noise 2 16 19 reference pulse current response 2 20 23 scaled references 2 19 20 supply range 2 15 tolerance 2 14 standard positive output three terminal hookup 2 13 14 startup behavior 2 1 temperature drift 2 1 various systems compared 2 15 types 2 2 4 series 2 2 shunt 2 2 three terminal 2 2 2 14 output capac
234. ght general issues of EMC electromagnetic compatibility to familiarize the system circuit designer with this subject and to illustrate proven techniques for protection against EMI A PRIMER ON EMI REGULATIONS The intent of this section is to summarize the different types of electromagnetic compatibility EMC regulations imposed on equipment manufacturers both voluntary and mandatory Published EMC regulations apply at this time only to equipment and systems and not to components Thus EMI hardened equipment does not necessarily imply that each of the components used integrated circuits especially in the equipment must also be EMI hardened Commercial Equipment The two driving forces behind commercial EMI regulations are the FCC Federal Communications Commission in the U S and the VDE Verband Deutscher Electrotechniker in Germany VDE regulations are more restrictive than the FCC s with regard to emissions and radiation but the European Community will be adding immunity to RF electrostatic discharge and power line disturbances to the VDE regulations and now requires mandatory compliance In Japan commercial EMC regulations are covered under the VCCI Voluntary Control Council for Interference standards and implied by the name are much looser than their FCC and VDE counterparts All commercial EMI regulations primarily focus on radiated emissions specifically to protect nearby radio and television receivers although both
235. gnates the SOT23 6 lead package The example circuit shown produces 5 0V with the use of the ADP3300 5 In operation the circuit will produce its rated 5V output for loads of 50mA or less and for input voltages above 5 3V Vo iT 0 3V when the shutdown input is in a HIGH state This can be accomplished either by a logic HIGH control input to the SD pin or by simply tying this pin to VrN When SD is LOW or tied to ground the regulator shuts down and draws a quiescent current of 14A or less 2 43 REFERENCES AND LOW DROPOUT LINEAR REGULATORS A BASIC ADP3300 50mA LDO REGULATOR CIRCUIT ADP3300 5 SD GND ON OFF Figure 2 37 The ADP3300 and other anyCAP series devices maintain regulation over a wide range of load input voltage and temperature conditions However when the regulator is overloaded or entering the dropout region for example by a reduction the input voltage the open collector ERR pin becomes active by going to LOW or conducting state Once set the ERR pin s internal hysteresis keeps the output low until some margin of operating range is restored In the circuit of Fig 2 37 R1 is a pullup resistor for the ERR output EUT This resistor can be eliminated if the load being driven provides a pullup current The ERR function can also be activated by the regulator s over temperature protection circuit which trips at 165 C These internal current and thermal limits are intended
236. gure 3 16 the basic relationship between the input and output voltage may be derived by inspecting the inductor current waveform and writing TN iu c OUT YIN See L Solving for VOUT ton toff V e toff 2 1 D VOUT VIN INPUT OUTPUT RELATIONSHIP FOR BOOST CONVERTER ton loft ton Write by Inspection from Inductor Input Current Waveforms toff VI Vi IN eton our IN L E Rearrange and Solve for ton t 1 V Vin e ON off _ Vin OUT IN toff IN 4 Figure 3 16 3 17 SWITCHING REGULATORS This discussion so far has assumed the boost converter is in the continuous mode of operation defined by the fact that the inductor current never goes to zero If however the output load current is decreased there comes a point where the inductor current will go to zero between cycles and the inductor current is said to be discontinuous It is necessary to understand this operating mode as well since many switchers must supply a wide dynamic range of output current where this phenomenon is unavoidable Discontinuous operation for the boost converter is similar to that of the buck converter Figure 3 17 shows the waveforms Note that when the inductor current goes to zero ringing occurs at the switch node at a frequency fp given by 1 2n JL Cp Cow fo BOOST CONVERTER WAVEFORMS DISCONTINUOUS MODE iin iL ip lour A 1 1 V IN OUT X
237. hange vs time temperature change temperature change vs time minimum current at full voltage charge time or various combinations of the above Battery voltage and temperature are the most popular methods of terminating the charge of NiCd and NiMH batteries Figure 5 9 shows the cell voltage and temperature as a function of charge time for these two types of batteries charging at the 1C rate Note that NiCd has a distinct peak in the cell voltage immediately preceding full charge NiMH has a much less pronounced peak as shown in the dotted portion of the curve A popular method of charge termination for NiCd is the AV method where the charge is terminated after the cell voltage falls 10 to 20mV after reaching its peak Note that for both types the temperature increases rather suddenly near full charge Because of the much less pronounced voltage peak in the NiMH characteristic the change in temperature with respect to time dT dt is most often used as a primary charge termination method NiCd NiMH BATTERY TEMPERATURE AND VOLTAGE CHARGING CHARACTERISTICS dV dt 0 EN Fail Safe CELL V CELL VOLTAGE dT dt Threshold CELL T CELL TEMP 7 Approx Time to charge ae Fail Safe Charge Time TIME Figure 5 9 In addition to the primary termination secondary terminations are used as backups for added protection The primary and secondary termination methods for NiCd and NiMH cells are summarized in Figure 5
238. he circuit or system designer must contend The first type of interference is that generated by and emitted from an instrument this is known as circuit system emission and can be either conducted or radiated An example of this would be the personal computer Portable and desktop computers must pass the stringent FCC Part 15 specifications prior to general use 8 62 HARDWARE DESIGN TECHNIQUES THREE TYPES OF INTERFERENCE EMISSIONS IMMUNITY INTERNAL Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA HANDHELD TRANSMITTER RADIO RADIATED TRANSMITTER EMISSIONS INTERNAL ELECTRONICS LIGHTNING HUMAN ESD EMISSIONS POWER DISTURBANCE Figure 8 60 The second type of interference is circuit or system immunity This describes the behavior of an instrument when it is exposed to large electromagnetic fields primarily electric fields with an intensity in the range of 1 to 10V m at a distance of 3 meters Another term for immunity is susceptibility and it describes circuit system behavior against radiated or conducted interference The third type of interference is internal Although not directly shown on the figure internal interference can be high speed digital circuitry within the equipment which affects sensitive analog or other digital circuitry or noisy power supplies which can contaminate both analog and digital circuits Internal
239. he SD pin Li Ion manufacturers recommend that the battery should not be left in trickle charge mode indefinitely Thus the ADP3801 3802 EOC signal makes the charger design simpler Periodically the system can remove the SD signal wait until the switching regulator output settles check the status of the EOC signal and then decide to resume charging if necessary This operation maintains a fully charged battery without having to resort to trickle charging The output stage of the ADP3801 3802 is designed to directly drive an external PMOS transistor Some discrete logic level PMOS transistors have a low Vag breakdown voltage specification To prevent damage the output swing is limited to approximately 8V below VCC For further details on specific design issues consult the ADP3801 3802 product data sheet Reference 9 UNIVERSAL CHARGER FOR LI ION NICD AND NIMH Many applications only require the charger to charge one specific battery The form factor physical dimensions of the battery pack is usually unique to prevent other battery types from being plugged in However some applications require the charger to handle multiple battery types and chemistries The design for these universal chargers is fairly complicated because the charger must first identify the type of battery program the charge current and voltage and choose the proper charge termination scheme Clearly such a charger requires some sort of microcontroller intellig
240. he energy could only be transferred from the higher to the lower value source In contrast an inductor ideally returns all the energy that 3 8 SWITCHING REGULATORS is stored in it and with the use of properly configured switches the energy can flow from any one source to another regardless of their respective values and polarities ENERGY TRANSFER USING AN INDUCTOR 1 E So m L SLOPE Figure 3 8 When the switches are initially placed in the position shown the voltage is applied to the inductor and the inductor current builds up at a rate equal to V1 L The peak value of the inductor current at the end of the interval t1 is V IPEAK T ety The average power transferred to the inductor during the interval t4 is 1 PAVG 5 IPEAK The energy transferred during the interval t4 is 1 E FAva tti IPRAK V1 etr Solving the first equation for t1 and substituting into the last equation yields E L IpgAK 3 9 SWITCHING REGULATORS When the switch positions are reversed the inductor current continues to flow into the load voltage Vo and the inductor current decreases at a rate Vo L At the end of the interval to the inductor current has decreased to zero and the energy has been transferred into the load The figure shows the current waveforms for the inductor the input current 11 and the output current io The ideal inductor dissipates no power so there is no power loss in this tr
241. hows how the Seebeck coefficient the change of output voltage with change of sensor junction temperature i e the first derivative of output with respect to temperature varies with sensor junction temperature we are still considering the case where the reference junction is maintained at 0 C When selecting a thermocouple for making measurements over a particular range of temperature we should choose a thermocouple whose Seebeck coefficient varies as little as possible over that range 6 3 TEMPERATURE SENSORS THERMOCOUPLE OUTPUT VOLTAGES FOR TYPE J K AND S THERMOCOUPLES 60 50 40 30 THERMOCOUPLE OUTPUT VOLTAGE mV 10 250 0 250 500 750 1000 1250 1500 1750 TEMPERATURE C Figure 6 4 THERMOCOUPLE SEEBECK COEFFICIENT _ VERSUS TEMPERATURE 70 60 50 40 Deest ee TM 2 ne 6 SEEBECK COEFFICIENT uV C 0 A B 250 0 250 500 750 1000 1250 1500 1750 TEMPERATURE C Figure 6 5 6 4 TEMPERATURE SENSORS For example a Type J thermocouple has a Seebeck coefficient which varies by less than 1pV C between 200 and 500 C which makes it ideal for measurements in this range Presenting these data on thermocouples serves two purposes First Figure 6 4 illustrates the range and sensitivity of the three thermocouple types so that the system designer can at a glance determine that a Type S thermocouple has the widest useful temperature range b
242. ically built of a platinum Pt wire wrapped around a ceramic bobbin the RTD exhibits behavior which is more accurate and more linear over wide temperature ranges than a thermocouple Figure 6 13 illustrates the temperature coefficient of a 1000 RTD and the Seebeck coefficient of a Type S thermocouple Over the entire range approximately 200 C to 850 C the RTD is a more linear device Hence linearizing an RTD is less complex 6 11 TEMPERATURE SENSORS RESISTANCE TEMPERATURE DETECTORs RTD Platinum Pt the Most Common 1000 10000 Standard Values Typical TC 0 385 0 3850 C for 1000 Pt RTD Good Linearity Better than Thermocouple Easily Compensated 0 400 11 5 RTD 4000PtRTD gt TYPES 1000 PtRTD RESISTANCE 2 0 10 5 THERMOCOUPLE 0 375 2 lt 6 TC C THERMOCOUPLE SEEBECK gt lt 9 50 0 350 24 4 8 50 0 325 7 50 0 300 6 50 0 275 0 400 800 TEMPERATURE C Figure 6 13 Unlike a thermocouple however an RTD is a passive sensor and requires current excitation to produce an output voltage The RTD s low temperature coefficient of 0 385 C requires similar high performance signal conditioning circuitry to that used by a thermocouple however the voltage drop across an RTD is much larger than a thermocouple output voltage A system designer may opt for large value RTDs with higher output but la
243. iety of low inductance terminations to minimize ESL Reference 12 Dependent upon their electrical and physical size film capacitors can be useful at frequencies to well above 10MHz At the highest frequencies only stacked film types should be considered Some manufacturers are now supplying film types in leadless surface mount packages which eliminates the lead length inductance Ceramic is often the capacitor material of choice above a few MHz due to its compact size low loss and availability up to several uF in the high K dielectric formulations X7R and 250 at voltage ratings up to 200V see ceramic families of Reference 8 NPO also called COG types use a lower dielectric constant formulation and have nominally zero TC plus a low voltage coefficient unlike the less stable high K types NPO types are limited to values of 0 1uF or less with 0 01uF representing a more practical upper limit Multilayer ceramic chip caps are very popular for bypassing filtering at 10MHz or more simply because their very low inductance design allows near optimum RF bypassing For smaller values ceramic chip caps have an operating frequency range to IGHz For high frequency applications a useful selection can be ensured by selecting a value which has a self resonant frequency above the highest frequency of interest The ripple current rating of electrolytic capacitors must not be ignored in switching regulator applications because unlike line
244. igital temperature sensors 6 36 37 temperature sensor 10 bit ADC characteristics 6 37 AD77XX high resolution ADC 6 13 15 high resolution ADCs 6 11 AD7705 16 bit sigma delta ADC 7 11 applications 7 13 battery monitor circuit 7 13 cell monitor battery charger 7 13 programmable gain amplifier 7 13 specifications 7 13 AD7817 7818 7819 digital temperature sensors 6 386 37 temperature sensor 10 bit ADC serial interface characteristics 6 37 AD22103 ratiometric output sensor 6 22 23 ADC 10 bit series temperature sensor 6 386 37 16 bit sigma delta 7 11 22 bit 2 23 high resolution 6 13 15 resistance temperature detector interfacing 6 15 high speed EMI RFI noise 8 73 on chip temperature sensors 6 386 37 sigma delta AD780 driven 2 23 24 internal digital filter 2 21 noise 2 20 reference input 2 20 21 switched capacitor input 2 20 21 successive approximation reference bypass capacitors 2 22 23 ADM660 efficiency 4 15 specifications 4 14 switched capacitor voltage converter 4 14 switched capacitor voltage inverter doubler 4 13 15 ADM8660 efficiency 4 15 specifications 4 14 switched capacitor voltage inverter doubler 4 13 15 ADM8691 application 7 1 3 block diagram 7 2 Chip Enable output 7 3 functionality 7 1 supervisory products 7 4 watchdog input 7 3 ADM9240 Index 1 INDEX block diagram 7 9 generic application circuit 7 11 microprocessor monitoring 7 9 specifications 7
245. ilability up to several uF in the high K dielectric formulations X7R 250 at voltage ratings up to 200V see ceramic families of Reference 3 NPO also called COG types use a lower dielectric constant formulation and have nominally zero TC plus a low voltage coefficient unlike the less stable high K types NPO types are limited to values of 0 1uF or less with 0 01uF representing a more practical upper limit Multilayer ceramic chip caps are very popular for bypassing filtering at 10MHz or more simply because their very low inductance design allows near optimum RF bypassing For smaller values ceramic chip caps have an operating frequency range to IGHz For high frequency applications a useful selection can be ensured by selecting a value which has a self resonant frequency above the highest frequency of interest All capacitors have some finite ESR In some cases the ESR may actually be helpful in reducing resonance peaks in filters by supplying free damping For example in most electrolytic types a nominally flat broad series resonance region can be noted an impedance vs frequency plot This occurs where Z falls to a minimum level nominally equal to the capacitor s ESR at that frequency This low Q resonance can generally be noted to cover a relatively wide frequency range of several octaves Contrasted to the very high Q sharp resonances of film and ceramic caps the low Q behavior of electrolytics can be
246. ill that modulation method tends to allow for less ripple current variation than does fixed frequency so it is often used In the case where very low duty cycles are needed e g under short circuit conditions sometimes the limitation of a minimum achievable duty cycle is encountered In such cases in order to maintain a steady state condition and prevent runaway of the switch current a pulse skipping function must be implemented This might take the form of a current monitoring circuit which detects that the switch current is too high to turn the switch on and ramp the current up any higher So either a fixed frequency cycle is skipped without turning on the switch or the off time is extended in some way to delay the turn on The pulse skipping technique for a fixed frequency controller can be applied even to operation at normal duty cycles Such a switch modulation technique is then referred to as pulse burst modulation PBM At its simplest this technique simply gates a fixed frequency fixed duty cycle oscillator to be applied to the switch or not The duty cycle of the oscillator sets the maximum achievable duty cycle for the converter and smaller duty cycles are achieved over an average of a multiplicity of pulses by skipping oscillator cycles This switch modulation method accompanies a simple control method of using a hysteretic comparator to monitor the output voltage versus a reference and decide whether to use the oscillator to turn
247. ill determine the peak to peak output voltage ripple but its impedance at high frequencies is not as critical due to the continuous nature of the output current waveform The situation is reversed in the case of the boost converter shown in Figure 3 59 Here the input waveform is continuous while the output waveform is pulsating The output capacitor must have good low and high frequency characteristics in order to minimize the output voltage ripple Boost converters are often followed by a post filter to remove the high frequency switching noise 3 59 SWITCHING REGULATORS BUCK CONVERTER INPUT AND OUTPUT CURRENT WAVEFORMS Vin BES gt tour Vout Figure 3 58 BOOST CONVERTER INPUT AND OUTPUT CURRENT WAVEFORMS i Vin IN lout Vout ER LOAD Cin INPUT CURRENT iiy OUTPUT CURRENT Figure 3 59 3 60 SWITCHING REGULATORS Switching regulator capacitors are generally of the electrolytic type because of the relatively large values required An equivalent circuit for an electrolytic capacitor is shown in Figure 3 60 In addition to the capacitance value itself the capacitor has some equivalent series resistance ESR and equivalent series inductance ESL It is useful to make a few assumptions and examine the approximate response of the capacitor to a fast current step input For the sake of the discussion assume the input current switches from 0 to 1A in 100ns Also assume that the ESR is 0 2Q and
248. in a low impedance over a wide range of frequencies With a good understanding of the behavior of real components a strategy can now be developed to find solutions to most EMI problems RADIO FREQUENCY INTERFERENCE The world is rich in radio transmitters radio and TV stations mobile radios computers electric motors garage door openers electric jackhammers and countless others All this electrical activity can affect circuit system performance and in extreme cases may render it inoperable Regardless of the location and magnitude of the interference circuits systems must have a minimum level of immunity to radio frequency interference RFI The next section will cover two general means by which RFI can disrupt normal instrument operation the direct effects of RFI sensitive analog circuits and the effects of RFI on shielded cables 8 65 HARDWARE DESIGN TECHNIQUES Two terms are typically used in describing the sensitivity of an electronic system to RF fields In communications radio engineers define immunity to be an instrument s susceptibility to the applied RFI power density at the unit In more general EMI analysis the electric field intensity is used to describe RFI stimulus For comparative purposes Equation 8 2 can be used to convert electric field intensity to power density and vice versa E 14 Eq 8 2 where E Electric Field Strength in volts per meter and Transmitted power in milliwatts per
249. inductor ripple current and current mode operation together with an optimal compensation design provide excellent line and load transient response The current limit level is user programmable with an external current sense resistor 3 46 SWITCHING REGULATORS ADP3153 POWER SUPPLY CONTROLLER FOR PENTIUM Il SIMPLIFIED SCHEMATIC Vec Vin 12V 5V 22uF 1uF 1500pF x 3 Vin PWRGD O L 5V DRIVE1 3 3uH VLDO IRL3103 a on Vo SUR IRL3103 ADP3153 o FB SENSE cd 14A 35kQ 1nF CMP SENSE 20kQ CT IRL3103 1500pF X Fi DRIVE2 TA 1N5818 1000pF 180pF nF AGND PGND VIDO VID4 NO 5 BIT VID CODE Figure 3 47 ADP3153 VID PROGRAMMABLE MICROPROCESSOR CONTROLLER KEY SPECIFICATIONS 5 Bit Digitally Programmable 1 8V to 3 5V Output Voltage Dual N Channel Driver Outputs Output Accuracy 1 0 C to 70 C Constant Off Time Variable Frequency Current Mode Control On Chip Adjustable Linear Regulator Controller 20 Lead TSSOP Package Suitable for Pentium Il Pentium Pro AMD K6 Processors Figure 3 48 3 47 SWITCHING REGULATORS INDUCTOR CONSIDERATIONS The selection of the inductor used in a switching regulator is probably the most difficult part of the design Fortunately manufacturers of switching regulators supply a wealth of applications information and standard off the shelf inductors from well known and reliable manufacturers are quite often recommended on the switching regulator data sheet
250. ing an output voltage which is either greater than or less than the absolute value of the input voltage A simple buck boost converter topology is shown in Figure 3 20 The input voltage is positive and the output voltage is negative When the switch is on the inductor current builds up When the switch is opened the inductor supplies current to the load through the diode Obviously this circuit can be modified for a negative input and a positive output by reversing the polarity of the diode BUCK BOOST CONVERTER 1 ViN VouT Vy SW Vour NEGATIVE The Absolute Value of the Output Can Be Less Than Or Greater Than the Absolute Value of the Input Figure 3 20 3 21 SWITCHING REGULATORS A second buck boost converter topology is shown in Figure 3 21 This circuit allows both the input and output voltage to be positive When the switches are closed the inductor current builds up When the switches open the inductor current is supplied to the load through the current path provided by D1 and D2 A fundamental disadvantage to this circuit is that it requires two switches and two diodes As in the previous circuits the polarities of the diodes may be reversed to handle negative input and output voltages BUCK BOOST CONVERTER 2 VouT Vn 5 1 Vout POSITIVE The Absolute Value of the Output Can Be Less Than Or Greater Than the Absolute Value of the Input Figure 3 21 Another way to accomplish the buck boost f
251. inimal radiated EMI Application circuits are simple and usually only two or three external capacitors are required Because there is no need for an inductor the final PCB component height can generally be made smaller than a comparable switching regulator This is important in many applications such as display panels Switched capacitor inverters are low cost and compact and are capable of achieving efficiencies greater than 90 Obviously the current output is limited by the size of the capacitors and the current carrying capacity of the switches Typical IC switched capacitor inverters have maximum output currents of about 150mA maximum Switched capacitor voltage converters do not maintain high efficiency for a wide range of ratios of input to output voltages unlike their switching regulator counterparts Because the input to output current ratio is scaled according to the basic voltage conversion i e doubled for a doubler inverted for an inverter regardless of whether or not regulation is used to reduce the doubled or inverted voltage any output voltage magnitude less than 2V y for a doubler or less than I VIN for an inverter will result in additional power dissipation within the converter and efficiency will be degraded proportionally 4 2 SWITCHED CAPACITOR VOLTAGE CONVERTERS SWITCHED CAPACITOR VOLTAGE CONVERTERS E No Inductors B Minimal Radiated EMI B Simple Implementation Only 2 External Capacitors Plus an Input
252. interference often occurs between digital and analog circuits or between motors or relays and digital circuits In mixed signal environments the digital portion of the system often interferes with analog circuitry In some systems the internal interference reaches such high levels that even very high speed digital circuitry can affect other low speed digital circuitry as well as analog circuits In addition to the source path receptor model for analyzing EMI related problems Kimmel Gerke Associates have also introduced the FAT ID concept Reference 1 FAT ID is an acronym that describes the five key elements inherent in any EMI problem These five key parameters are frequency amplitude time impedance and distance 8 63 HARDWARE DESIGN TECHNIQUES The frequency of the offending signal suggests its path For example the path of low frequency interference is often the circuit conductors As the interference frequency increases it will take the path of least impedance usually stray capacitance In this case the coupling mechanism is radiation Time and frequency in EMI problems are interchangeable In fact the physics of EMI shows that the time response of signals contains all the necessary information to construct the spectral response of the interference In digital systems both the signal rise time and pulse repetition rate produce spectral components according to the following relationship fEMI c Eq 8 1 T trise For
253. io per the equations Temperature 235 T2 Temperature 455 E Popular microcontrollers such as the 80C51 and 68HC11 have on chip timers which can easily decode the mark space ratio of the TMP03 TMP04 A typical interface to the 80C51 is shown in Figure 6 31 Two timers labeled Timer 0 and Timer 1 are 16 bits in length The 80C51 s system clock divided by twelve provides the source for the timers The system clock is normally derived from a crystal oscillator so timing measurements are quite accurate Since the sensor s output is ratiometric the actual clock frequency is not important This feature is important because the microcontroller s clock frequency is often defined by some external timing constraint such as the serial baud rate INTERFACING TMP04 TO A MICROCONTROLLER 45V 80C51 MICROCONTROLLER NOTE ADDITIONAL PINS OMITTED FOR CLARITY Figure 6 31 Software for the sensor interface is straightforward The microcontroller simply monitors I O port P1 0 and starts Timer 0 on the rising edge of the sensor output The microcontroller continues to monitor P1 0 stopping Timer 0 and starting Timer 1 when the sensor output goes low When the output returns high the sensor s T1 and T2 times are contained in registers Timer 0 and Timer 1 respectively Further software routines can then apply the conversion factor shown in the equations above and calculate the temperature 6 28
254. ioned and amplified before further processing can occur Except for IC sensors all temperature sensors have nonlinear transfer functions In the past complex analog conditioning circuits were designed to correct for the sensor nonlinearity These circuits often required manual calibration and precision resistors to achieve the desired accuracy Today however sensor outputs may be 6 1 TEMPERATURE SENSORS digitized directly by high resolution ADCs Linearization and calibration is then performed digitally thereby reducing cost and complexity Resistance Temperature Devices RTDs are accurate but require excitation current and are generally used in bridge circuits Thermistors have the most sensitivity but are the most non linear However they are popular in portable applications such as measurement of battery temperature and other critical temperatures in a system Modern semiconductor temperature sensors offer high accuracy and high linearity over an operating range of about 55 C to 150 C Internal amplifiers can scale the output to convenient values such as 10mV C They are also useful in cold junction compensation circuits for wide temperature range thermocouples Semiconductor temperature sensors can be integrated into multi function ICs which perform a number of other hardware monitoring functions Figure 6 2 lists the most popular types of temperature transducers and their characteristics TYPES OF TEMPERATURE SENSO
255. is the approximate average voltage drop across the switch When the switch is off the inductor current is discharged into a 3 35 SWITCHING REGULATORS voltage equal to VOUT Vp where Vp is the approximate average forward drop across the diode The basic inductor equation used to derive the relationship between the input and output voltage becomes VIN VOUT VSW VOUT VD X L on L In the actual regulator circuit negative feedback will force the duty cycle to maintain the correct output voltage but the duty cycle will also be affected by the switch and the diode drops to a lesser degree When the switch is on in a boost converter the voltage applied to the inductor is equal to VIN Vow When the switch is off the inductor current discharges into voltage equal to VOUT VIN Vp The basic inductor current equation becomes VIN NSW pc oa OUT SYN END h i L on L 0 From the above equations the basic relationships between input voltage output voltage duty cycle switch and diode drops can be derived for the buck and boost converters The ADP3000 is a switching regulator that uses the NPN type switch just discussed A block diagram is shown in Figure 3 33 and key specifications are given in Figure 3 34 ADP3000 SWITCHING REGULATOR BLOCK DIAGRAM GAIN BLOCK ERROR AMP 1 245V COMPARATOR REFERENCE DRIVER Figure 3 33 3 36 SWITCHING REGULATORS ADP3000 SWITCHING REGULATO
256. istance An actual PMOS pass device selected must satisfy all of these electrical requirements plus physical and thermal parameters There are a number of manufacturers offering suitable devices in packages ranging from SO 8 up through TO 220 in size To ensure that the maximum available drive from the controller will adequately drive the FET under worst case conditions of temperature range and manufacturing tolerances the maximum drive from the controller VGS DRIVE to the pass device must be determined This voltage is calculated as follows VGS DRIVE VIN VBE 2 50 REFERENCES AND LOW DROPOUT LINEAR REGULATORS where VIN is the minimum input voltage Ip MAX is the maximum load current Rg the sense resistor and 1 a voltage internal to the ADP3310 0 5 high temp 0 9 cold and 0 7V at room temp Note that since Ip MAX x Rg will be no more than 75mV and at cold temperature 0 9V this equation can be further simplified to VGS DRIVE VIN 1V In the Figure 2 43 example VIN 6V and Voy 5V so VGS DRIVE is6 1 BV It should be noted that the above two equations apply to FET drive voltages which are less than the typical gate to source clamp voltage of 8V built into the ADP3310 for the purposes of FET protection An overall goal of the design is to then select an FET which will have an RDS ON sufficiently low so that the resulting dropout voltage will be less than VIN
257. istics Cost and Complexity of Fast Charging Circuits Figure 5 1 There are an enormous number of tradeoffs to be made in selecting the battery and designing the appropriate charging circuits Weight capacity and cost are the primary considerations in most portable electronic equipment Unfortunately these considerations are not only interacting but often conflicting While slow charging charging time greater than 12 hours circuits are relatively simple fast charging circuits must be tailored to the battery chemistry and provide both reliable charging and charge termination Overcharging batteries can cause reduced battery life overheating the emission of dangerous corrosive gasses and sometimes total destruction For this reason fast charging circuits generally have built in backup means to terminate the charge should the primary termination method fail 5 1 BATTERY CHARGERS Understanding battery charger electronics requires a knowledge of the battery charge and discharge characteristics as well as charge termination techniques BATTERY FUNDAMENTALS Battery capacity C is expressed in Amp hours or mA hours and is a figure of merit of battery life between charges Battery current is described in units of C Rate For instance a 1000mA h battery has a C Rate of 1000mA The current corresponding to 1C is 1000mA and for 0 1C 100mA For a given cell type the behavior of cells with varying capacity is similar at the same C R
258. istors are available at the package pins so that the circuit can be recalibrated for other thermocouple types by the addition of resistors These terminals also allow more precise calibration for both thermocouple and thermometer applications The AD594 AD595 is available in two performance grades The C and the A versions have calibration accuracies of 1 C and 8 C respectively Both are designed to be used with cold junctions between 0 to 50 C The circuit shown in Figure 6 11 will provide a direct output from a type J thermocouple AD594 or a type K thermocouple AD595 capable of measuring 0 to 300 C AD594 AD595 MONOLITHIC THERMOCOUPLE AMPLIFIERS WITH COLD JUNCTION COMPENSATION 5V BROKEN THERMOCOUPLE ALARM Vout 10mV C OVERLOAD TYPE J AD594 DETECT TYPE K AD595 THERMOCOUPLE AD594 AD595 Figure 6 11 The AD596 AD597 are monolithic set point controllers which have been optimized for use at elevated temperatures as are found in oven control applications The device cold junction compensates and amplifies a type J K thermocouple to derive an internal signal proportional to temperature They can be configured to provide a voltage output 10mV C directly from type J K thermocouple signals The device is packaged in a 10 pin metal can and is trimmed to operate over an ambient range from 25 C to 100 C The AD596 will amplify thermocouple signals covering the entire 200 C to 760 temperature ra
259. itor 2 14 two terminal 2 2 XFET 2 10 14 characteristics 2 13 Voltage mode control 3 28 29 voltage feedforward 3 29 W Wainwright Instruments GmbH firm 8 87 Webster John G 6 38 Widlar Bob 2 24 2 57 Williams Jim 6 38 8 13 8 88 Winding resistance 3 58 Wong James 6 38 Wurcer Scott 8 88 Wynne J 8 77 Wynne John 8 87 X XFET voltage reference basic topology 2 10 11 characteristics 2 13 performance improvements 2 12 Z Zener buried 2 10 Zener diode breakdown 2 3 monolithic 2 4 temperature compensated 2 4 Index 13 INDEX Index 14 Analog Devices Parts Index A AD29X 2 16 AD524 8 47 AD574 2 23 AD580 2 5 6 AD582 8 47 AD584 2 6 AD585 8 47 AD586 2 10 2 14 2 16 2 18 AD587 2 18 AD588 2 10 2 14 2 16 AD589 2 5 2 7 2 19 AD592 6 21 22 AD594 595 6 9 10 AD596 597 6 10 AD680 2 5 2 16 AD688 2 16 AD712 8 47 AD7138 8 47 AD780 2 5 2 14 16 2 22 24 AD797 2 18 AD811 8 47 AD813 8 47 AD815 8 47 AD820 822 2 20 AD823 8 47 AD841 8 47 2 23 24 6 11 6 13 15 AD815 8 46 AD1580 2 5 2 7 2 19 AD1582 1585 series 2 5 2 8 9 2 14 16 2 22 AD7416 7417 7418 6 36 37 7547 8 48 7575 8 47 AD7705 7 11 AD7710 series 2 23 AD7817 7818 7819 6 36 37 AD8531 32 34 2 20 AD22108 6 22 23 ADM660 4 13 15 ADM707 7 4 ADMS800L 7 2 3 ADMS800M 7 2 3 ADM809 810 7 4 ADMS811 812 7 4 ADMS869X 7 4 ADM1232
260. ixed frequency 3 26 variable frequency constant off time 3 26 variable frequency constant on time 3 26 SEPIC topology 3 23 24 topology 3 1 valley control 3 31 versus controller 3 3 voltage mode control 3 28 29 Zeta converter 3 24 Synchronous rectifier 3 2 3 43 Synchronous switch 3 43 Systems Application Guide Analog Devices 8 77 8 86 T Tantalum electrolytic capacitor 3 63 8 20 22 Temperature 7 1 Temperature sensors 6 1 38 applications 6 1 resistance devices 6 2 6 11 15 semiconductor 6 2 6 19 38 bandgap 6 21 thermistor 6 2 6 16 19 thermocouple 6 2 11 types 6 2 Thandi Gurjit 4 1 Thermal Coastline 2 46 47 anyCAP applications 2 47 leadframe device 8 49 package details 2 47 8 49 Thermal design basics 8 46 Thermal management 8 45 58 airflow control 1 1 basics 8 45 50 design basics 8 46 heat sinks and airflow 8 51 58 overview 1 2 scope 1 1 temperature control 1 1 temperature sensors and control circuits 1 7 Index 12 Thermal mass 6 25 Thermal resistance 8 45 derating curves 8 48 49 heat sink 8 48 IC packages summary 8 46 48 junction ambient air measurement 8 46 junction device case measurement 8 48 Thermistor 6 16 19 advantages 6 16 17 high sensitivity 6 16 linearization 6 17 18 using shunt resistor 6 18 negative temperature coefficient 6 16 temperature coefficient 6 17 temperature sensors 6 2 Thermocouple basic principles 6 5 cold junction compensation
261. ize switching regulator output voltage ripple it is often necessary to add additional filtering In many cases this is more efficient than simply adding parallel capacitors to the main output capacitor to reduce ESR Output ripple current in a boost converter is pulsating while that of a buck converter is a sawtooth In any event the high frequency components in the output ripple current can be removed with a small inductor 2 to 10nH or so followed by a low ESR capacitor Figure 3 65 shows a simple LC filter on the output of a switching regulator whose switching frequency is f Generally the actual value of the filter capacitor is not as important as its ESR when filtering the switching frequency ripple For instance the reactance of a 100uF capacitor at 100kHz is approximately 0 016Q which is much less than available ESRs 3 66 SWITCHING REGULATORS The capacitor ESR and the inductor reactance attenuate the ripple voltage by a factor of approximately 27fL ESR The example shown in Figure 3 65 uses a 100H inductor and a capacitor with an ESR of 0 2Q This combination attenuates the output ripple by a factor of about 32 The inductor core material is not critical but it should be rated to handle the load current Also its DC resistance should be low enough so that the load current does not cause a significant voltage drop across it SWITCHING REGULATOR OUTPUT FILTERING p p filtered SWITCHING REGULATOR SWITCHING FREQUE
262. jor influence on almost all major regulator performance issues Most notable among these is the dropout voltage VMIN Figure 2 25a through 2 25e illustrates a number of pass devices which are useful within voltage regulator circuits shown in simple schematic form On the figure is also listed the salient for the device as it would typically be used which directly indicates its utility for use in an LDO Not shown in these various mini figures are the remaining circuits of a regulator It is difficult to fully compare all of the devices from their schematic representations since they differ in so many ways beyond their applicable dropout voltages For this reason the chart of Figure 2 26 is useful This chart compares the various pass elements in greater detail allowing easy comparison between the device types dependent upon which criteria is most important Note that columns A E correspond to the schematics of Figure 2 25a 2 25e Note also that the pro con comparison items are in relative terms as opposed to a hard specification limit for any particular pass device type For example it can be seen that the all NPN pass devices of columns A and B have the attributes of a follower circuit which allows high bandwidth and provides relative immunity to cap loading because of the characteristic low ZOUT However neither the single NPN nor the Darlington NPN can achieve low dropout for any load current This is because the Vpg s of the
263. l Voltage Environmental Concerns Operating Temperature Range Figure 5 3 Self Discharge Rate Measured in month or day Capacity C Measured in Amp hours A h or mA hours mA h Energy Density Volume Measured in Watt hours liter Wh l Energy Density Weight Measured in Watt hours kilogram Wh kg Cost Measured in Wh Memory Effect RECHARGEABLE BATTERY TECHNOLOGIES Sealed Nickel Nickel Lithium Lithium Lead Cadmium Metal lon Metal Acid Hydride Average Cell Voltage V 2 1 20 1 25 3 6 3 0 Energy Density Wh kg 35 45 55 100 140 Energy Density Wh l 85 150 180 225 300 Cost Wh 0 25 0 50 0 75 1 5 1 5 3 0 2 5 3 5 1 4 3 0 Memory Effect No Yes No No No Self Discharge month 5 10 25 20 25 8 1 2 Discharge Rate lt 5C gt 10C lt 3C lt 2C lt 2C Charge Discharge Cycles 500 1000 800 1000 1000 Temperature Range C Oto 50 10 to 50 10 to 50 10 to 50 30 to 55 Environmental Concerns Yes Yes No No No Based on AA Size Cell Figure 5 4 5 3 BATTERY CHARGERS Memory occurs only in NiCd batteries and is relatively rare It can occur during cyclic discharging to a definite fixed level and subsequent recharging Upon discharging the cell potential drops several tenths of a volt below normal and remains there for the rest of the discharge The total ampere hour capacity of the cell is not significantly affected Memory usu
264. l go to zero between cycles and the inductor current is said to be discontinuous It is necessary to understand this operating mode as well since many switchers must supply a wide dynamic range of output current where this phenomenon is unavoidable Waveforms for discontinuous operation are shown in Figure 3 12 BUCK CONVERTER WAVEFORMS DISCONTINUOUS MODE iL igut Lower Case Instantaneous Value sw Upper Case Average Value 0 Figure 3 12 Behavior during the switch on time is identical to that of the continuous mode of operation However during the switch off time there are two regions of unique behavior First the inductor current ramps down at the same rate as it does during continuous mode but then the inductor current goes to zero When it reaches zero the current tries to reverse but cannot find a path through the diode any longer So the voltage on the input side of the inductor same as the diode and switch junction 3 13 SWITCHING REGULATORS jumps up to VoyT such that the inductor has no voltage across it and the current can remain at zero Because the impedance at diode node vp is high ringing occurs due to the inductor L resonating with the stray capacitance which is the sum of the diode capacitance Cp and the switch capacitance Cow The oscillation is damped by stray resistances in the circuit and occurs at a frequency given by 1 JL CD fo A
265. lator Topologies Switch Modulation Techniques Control Techniques Diode and Switch Considerations a Inductor Considerations Capacitor Considerations E Input and Output Filtering SECTION 4 SWITCHED CAPACITOR VOLTAGE CONVERTERS B Charge Transfer Using Capacitors B Unregulated Switched Capacitor Inverter and Doubler Implementations Voltage Inverter and Doubler Dynamic Operation B Switched Capacitor Voltage Converter Power Losses BH Unregulated Inverter Doubler Design Example B Regulated Output Switched Capacitor Voltage Converters SECTION 5 BATTERY CHARGERS Battery Fundamentals E Battery Charging B X linear Battery Charger Switch Mode Dual Charger for Li lon NiCd and NiMH Batteries Hi Universal Charger for Li lon NiCd and NiMH SECTION 6 TEMPERATURE SENSORS B X Thermocouple Principles and Cold Junction Compensation E Resistance Temperature Detectors RTDs m Thermistors B Semiconductor Temperature Sensors SECTION 7 HARDWARE MONITORING SECTION 8 HARDWARE DESIGN TECHNIQUES Analog Circuit Simulation Prototyping Techniques Evaluation Boards Grounding Techniques for Regulator Circuits Power Supply Noise Reduction and Filtering Thermal Management EMI RFI Considerations Shielding Concepts PRACTICAL DESIGN TECHNIQUES FOR POWER AND THERMAL MANAGEMENT INTRODUCTION REFERENCES AND LOW DROPOUT LINEAR REGULATORS SWITCHING REGULATORS SWITCHED CAPACITOR VOLTAGE CONVERTERS BATTERY CHARGERS TEM
266. lel as well as to the corresponding Rg VOUT large area PCB lands Using 2 oz copper PCB material and one square inch of copper PCB land area as a heatsink it is possible to achieve net thermal resistance 0JA for mounted SO 8 devices on the order of 60 C W or less Such data is available for SO 8 power FETs see Reference 11 There are also a variety of larger packages with lower thermal resistance than the SO 8 but still useful with surface mount techniques Examples are the DPAK and D PAK etc For higher power dissipation applications corresponding to thermal resistance of 50 C W or less a bolt on external heat sink is required to satisfy the requirement Compatible package examples would be the TO 220 family which is used with the NDP6020P example of Fig 2 43 Calculating thermal resistance for VIN 6 7V VoyT 5V and Ij 1A _ Ty TA MAX VDS MAX IL where T is the pass device junction temperature limit is the maximum ambient temperature VDS MAX is the maximum pass device drain source voltage and Ip MAX is the maximum load current 9JA Inserting some example numbers of 125 C as a max junction temp for the NDP6020P a 75 C expected ambient and the VDS MAX and Ip MAX figures of 1 7V and 1 the required 0j A works out to be 125 75 1 7 29 4 C W This be met with a very simple heat sink which is derived as follows The NDP6020P in the TO 220 package has
267. less than 10mV peak to peak a value suitable for driving most analog circuits Measurements were made using a Tektronix wideband digitizing oscilloscope with the input bandwidth limited to 20MHz so that the ripple generated by the switching 8 26 HARDWARE DESIGN TECHNIQUES regulators could be more readily observed In a system power supply ripple frequencies above 20MHz are best filtered locally at each IC power pin with a low inductance ceramic capacitor and perhaps a series connected ferrite bead Probing techniques are critical for accurate ripple measurements A standard passive 10X probe was used with a bayonet probe tip adapter for making the ground connection as short as possible see Figure 8 20 Use of the ground clip lead is not recommended in making this type of measurement because the lead length in the ground connection forms an unwanted inductive loop which picks up high frequency switching noise thereby corrupting the signal being measured PROPER PROBING TECHNIQUES PROBE GROUND CLIP CONNECTOR SLIP ON BAYONET GROUND ADAPTER IN SIGNAL CONTACT Eom GROUND CLIP LEAD DO NOT USE GROUND PLANE CONTACT Figure 8 20 Note Schematic representation of proper physical grounding is almost impossible In all the following circuit schematics the connections to ground are made to the ground plane using the shortest possible connecting path regardless of how they are indicated i
268. lications 3 5 7 AC DC conversion 3 5 power distribution 3 6 advantages 3 6 buck boost cascaded 3 23 capacitor electrolytic equivalent circuit 3 61 impedance vs frequency 3 61 62 function 3 59 capacitors ripple currents 3 2 charge control 3 31 components 3 1 2 control techniques 3 28 31 controller 3 1 coupled inductor single ended primary inductance topology 3 23 24 Cuk converter 3 24 current mode control 3 29 30 diodes and switches 3 34 47 disadvantages 3 3 filtering output experiments 8 25 41 results summary 8 40 41 probing techniques 8 27 flyback buck boost circuit 3 25 forward converter circuit 3 26 gated oscillator control 3 31 34 scheme 3 32 grounding techniques 8 15 17 high frequency noise reduction 8 19 20 hysteretic current control 3 31 ideal efficiency 3 5 input voltage range 3 4 5 output current 3 5 output line load regulation 3 5 output ripple voltage 3 4 5 scheme 3 3 4 switching frequency 3 5 ideal lossless 3 34 inductor considerations 3 48 59 input filtering 3 67 68 circuit 3 68 isolated topologies 3 24 26 limitations 3 2 modulation 3 26 28 Index 11 INDEX noise 3 2 non isolated topologies 3 23 24 NPN switches 3 34 output analog ready 8 19 output filtering 3 66 67 circuit 3 67 pulse burst modulation 3 27 3 27 28 3 31 34 disadvantages 3 28 voltage mode control 3 28 29 pulse skipping normal duty cycles 3 27 pulse width modulation f
269. linear regulator 2 25 57 in cell phones 1 5 6 Low dropout regulator see LDO M McDaniel Wharton 2 57 8 58 Magnetic flux density versus inductor current 3 56 Index 8 Magnetic hysteresis 3 58 Magnetizing current 3 25 Marsh Dick 3 69 8 44 Marsh Richard 8 88 Massobrio Guiseppi 8 13 Memory battery 5 4 Metals dissimilar thermoelectric e m f 6 6 Microconverters 7 13 Micromodel 8 1 Microprocessor Chip Enable inhibiting 7 1 supervisory products 7 1 2 supply voltage 7 1 Mixed signal circuit definition 8 1 Morrison Ralph 8 77 8 86 8 87 MOSFET manufacturer listing 3 70 Muncy Neil 8 86 N NDP6020P NDB6020P P Channel Logic Level Enhancement Mode Field Effect Transistor 2 57 Nickel cadmium battery 5 3 4 5 6 fast charge termination methods 5 7 8 fast charging characteristics 5 6 slow charging characteristics 5 6 temperature and voltage charging characteristics 5 7 Nickel metal hydride battery 5 3 4 5 6 fast charge termination methods 5 7 8 fast charging characteristics 5 6 slow charging characteristics 5 6 temperature and voltage charging characteristics 5 7 Noise conducted 3 2 radiated 3 2 voltage references 2 16 19 Noise reduction and filtering manufacturer listing 8 44 NTC see negative temperature coefficient O Oersted 3 55 Off line charger laptop computers 5 16 17 OMEGA Temperature Measurement Handbook 6 38 OP113EP precision low noise unity gain follower 2 1
270. lk is significant above 50MHz Traces are protected B Disadvantages of Embedding Lower interboard capacitance harder to decouple Impedances may be too low for matching Hard to prototype and troubleshoot buried traces Figure 8 72 Much has been written about terminating printed circuit board traces in their characteristic impedance to avoid reflections A good rule of thumb to determine when this is necessary is as follows Terminate the line in its characteristic impedance when the one way propagation delay of the PCB track is equal to or greater than one half the applied signal rise fall time whichever edge is faster A conservative approach is to use a 2 inch PCB track length nanosecond rise fall time criterion For example PCB tracks for high speed logic with rise fall time of 5ns should be terminated in their characteristic impedance if the track length is equal to or greater than 10 inches including any meanders The 2 inch nanosecond track length criterion is summarized in Figure 8 73 for a number of logic families This same 2 inch nanosecond rule of thumb should be used with analog circuits in determining the need for transmission line techniques For instance if an amplifier must output a maximum frequency of fmax then the equivalent risetime ty can be calculated using the equation t 0 35 fj 45 The maximum PCB track length is then calculated by multiplying the risetime by 2 inch nanosecond For example
271. lly in the diagram for clarity The output voltage is set by the resistor divider ratio and the reference voltage R2 VOUT vner 1 3 28 SWITCHING REGULATORS The internal resistor ratios and the reference voltage are set to produce standard output voltage options such as 12V 5V 3 3V or 3V In some regulators the resistor divider can be external allowing the output voltage to be adjusted VOLTAGE FEEDBACK FOR PWM CONTROL SWITCHING REG IC INDUCTOR DIODE NOTE RESISTORS AMPLIFIER AND VREF INCLUDED IN SWITCHING REGULATOR IC Figure 3 27 A simple modification of VM control is voltage feedforward This technique adjusts the duty cycle automatically as the input voltage changes so that the feedback loop does not have to make an adjustment or as much of an adjustment Voltage feedforward can even be used in the simple PBM regulators Feedforward is especially useful in applications where the input voltage can change suddenly or perhaps due to current limit protection limitations it is desirable to limit the maximum duty cycle to lower levels when the input voltage is higher In switchers the VM control loop needs to be compensated to provide stability considering that the voltage being controlled by the modulator is the average voltage produced at the switched node whereas the actual output voltage is filtered through the switcher s LC filter The phase shift produced by the filter can make it difficult to
272. lly any good quality output capacitor used for Cy as is true with the other anyCAP devices The actual CT value required and its associated ESR depends on the gm and capacitance of the external PMOS device In 2 49 REFERENCES AND LOW DROPOUT LINEAR REGULATORS general a 10uF capacitor at the output is sufficient to ensure stability for load currents up to 10A Larger capacitors can also be used if high output surge currents are present In such cases low ESR capacitors such as OS CON electrolytics are preferred because they offer lowest ripple on the output For less demanding requirements a standard tantalum or aluminum electrolytic can be adequate When an aluminum electrolytic is used it should be qualified for adequate performance over temperature The input capacitor is only necessary when the regulator is several inches or more distant from the raw DC filter capacitor However since it is a small type it is usually prudent to use it in most instances located close to the VIN Pin of the regulator A BASIC ADP3310 PMOS FET 1A LDO REGULATOR CONTROLLER CIRCUIT Rs 6V MIN sog NDP6020P OR NDB6020P Vour 25V FAIRCHILD C GATE Vout Cin ADP3310 5 Figure 2 43 Selecting the Pass Device The type and size of the pass transistor are determined by a set of requirements for threshold voltage input output voltage differential load current power dissipation and thermal res
273. luding telephone listings is on the outside back cover of the 1997 Short Form Designers Guide DSP Support Center Fax requests to 49 89 57005 200 or e mail dsp europe analog com The Bulletin Board Service is at 43 1 8887656 Australia and New Zealand B Fax Retrieval Telephone number 61 59 864377 Follow the voice prompts India bz Call 91 80 526 3606 or fax 91 80 526 3713 and request the data sheet of interest Other Locations World Wide Web Our address is http www analog com Use the browser of your choice and follow the prompts Analog Devices Sales Offices Call your local sales office and request a data sheet A Worldwide Sales Directory including telephone numbers is listed on the back cover of the 1997 Short Form Designers Guide TECHNICAL SUPPORT AND CUSTOMER SERVICE In the U S A and Canada call 800 ANALOGD 800 262 5643 For technical support on all products select option one then select the product area of interest For price and delivery select option three For literature and samples select option two Non 800 Number 781 937 1428 PRACTICAL DESIGN TECHNIQUES FOR POWER AND THERMAL MANAGEMENT ANALOG DEVICES ACKNOWLEDGMENTS Thanks are due the many technical staff members of Analog Devices in Engineering and Marketing who provided invaluable inputs during this project Particular credit is due the individual authors whose names appear at the beginning of their material S
274. ly voltage can range between 2V and 12V in the boost mode and up to 30V in the buck mode It should be noted that when the oscillator is turned off the internal switch is opened so that the inductor current does not continue to increase 3 32 SWITCHING REGULATORS REPRESENTATIVE OUTPUT VOLTAGE WAVEFORM FOR GATED OSCILLATOR CONTROLLED PBM BUCK REGULATORS R2 VOUT vner ij Figure 3 30 In the gated oscillator method the comparator hysteresis serves to stabilize the feedback loop making the designs relatively simple The disadvantage of course is that the peak to peak output voltage ripple can never be less than the comparator hysteresis multiplied by the reciprocal of the attenuation factor Output Ripple 2 Vhysteresis 22 Because the gated oscillator controlled switching regulator operates with a fixed duty cycle output regulation is achieved by changing the number of skipped pulses as a function of load current and voltage From this perspective PBM controlled switchers tend to operate in the discontinuous mode under light load conditions Also the maximum average duty cycle is limited by the built in duty cycle of the oscillator Once the required duty cycle exceeds that limit no pulse skipping occurs and the device will lose regulation One disadvantage of the PBM switching regulator is that the frequency spectrum of the output ripple is fuzzy because of the burst mode of
275. m peak and continuous current levels by a factor of 20 or so the inductor should be satisfactory for the application If these simple guidelines are observed then the designer can be reasonable confident that the major sources of efficiency losses will be due to other parts of the regulator i e the switch I2R gate charge on voltage the diode on voltage and the quiescent power dissipation of the regulator itself One method to ensure that the inductor losses do not significantly degrade the regulator performance is to measure the Q of the inductor at the switching frequency If the Q is greater than about 25 then the losses should be insignificant 3 58 SWITCHING REGULATORS There are many possible choices in inductor core materials ferrite molypermalloy MPP ferrite powdered iron etc High efficiency converters generally cannot accommodate the core loss found in the low cost powdered iron cores forcing the use of more expensive ferrite molypermalloy MPP or Kool My cores Ferrite core material saturates hard which causes the inductance to collapse abruptly when the peak current is exceeded This results in a sharp increase in inductor ripple current Molypermalloy from Magnetics Inc is a very good low loss core material for toroids but is more expensive than ferrite A reasonable compromise from the same manufacturer is Kool The final consideration is the inductor self resonant freque
276. minate the switch on time Since it is inductor current that turns off the switch and thereby sets the duty cycle this method is commonly referred to as current mode control even though there are actually two feedback control loops the fast responding current loop and the slower responding output voltage loop Note that inductor current is being controlled on a pulse by pulse basis which simplifies protection against switch over current and inductor saturation conditions In essence then in CM control rather than controlling the average voltage which is applied to the LC filter as in VM control the inductor current is controlled directly on a cycle by cycle basis The only phase shift remaining between the inductor current and the output voltage is that produced by the impedance of the output capacitor s The correspondingly lower phase shift in the output filter allows the loop response to be faster while still remaining stable Also instantaneous changes in input voltage are immediately reflected in the inductor current which provides excellent line transient response The obvious disadvantage of CM control is the requirement of sensing current and if needed an additional amplifier With increasingly higher performance requirements in modern electronic equipment the performance advantage of CM control typically outweighs the cost of 3 30 SWITCHING REGULATORS implementation Also some sort of current limit protection is often re
277. minated entirely However bypassing the prototype phase in high speed high performance analog or mixed signal circuit designs can be risky for a number of reasons For the purposes of this discussion an analog circuit is any circuit which uses ICs such as op amps instrumentation amps programmable gain amps PGAs voltage controlled amps VCAs log amps mixers analog multipliers voltage references etc A mixed signal circuit is an A D converter ADC D A converter DAC or combinations of these in conjunction with some amount of digital signal processing which may or may not be on the same IC as the converters Switching regulators must be classified as high speed analog circuits because of the frequencies generated by the internal or external switching action Consider a typical IC operational amplifier It may contain some 20 40 transistors almost as many resistors and a few capacitors A complete SPICE Simulation Program with Integrated Circuit Emphasis see Reference 1 model will contain all these components and probably a few of the more important parasitic capacitances and spurious diodes formed by the various junctions in the op amp chip For high speed ICs the package and wirebond parasitics may also be included This is the type of model that the IC designer uses to optimize the device during the design phase and is typically run on a CAD workstation Because it is a detailed model it will be referred to as a micromodel In
278. mmunity minimizes the possibility of spurious triggering by noise spikes on the supplies being monitored A block diagram of the device is shown in Figure 7 5 key features in Figure 7 6 and a pager application circuit in Figure 7 7 7 4 HARDWARE MONITORING ADM9261 TRIPLE COMPARATOR AND REFERENCE bo 2 5 TO 43 6V 501 9V BAT 4V THRESHOLD ERR1 SU2 3 3V 1 3 0V THRESHOLD ERR2 SU3 3 3V 2 2 8V THRESHOLD Ll ERR3 GND eal REFERENCE Figure 7 5 ADM9261 KEY SPECIFICATIONS Simultaneous Monitoring of 9V and two 3 3V Supplies Limits set at 4V for SU1 9V Input 3 0V for SU2 3 3V Input and 2 8V for SU3 3 3V Input Vec 2 5V to 3 6V Low Power 10pA Typical Internal Comparator Hysteresis 3 Power Supply Glitch Immunity 20 5 100mV 501 503 Guaranteed from 10 to 60 C No External Components Required 8 pin Micro SOIC Package Figure 7 6 7 5 HARDWARE MONITORING ADM9261 PAGER POWER SYSTEM APPLICATION CIRCUIT ADM9261 ERR2 503 GND ERR3 Figure 7 7 The ADM9264 is a quad power supply monitor IC which simultaneously monitors four separate supply voltage and outputs error signals if any of the supply voltages go above or below preset limits It is designed for desktop PC supply monitoring but can be used in any system where multiple power supplies require monitoring Each power supply monitor circuit uses a proprietary window comparator design whe
279. most IC switching regulators the impedance is limited by the ESR to 0 2Q Above about 1MHz the capacitor behaves like an inductor due to the ESL of 20nH These values although they may vary somewhat depending upon the actual type of electrolytic capacitor aluminum general purpose aluminum 3 61 SWITCHING REGULATORS switching type tantalum or organic semiconductor are representative and illustrate the importance of understanding the limitations of capacitors in switching regulators TYPICAL ELECTROLYTIC CAPACITOR IMPEDANCE VERSUS FREQUENCY C 100pF ESL 20nH x REGION REGION ET LOG 12 ESR 0 20 REGION ESR 020 529 10kHz 1MHz LOG FREQUENCY Figure 3 61 From the electrolytic capacitor impedance characteristic it is clear that the ESR and ESL of the output capacitor will determine the peak to peak output voltage ripple caused by the switching regulator output ripple current In most electrolytic capacitors ESR degrades noticeably at low temperature by as much as a factor of 4 6 times at 55 C vs the room temperature value For circuits where ESR is critical to performance this can lead to problems Some specific electrolytic types do address this problem for example within the HFQ switching types the 10 C ESR at 100kHz is no more than 2x that at room temperature The OS CON electrolytics have a ESR vs temperature characteristic which is relatively flat There are generally th
280. mple Figure 8 61 illustrates the real behavior of the passive components used in circuit design At very high frequencies wires become transmission lines capacitors become inductors inductors become capacitors and resistors behave as resonant circuits 8 64 HARDWARE DESIGN TECHNIQUES ALL PASSIVE COMPONENTS EXHIBIT NON IDEAL BEHAVIOR Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA COMPONENT LF BEHAVIOR HF BEHAVIOR RESPONSE 2 0 AN o Mm ub WIRE 54 QJ om ype CAPACITOR os oM 0 INDUCTOR 2 rd NES ANN o0 v RESISTOR gt f Figure 8 61 A specific case in point is the frequency response of a simple wire compared to that of a ground plane In many circuits wires are used as either power or signal returns and there is no ground plane A wire will behave as a very low resistance less than 0 02Q ft for 22 gauge wire at low frequencies but because of its parasitic inductance of approximately 20nH inch it becomes inductive at frequencies above 13kHz Furthermore depending on size and routing of the wire and the frequencies involved it ultimately becomes a transmission line with an uncontrolled impedance From our knowledge of RF unterminated transmission lines become antennas with gain On the other hand large area ground planes are much more well behaved and mainta
281. n 2 57 8 58 Linden David 5 25 Line sensitivity voltage references 2 16 Linear Design Seminar Analog Devices 8 87 Linear regulator low dropout routing techniques 8 14 15 Linear voltage regulator 2 25 basic considerations 2 25 29 block diagram 2 28 Index 7 INDEX controller charging lithium ion battery 5 17 18 current limiting 2 28 dropout voltage 2 25 low noise 2 25 pass devices advantages disadvantages 2 30 Darlington NPN 2 29 30 dropout voltage 2 29 PMOS 2 30 PNP NPN 2 30 single NPN 2 29 30 single PNP 2 30 tradeoffs 2 29 33 temperature sensing 2 28 29 three terminal diagram 2 26 positive leg series style 2 25 power dissipation 2 26 27 types 2 25 26 Lithium ion battery 5 3 4 5 6 charge termination techniques 5 9 charger linear regulator controller 5 17 18 charging caveats 5 10 charging chemistry 5 8 end of charge detect 5 20 22 fast charging characteristics 5 6 5 9 multiple cell packs 5 9 overcharging sensitivity 5 8 slow charging characteristics 5 6 undercharging effects 5 10 LM109 309 bandgap voltage reference 2 31 LM309 fixed voltage regulator 2 32 33 dominant pole 2 36 emitter follower 2 36 LM317 high dropout voltage from Darlington pass transistors 2 33 pass device topology modifications 2 33 simplified schematic form 2 32 33 LO PADS 8 5 Load sensitivity voltage references 2 15 16 Long term drift in precision analog circuits 2 14 Low dropout
282. n a monolithic chip see Figure 6 11 It combines an ice point reference with a precalibrated amplifier to provide a high level 10mV C output directly from the thermocouple signal Pin strapping options allow it to be used as a linear amplifier compensator or as a switched output set point controller using either fixed or remote set point control It can be used to amplify its compensation voltage directly thereby becoming a stand alone Celsius transducer with 10mV C output In such applications it is very important that the IC chip is at the same temperature as the cold junction of the thermocouple which is usually achieved by keeping the two in close proximity and isolated from any heat sources The AD594 AD595 includes a thermocouple failure alarm that indicates if one or both thermocouple leads open The alarm output has a flexible format which includes TTL drive capability The device can be powered from a single ended supply which may be as low as 45V but by including a negative supply temperatures below 0 C can be measured To minimize self heating an unloaded AD594 AD595 will operate with a supply current of 160pA but is also capable of delivering 5mA to a load The AD594 is precalibrated by laser wafer trimming to match the characteristics of type J iron constantan thermocouples and the AD595 is laser trimmed for type K 6 9 TEMPERATURE SENSORS chromel alumel The temperature transducer voltages and gain control res
283. n output voltage of 5V its output voltage can be adjusted between 5V and 2V N with an external resistor using the equation VOUT DEG 5V for VOUT lt 2VIN 4 19 SWITCHED CAPACITOR VOLTAGE CONVERTERS ADP3607 ADP3607 5 APPLICATION CIRCUIT 1N5817 Vout 5V Vin 3V TO 5V for ADP3607 5 ADP3607 5 ADP3607 SEE TEXT Figure 4 22 When using either the ADP3607 or the ADP3607 5 in the adjustable mode the output current should be no greater than 30mA in order to maintain good regulation The circuit shown in Figure 4 23 generates a regulated 12V output from a 5V input using the ADP3607 5 in a voltage tripler application Operation is as follows First assume that the VgENSE pin of the ADP3607 5 is grounded and that the resistor R is not connected The output of the ADP3607 5 is an unregulated voltage equal to 2VIN The voltage at the Cp pin of the ADP3607 5 is a square wave with a minimum value of VIN and a maximum value of 2VjN When the voltage at Cp is VIN capacitor C2 is charged to VIN less the D1 diode drop from via diode D1 When the voltage at Cp is ZVIN the output capacitor C4 is charged to a voltage 3VrN less the diode drops of D1 and D2 The final unregulated output voltage of the circuit VOUT is therefore approximately 2Vp where Vp is the Schottky diode voltage drop The addition of the feedback resistor R ensures that the output is regulated for values of VOUT
284. n the actual circuit schematic diagram ADP3000 2V To 5V 100MA BOOST REGULATOR Figure 8 21 shows the connection diagram for the ADP3000 used as a 2V to 5V 100mA boost regulator The actual switch is internal to the device Multiple capacitors are used on both the input and output in order to lower the ESR and ESL 8 27 HARDWARE DESIGN TECHNIQUES ADP3000 2V TO 5V BOOST REGULATOR L1 12 5pH 1 5817 33uF 33uF 33 16V 16V 16V Vour C2 5V 100 400 16V ADP3000 ADJ 33yF x 3 100uF S3yF 16V x 3 GND SW2 33yF 33uF L1 COILTRONICS 25 4 C1 C2 SPRAGUE 293D SERIES SURFACE MOUNT TANTALUM Figure 8 21 The input waveform of the boost regulator is shown in Figure 8 22 and is typical of the gated oscillator type of regulation used the ADP3000 It consists of series of gradually decreasing ramp waveforms during the time the inductor is being switched at the 400kHz internal oscillator rate When the output voltage reaches the proper value the internal oscillator is turned off and the input capacitors recharge as indicated by the positive going ramp voltage During this interval the output voltage gradually decays until the point at which the internal oscillator is gated on again and the cycle repeats itself The output waveform for the circuit is shown in Figure 8 23 This waveform is also characteristic of gated oscillator boost regulators as i
285. nalog Multiplexer On Chip Temperature Sensor and Bandgap Voltage Reference 2 DACs 8 bits with Voltage Output Buffers 3V or 5V Single Supply Operation 64 Digital I O for Address Data Interrupts LEDs LCDs UART and I C Compatible SPI Serial Interfaces 3 x 16 bit Timers Counters 2 Muxed for 4 Channel Fans Independent Watchdog Clock and Supply Monitor Power Management of Peripherals and I O Figure 7 18 ADuC810PC PROCESSOR DC TO 16MHz Static Industry Standard 8051 MCU for up to 1MIP Operation 48k Bytes Flash Program Memory 2k Bytes Flash EEPROM Lockable User Data Memory 1k Bytes Data RAM 64k Bytes External Program and Data Memory Space Enhanced Hooks Emulation and Debugging Tools Resident Loader and Debugger Simplified I O through Special Function Registers Multi Level Maskable Interrupts 100 pin PQFP 14x14mm Package Figure 7 19 HARDWARE MONITORING REFERENCE Bill Schweber Supervisory ICs Establish System Boundaries EDN Sept 28 1995 p 71 7 15 HARDWARE DESIGN TECHNIQUES SECTION 8 HARDWARE DESIGN TECHNIQUES Walt Kester Walt Jung James Bryant Bill Chestnut ANALOG CIRCUIT SIMULATION In recent years there has been much pressure placed on system designers to verify their designs with computer simulations before committing to actual printed circuit board layouts and hardware Simulating complex digital designs is extremely beneficial and very often the prototype phase can be eli
286. ncy A practical example would be an inductor of 10nH which has an equivalent distributed capacitance of 5pF self resonant frequency can be calculated as follows 1 1 fresonance VLC 22MHz The switching frequency of the regulator should be at least ten times less than the resonant frequency In most practical designs with switching frequencies less than 1MHz this will always be the case but a quick calculation is a good idea CAPACITOR CONSIDERATIONS Capacitors play a critical role in switching regulators by acting as storage elements for the pulsating currents produced by the switching action Although not shown on the diagrams previously all switching regulators need capacitors on their inputs as well as their outputs for proper operation The capacitors must have very low impedance at the switching frequency as well as the high frequencies produced by the pulsating current waveforms Recall the input and output current waveforms for the simple buck converter shown in Figure 3 58 Note that the input current to the buck converter is pulsating while the output is continuous Obviously the input capacitor is critical for proper operation of the regulator It must maintain the input at a constant voltage during the switching spikes This says that the impedance of the capacitor must be very low at high frequencies much above the regulator switching frequency The load capacitor is also critical in that its impedance w
287. ncy slightly even if not required 3 43 SWITCHING REGULATORS BUCK CONVERTER WITH SYNCHRONOUS SWITCH USING P AND N CHANNEL MOSFETS P DRIVE N DRIVE SCHOTTKY DIODE PREVENTS BODY DIODE OF N CHANNEL MOSFET FROM CONDUCTING DURING DEADTIME Figure 3 43 The ADP1148 is a high efficiency synchronous step down switching regulator controller with an input voltage range of 3 5V to 18V It utilizes a constant off time variable frequency current mode control topology and is available in three versions the ADP1148 3 3 3 3V output the ADP1148 5 5V output and the ADP1148 adjustable output At low output currents the device switches into a power saving mode to maintain high efficiency An application circuit for the ADP1148 synchronous step down regulator controller is shown in Figure 3 44 Operation of the ADP1148 is similar to the ADP1147 with the addition of the drive circuitry for the synchronous N channel MOSFET The input voltage can range from 5 2V to 18V and the output is 5V at 2A A breakdown of the ADP1148 efficiency losses is shown in Figure 3 45 where the lower curve represents the total efficiency Key specifications for the device are given in Figure 3 46 3 44 SWITCHING REGULATORS HIGH EFFICIENCY STEP DOWN REGULATOR USING THE ADP1148 CONTROLLER Vin 5 2V TO 18V QO 1yF O 10nF Vin IRF7204 100yF P DRIVE9 gt P CH R 0V NORMAL ADP1148 5 I L 62uH your 1 5V SHUTDOWN
288. nd ripple currents A subset of aluminum electrolytic capacitors is the switching type designed for handling high pulse currents at frequencies up to several hundred kHz with low losses Reference 4 This capacitor type competes directly with tantalums in high frequency filtering applications with the advantage of a broader range of values A more specialized high performance aluminum electrolytic capacitor type uses an organic semiconductor electrolyte Reference 5 The OS CON capacitors feature appreciably lower ESR and higher frequency range than do other electrolytic types with an additional feature of low low temperature ESR degradation Film capacitors are available in very broad value ranges and an array of dielectrics including polyester polycarbonate polypropylene and polystyrene Because of the low dielectric constant of these films their volumetric efficiency is quite low and a 10uF 50V polyester capacitor for example is actually a handful Metalized as opposed to foil electrodes does help to reduce size but even the highest dielectric constant units among film types polyester polycarbonate are still larger than any electrolytic even using the thinnest films with the lowest voltage ratings 50V Where film types excel is in their low dielectric losses a factor which may not necessarily be a practical advantage for filtering switchers For example ESR in film capacitors can be as low as 10m2 or less and the behavior of
289. ndicated by the pulsating waveforms followed by the decaying ramp voltage It should be noted that the ripple in this waveform is almost entirely determined by the equivalent ESR of the parallel combination of the output capacitors Adding more capacitors would reduce the ripple but a more effective method is to add an LC filter on the output as shown in Figure 8 24 8 28 HARDWARE DESIGN TECHNIQUES ADP3000 BOOST INPUT WAVEFORM P od Vour S 5V 100mA 37mV ADP3000 BOOST REGULATOR CIRCUIT C1 100yF C2 100 S3pgF 16V x 3 S3pF 16V x 3 TUE 10 0mV 5 0045 Chi 7 62 0mV VERTICAL SCALE 10mV DIV HORIZ SCALE 5ps DIV C1 C2 33uF 16V x 3 SPRAGUE 293D SURFACE MOUNT TANTALUM Figure 8 22 ADP3000 BOOST OUTPUT WAVEFORM Vout 5V 100mA ADP3000 BOOST REG CIRCUIT C1 100yF C2 100 i S3pF 16V x 3 S3pF 16V x 3 TED 10 0mv 5 0045 Chi 7 62 0mV VERTICAL SCALE 10mV DIV HORIZ SCALE 5ps DIV C1 C2 33pgF 16V x 3 SPRAGUE 293D SURFACE MOUNT TANTALUM Figure 8 23 8 29 HARDWARE DESIGN TECHNIQUES ADP3000 BOOST FILTERED OUTPUT CONDITION 1 x 12 5HH 100 mA Lr Vour 14mV p p C1 100yF C2 100 16 x 3 S3pF 16V x 3 10 0mV M5 00us Chi 7 200gV VERTICAL SCALE 10mV DIV HORIZ SCALE 5ps DIV OUTPUT FILTER Lr 12 5pH COILTRONICS CTX25 4 Cp 479F 10V SURFACE MOU
290. nductance ESL determines the frequency where the net impedance characteristic switches from capacitive to inductive This varies from as low as 10kHz in some electrolytics to as high as 100MHz or more in chip ceramic types Both ESR and ESL are minimized when a leadless package is used All capacitor types mentioned are available in surface mount packages preferable for high speed uses The electrolytic family provides an excellent cost effective low frequency filter component because of the wide range of values a high capacitance to volume ratio and a broad range of working voltages It includes general purpose aluminum electrolytic types available in working voltages from below 10V up to about 500V and in size from 1 to several thousand uF with proportional case sizes All electrolytic capacitors are polarized and thus cannot withstand more than a volt or so of reverse bias without damage They also have relatively high leakage currents up to tens of uA and strongly dependent upon design specifics A subset of the general electrolytic family includes tantalum types generally limited to voltages of 100V or less with capacitance of 500uF or less Reference 3 In a given size tantalums exhibit a higher capacitance to volume ratios than do general purpose electrolytics and have both a higher frequency range and lower ESR They are generally more expensive than standard electrolytics and must be carefully applied with respect to surge a
291. neric synchronous switching regulator controller IC and the associated external MOSFET switching transistors The heavy bold lines indicate the paths where there are large switching currents and or high DC currents Notice that all these paths are connected together at a single point ground which in turn connects to a large area ground plane GROUNDING AND SIGNAL ROUTING TECHNIQUES FOR SWITCHING REGULATORS METHOD 1 SWITCHING REGULATOR CONTROLLER HIGH TRANSIENT CURRENTS USE SHORT HEAVY TRACES SGND PGND COMMON Figure 8 12 In order to minimize stray inductance and resistance each of the high current paths should be as short as possible Capacitors C1 and C2A must absorb the bulk or the input and output switching current and shunt it to the single point ground Any additional resistance or inductance in series with these capacitors will degrade their effectiveness Minimizing the area of all the loops containing the switching currents prevents them from significantly affecting other parts of the circuit In actual practice however the single point concept in Figure 8 12 is difficult to implement without adding additional lead length in series with the various components The added lead length required to implement the single point grounding scheme tends to degrade the effects of using the single point ground in the first place A more practical solution is to make multiple connections to the ground plane and
292. ng the inductor value however will result in a larger peak to peak output ripple current while increasing the value results in smaller ripple There are many other tradeoffs involved in selecting the inductor and these will be discussed in a later section 3 12 SWITCHING REGULATORS In this simple model line and load regulation of the output voltage is achieved by varying the duty cycle using a pulse width modulator PWM operating at a fixed frequency f The PWM is in turn controlled by an error amplifier an amplifier which amplifies the error between the measured output voltage and a reference voltage As the input voltage increases the duty cycle decreases and as the input voltage decreases the duty cycle increases Note that while the average inductor current changes proportionally to the output current the duty cycle does not change Only dynamic changes in the duty cycle are required to modulate the inductor current to the desired level then the duty cycle returns to its steady state value In a practical converter the duty cycle might increase slightly with load current to counter the increase in voltage drops in the circuit but would otherwise follow the ideal model This discussion so far has assumed the regulator is in the continuous mode of operation defined by the fact that the inductor current never goes to zero If however the output load current is decreased there comes a point where the inductor current wil
293. nge Supply Current 654A max Initial Accuracy 0 1 max Temperature Coefficient 50 ppm C max Noise 50pV rms 10Hz 10kHz Long Term Drift 100ppm 1khrs High Output Current 5mA min Temperature Range 40 C to 85 C Low Cost SOT 23 Package Figure 2 6 The circuit diagram for the series shown in Figure 2 7 may be recognized as a variant of the basic Brokaw bandgap cell as described under Figure 2 4 In this case Q1 Q2 form the core and the overall loop operates to produce the stable reference voltage VBG at the base of Q1 A notable difference here is that the op amp s output stage is designed with push pull common emitter stages This has the effect of requiring an output capacitor for stability but it also provides the IC with relatively low dropout operation The low dropout feature means essentially that VIN can be lowered to as close as several hundred mV above the VOUT level without disturbing operation The push pull operation also means that this device series can actually both sink and source currents at the output as opposed to the classic reference operation of sourcing current only For the various output voltage ratings the divider R5 R6 is adjusted for the respective levels The AD1582 series is designed to operate with quiescent currents of only 65 maximum which allows good power efficiency when used in low power systems with varying voltage inputs The rated output current for the series is 5 mA and
294. nge recommended for type J thermocouples while the AD597 can accommodate 200 C to 1250 C type K inputs They have a calibration accuracy of 4 C at an ambient temperature of 60 C and an ambient temperature stability specification of 0 05 C C from 25 C to 100 C None of the thermocouple amplifiers previously described compensate for thermocouple non linearity they only provide conditioning and voltage gain High resolution ADCs such as the AD77XX family can be used to digitize the 6 10 TEMPERATURE SENSORS thermocouple output directly allowing a microcontroller to perform the transfer function linearization as shown in Figure 6 12 The two multiplexed inputs to the ADC are used to digitize the thermocouple voltage and the cold junction temperature sensor outputs directly The input PGA gain is programmable from 1 to 128 and the ADC resolution is between 16 and 22 bits depending upon the particular ADC selected The microcontroller performs both the cold junction compensation and the linearization arithmetic AD77XX ADC USED WITH TMP35 TEMPERATURE SENSOR FOR CJC 5V DEPENDING ON ADC CONTROL REGISTER OUTPUT REGISTER SERIAL INTERFACE THERMO COUPLE G 1 TO 128 AD77XX SERIES 16 22 BITS TO MICROCONTROLLER Figure 6 12 RESISTANCE TEMPERATURE DETECTORS RTDs The Resistance Temperature Detector or the RTD is a sensor whose resistance changes with temperature Typ
295. ns They have a relatively high capacitance to ground and therefore serve as low inductance decoupling capacitors They come in sheet form and may be cut with a knife or scissors A few of the many types of Solder Mount building block components are shown in Figure 8 3 SAMPLES OF SOLDER MOUNT COMPONENTS HATTIN Figure 8 3 8 5 HARDWARE DESIGN TECHNIQUES The main advantage of Solder Mount construction over bird s nest or deadbug is that the resulting circuit is far more rigid and if desired may be made far smaller the latest Solder Mounts are for surface mount devices and allow the construction of breadboards scarcely larger than the final PC board although it is generally more convenient if the prototype is somewhat larger Solder Mount is sufficiently durable that it may be used for small quantity production as well as prototyping Both the deadbug and the Solder Mount prototyping techniques become somewhat tedious for complex analog or mixed signal circuits Larger circuits are often better prototyped using more formal layout techniques Another approach to prototyping analog circuits is to actually lay out a single or double sided board using CAD techniques PC based software layout packages offer ease of layout as well as schematic capture to verify connections Reference 9 Although most layout software has some amount of auto routing capability this feature is best left to digital designs Afte
296. o a voltage VSETPOINT VSETPOINT 1 49V 5mV C x TSETPOINT T25 1 where TSETPOINT is the desired setpoint temperature and T95 is 25 C For a 50 C high setpoint this works out to be VSETPOINT HI 1 615V For a lower setpoint of 35 C the voltage VSETPOINT LO would be 1 59V The divider resistors are then chosen to draw the required current IREF while setting the two tap voltages corresponding to VSETPOINT HI and VSETPOINT LO RTOTAL VREF IREF 2 5V IREF R1 VREF VSETPOINT HI 1 IREF 2 5 VSETPOINT HD 1 IREF R2 VSETPOINT HI VSETPOINT LO 1 IREF R3 VSETPOINT LO IREF 6 34 TEMPERATURE SENSORS In the example of the figure the resulting standard values for R1 R3 correspond to the temperature voltage setpoint examples noted above Ideal 1 values shown give resistor related errors of only 0 1 C from ideal Note that this is error is independent of the TMP12 temperature errors which are 2 C As noted above both comparators of the device need not always be used and in this case the lower comparator output is not used For a single point 50 C controller the 35 C setpoint is superfluous One resistor can be eliminated by making R2 R3 a single value of 95 3kQ and connecting pin 3 to GND Pin 6 should be left as a no connect If a greater hysteresis is desired the resistor values will be proportionally lowered It is also important to minimize potential parasitic tempe
297. ocessing Systems presented at 97th Audio Engineering Society Convention Nov 1994 HARDWARE DESIGN TECHNIQUES GENERAL REFERENCES HARDWARE DESIGN TECHNIQUES 1 2 10 11 12 Linear Design Seminar Section 11 Analog Devices Inc 1995 E S D Prevention Manual Available free from Analog Devices Inc B I amp B Bleaney Electricity amp Magnetism OUP 1957 pp 23 24 amp 52 Paul Brokaw An I C Amplifier User s Guide to Decoupling Grounding and Making Things Go Right for a Change Analog Devices Application Note Available free of charge from Analog Devices Inc Jeff Barrow Avoiding Ground Problems in High Speed Circuits R F Design July 1989 AND Paul Brokaw amp Jeff Barrow Grounding for Low and High Frequency Circuits Analog Dialogue 23 3 1989 Free from Analog Devices International EMI Emission Regulations Canada CSA C108 8 M1983 FDR VDE 0871 VDE 0875 Japan CISPR VCCI PUB 22 USA FCC 15 Part J Bill Slattery amp John Wynne Design amp Layout of a Video Graphics System for Reduced EMI Analog Devices Application Note E1309 15 10 89 Free from Analog Devices William R Blood Jr MECL System Design Handbook HB205 Rev 1 Motorola Semiconductor Products Inc 1988 RDI Wainwright 69 Madison Ave Telford PA 18969 1829 Tel 215 723 4333 Fax 215 723 4620 Wainwright Instruments GmbH Widdersberger Strasse 14 DW 8138 Andechs Frieding Germany Tel 49 8152 3162
298. ocessor but also with respect to the battery charging function see Figure 1 7 Most laptops have internal fans to cool the microprocessor when the internal temperature exceeds safe levels but the fan should only operate when necessary to conserve battery life LAPTOPS ARE GREAT BUT PRESENT SIGNIFICANT DESIGN CHALLENGES High Levels of Functionality and Performance Light Weight Longer Battery Life Fast Battery Charging Li ion Batteries Emerging as the Battery of Choice Lower Cost Figure 1 6 Cell phones and other types of hand held electronic equipment make wide use of power management techniques see Figure 1 8 Certain critical parts of a cell phone such as the oscillator and frequency synthesis circuits are generally powered by low dropout linear regulators for low noise and accuracy while high efficiency switching regulators are most often used in the high power transmitter circuits Shutdown features are also vital to preserve battery life while the phone is idle 1 5 INTRODUCTION LAPTOP COMPUTERS REQUIRE Battery Charger Circuits Switching Regulators Low Dropout Linear Regulators Temperature Sensors and Control uP Supervisory Circuits Airflow Fan Control AC DC MAY INCLUDE CHARGER Figure 1 7 CELL PHONES TRANSMITTER Switching Regulator DISPLAY Regulated Charge Pump Converter POWER UNIT Battery Charger POWER MANAGEMENT Voltage Reference Low d
299. oise immunity 6 25 TEMPERATURE SENSORS THERMAL RESPONSE IN FORCED AIR FOR SOT 23 3 35 SOT 23 3 SOLDERED TO 0 338 x 0 307 Cu PCB M Vt 2 7V TO 5V NO LOAD E eee TIME 2 2 2 2 2 2 CONSTANT SECONDS 20 Sere ostii i i TAT doses efe ds esce ee 2 227 ee eed pem M s IE FEAR 0 100 200 300 400 500 600 700 AIR VELOCITY LFPM Figure 6 28 Digital Output Temperature Sensors Temperature sensors which have digital outputs have a number of advantages over those with analog outputs especially in remote applications Opto isolators can also be used to provide galvanic isolation between the remote sensor and the measurement system A voltage to frequency converter driven by a voltage output temperature sensor accomplishes this function however more sophisticated ICs are now available which are more efficient and offer several performance advantages The TMP03 TMP04 digital output sensor family includes a voltage reference VPTAT generator sigma delta ADC and a clock source see Figure 6 29 The sensor output is digitized by a first order sigma delta modulator also known as the charge balance type analog to digital converter This converter utilizes time domain oversampling and a high accuracy comparator to deliver 12 bits of effective accuracy in an extremely compact circuit
300. oltage is not directly accessible but instead it exists in the virtual form described above It acts as it would be seen at the output of a zero impedance divider of a numeric ratio of R1 R2 which is then fed into the R3 D1 series string through a Thevenin resistance of R11 I R2 in series with D1 With the closed loop regulator at equilibrium the voltage at the virtual reference node will be R2 V V OUT 1 2 39 REFERENCES AND LOW DROPOUT LINEAR REGULATORS With minor re arrangement this can be put into the standard form to describe the regulator output voltage as R1 VOUT Vrer 1 z In the various devices of the ADP330X series the R1 R2 divider is adjusted to produce standard output voltages of 2 7 3 0 3 2 3 3 and 5 0V As can be noted from this discussion unlike a conventional reference setup there is no power wasting reference current such as used in a conventional regulator topology IpERF of Fig 2 24 In fact the Fig 2 32 regulator behaves as if the entire error amplifier has simply an offset voltage of VREF volts as seen at the output of a conventional R1 R2 divider Design Features Related to AC Performance While the above described DC performance enhancements of the ADP330X series are worthwhile the most dramatic improvements come in areas of AC related performance These improvements are in fact the genesis of the anyCAP series name Capacitive loading and the
301. oltage reference combined with a pair of matched comparators The reference provides both a constant 2 5V output and a PTAT output voltage which has a precise temperature coefficient of 5mV K and is 1 49V nominal at 25 C The comparators compare VPTAT with the externally set temperature trip points and generate an open collector output signal when one of their respective thresholds has been exceeded Hysteresis is also programmed by the external resistor chain and is determined by the total current drawn out of the 2 5V reference This current is mirrored and used to generate a hysteresis offset voltage of the appropriate polarity after a comparator has been tripped The comparators are connected in parallel which guarantees that there is no hysteresis overlap and eliminates erratic transitions between adjacent trip zones The TMP01 utilizes laser trimmed thin film resistors to maintain a typical temperature accuracy of 1 C over the rated temperature range The open collector outputs are capable of sinking 20mA enabling the TMPO1 to drive control relays directly Operating from a 5V supply quiescent current is only 500 maximum 6 31 TEMPERATURE SENSORS TMP01 SETPOINT CONTROLLER KEY FEATURES Vc 4 5 to 13 2V Temperature Output VPTAT 5mV K Nominal 1 49V Output 25 C 1 C Typical Accuracy Over Temperature Specified Operating Range 55 C to 125 C Resistor Programmable Hysteresis Resistor Programmable Setpoints
302. ombination is reduced to approximately 10 C W with a reasonable amo unt of airflow 200 linear feet per minute The curve also shows the thermal resistance with no heat sink as a function of airflow clearly indicating that a heat sink is required in order to meet the design requirements in a surface mount package 8 56 HARDWARE DESIGN TECHNIQUES AAVID 573300 HEAT SINK FOR TO 263 2 Courtesy AAVID Thermal Technologies Inc Figure 8 56 THERMAL RESISTANCE OF AAVID 573300 SURFACE MOUNT HEAT SINK VS AIRFLOW 25 261 etra o e ui 8 p gt _ PAD AREA gt 1 6in sa T C W 15 pua Courtesy AAVID Thermal Technologies Inc AAVID 573300 HEAT SINK 0 z 0 100 200 300 400 500 AIRFLOW LFPM Figure 8 57 8 57 HARDWARE DESIGN TECHNIQUES These examples illustrate the basic process of thermal design and heat sink selection Larger heat sinks may lessen or even eliminate the need for airflow However when operating heat sinks with no air flow the heatsink must be oriented such that thermal convection currents can carry the heat away from the heat sink If additional airflow is required the air must be allowed to pass freely around the heat sink with no obstruction Note that small SOIC packages are also useful in conjunction with PCB copper heatsink areas see References 6 and 7 Further information on thermal management using heat sinks can be obtained from References
303. ompany Englewood CO An excellent general purpose trade journal on issues of EMI and EMC 8 77 HARDWARE DESIGN TECHNIQUES SHIELDING CONCEPTS The concepts of shielding effectiveness presented next are background material Interested readers should consult References 1 2 and 6 cited at the end of the section for more detailed information Applying the concepts of shielding requires an understanding of the source of the interference the environment surrounding the source and the distance between the source and point of observation the receptor or victim If the circuit is operating close to the source in the near or induction field then the field characteristics are determined by the source If the circuit is remotely located in the far or radiation field then the field characteristics are determined by the transmission medium A circuit operates in a near field if its distance from the source of the interference is less than the wavelength of the interference divided by 27 or 2 If the distance between the circuit and the source of the interference is larger than this quantity then the circuit operates in the far field For instance the interference caused by a Ins pulse edge has an upper bandwidth of approximately 350MHz The wavelength of a 350MHz signal is approximately 32 inches the speed of light is approximately 12 ns Dividing the wavelength by 2r yields a distance of approximately 5 inches the bound
304. on CHARGE TERMINATION TECHNIQUES B Primary Detection of Minimum Threshold Charging Current with Cell Voltage Limited to 4 2V B Secondary TCO Absolute Temperature Cutoff Timer Accurate Control 50mV of Final Battery Voltage Required for Safety B Multiple Cell Li lon Battery Packs Require Accurate Cell Matching and or Individual Cell Monitors and Charge Current Shunts for Safety Figure 5 12 5 9 BATTERY CHARGERS Under no circumstances should a multiple cell Li Ion battery pack be constructed from individual cells without providing this voltage equalization function While the dangers of overcharging cannot be overstated undercharging a Li Ion cell can greatly reduce capacity as shown in Figure 5 13 Notice that if the battery is undercharged by only 100mV 10 of the battery capacity is lost For this reason accurate control of the final charging voltage is mandatory in Li Ion chargers EFFECT OF UNDERCHARGE ON Li lon BATTERY CAPACITY 100 CAPACITY 98 96 94 92 90 4 100 4 125 4 150 4 175 4 200 FINAL BATTERY VOLTAGE V Figure 5 13 From the above discussion it is clear that accurate control of battery voltage and current is key to proper charging regardless of cell chemistry The ADP3810 3811 series of ICs makes this job much easier to implement A block diagram of the IC is shown in Figure 5 14 Because the final voltage is critical in charging Li Ion cells the ADP3810 has precision
305. on cell 8 4V two Li Ion cells 12 6V three Li Ion cells 4 5V three NiCd NiMH cells 9 0V six NiCd NiMH cells or 13 5V nine NiCd NiMH cells In addition a pin is provided for changing the final battery voltage by up to 10 to adjust for variations in battery chemistry from different Li Ion manufacturers A functional diagram along with a typical application circuit is shown in Figure 5 21 5 18 BATTERY CHARGERS The ADP3801 and ADP3802 directly drive an external PMOS transistor Switching frequencies of the family are 200kHz ADP3801 and 500kHz ADP3802 An on chip end of charge comparator indicates when the charging current drops to below 80mA 50mA of hysteresis prevents comparator oscillation ADP3801 ADP3802 BUCK BATTERY CHARGER 5 100uH 0 10 CS CURRENT LOOP AMP SHUTDOWN UVLO RESET SD UVLO VOLTAGE 1 EOC BATTERY ADP3801 3802 COMPARATOR VOLTAGE BAT ADJUST ADJ Figure 5 21 ADP3801 ADP3802 SWITCH MODE BATTERY CHARGER KEY SPECIFICATIONS Programmable Charge Current with High Side Sensing 0 75 End of Charge Voltage Pin Programmable Battery Chemistry and Cell Number Select On Chip LDO Regulator 3 3V Drives External PMOS Transistor PWM Oscillator Frequency ADP3801 200kHz ADP3802 500kHz End of Charge Output Signal SO 16 Package Figure 5 22 5 19 BATTERY CHARGERS Both devices offer a 3 3V LDO The LDO can deliver up to
306. on of the ADP1147 is shown in Figure 3 40 Input voltage in the circuit can range from 5 2V to 14V and the output is 5V at 2A The external resistor and capacitor connected to the pin serves to control the frequency response of the voltage feedback loop The off time of the regulator is determined by the external capacitor Current feedback is obtained from the voltage developed across the external RSgENSE resistor Typical efficiency is shown in the composite Figure 3 41 where the contributions of each source of efficiency loss are given switch I R loss gate charge loss quiescent power loss and Schottky diode loss The lower curve represents the total efficiency Note that for output currents between about 100mA and 1A the power required to drive the MOSFET gate gate charge is the largest contributor to efficiency loss At the higher current levels the I R loss due to MOSFET on resistance dominates Key specifications for the ADP1147 are summarized in Figure 3 42 3 41 SWITCHING REGULATORS HIGH EFFICIENCY STEP DOWN REGULATOR USING THE ADP1147 CONTROLLER Vin 5 2V TO 14V 1 100pF IFR7204 P DRIVE L 50H RSENSE Vout 0V NORMAL ADP1147 5 0 050 5V 2A gt 1 5V SHUTDOWN SHUTDOWN O SENSE 1000 1kQ 390uF OCI SENSE 3300pF A70pF e N 30BQ040 L COILTRONICS CTX50 2MP Figure 3 40 ADP1147 TYPICAL EFFICIENCY LOSSES
307. ong distances using twisted pair shielded cables In these applications the shield again offers protection against low frequency interference and an accepted approach is to ground the shield at the driver end LF and HF ground and ground it at the receiver with a capacitor HF ground only In those applications where the length of the cable is electrically long or protection against high frequency interference is required then the preferred method is to connect the cable shield to low impedance points at both ends direct connection at the driving end and capacitive connection at the receiver Otherwise unterminated transmission lines effects can cause reflections and standing waves along the cable At frequencies of 10MHz and above circumferential 360 shield bonds and metal connectors are required to main low impedance connections to ground In summary for protection against low frequency 1MHZ electric field interference grounding the shield at one end is acceptable For high frequency interference gt 1MHz the preferred method is grounding the shield at both ends using 360 circumferential bonds between the shield and the connector and maintaining metal to metal continuity between the connectors and the enclosure Low frequency ground loops can be eliminated by replacing one of the DC shield connections to ground with a low inductance 0 01yF capacitor This capacitor prevents low frequency ground loops and shunts high frequ
308. ongest dimension of any opening to less than 1 20 wavelength Any cables wires connectors indicators or control shafts penetrating the enclosure should have circumferential metallic shields physically bonded to the enclosure at the point of entry In those applications where unshielded cables wires are used then filters are recommended at the point of shield entry Sensors and Cable Shielding The improper use of cables and their shields is a significant contributor to both radiated and conducted interference As illustrated in Figure 8 77 effective cable and enclosure shielding confines sensitive circuitry and signals within the entire shield without compromising shielding effectiveness Depending on the type of interference pickup radiated low high frequency proper cable shielding is implemented differently and is very dependent on the length of the cable The first step is to determine whether the length of the cable is electrically short or electrically long at the frequency of concern A cable is considered electrically short if the length of the cable is less than 1 20 wavelength of the highest frequency of the interference otherwise it is electrically long For example at 50 60Hz an electrically short cable is any cable length less than 150 miles where the primary coupling mechanism for these low frequency electric fields is capacitive As such for any cable length less than 150 miles the amplitude of the interference will be the
309. only on the order or 10 or so Since Q1 is driven from the collector of Q2 the relatively high base current demanded by a lateral PNP results in relatively high emitter current in Q2 or a high Iground For a typical lateral PNP based regulator operating with a 5V 150mA output Iground will be typically 18mA and can be as high as 40mA To compound the problem of high Iground in PNP LDOs there is also the spike in Iground as the regulator is operating within its dropout region Under such conditions the output voltage is out of tolerance and the regulation loop forces higher drive to the pass device in an attempt to maintain loop regulation This results in a substantial spike upward in Iground which is typically internally limited by the regulator s saturation control circuits PMOS pass devices do not demonstrate a similar current spike in Iground since they are voltage controlled But while devoid of the Iground spike PMOS pass 2 34 REFERENCES AND LOW DROPOUT LINEAR REGULATORS devices do have some problems of their own Problem number one is that high quality low Ron low threshold PMOS devices generally aren t compatible with many IC processes This makes the best technical choice for a PMOS pass device an external part driven from the collector of Q2 in the figure This introduces the term LDO controller where the LDO architecture is completed by an external pass device While in theory NMOS pass devices would of
310. onnect dual and VOUT pins in parallel as well as to the corresponding Vin and VOUT large area PCB lands 3 In cases where maximum heat dissipation is required use double sided copper planes connected with multiple vias 4 Where possible increase the thermally conducting surface area s openly exposed to moving air so that heat can be removed by convection or forced air flow if available 5 Do not use solder mask or silkscreen on the heat dissipating traces as they increase the net thermal resistance of the mounted IC package 2 45 REFERENCES AND LOW DROPOUT LINEAR REGULATORS A real life example visually illustrates a number of the above points far better than words can do and is shown in Figure 2 38 a photo of the ADP3300 1 5 square evaluation PCB The boxed area on the board represents the actual active circuit area ADP3300 EVALUATION BOARD CAPACITOR SIZE CAN MAKE A DIFFERENCE ANALOG DEVICES 5300 SANTA CLARA DMSIO TOTAL 10pF 16V BOARD SIZE TANTALUM 1 5 X 1 5 a CAPACITOR KEMET T491C SERIES Figure 2 38 In this figure a large cross section conductor area can be seen associated with pin 4 and Vout the large U shaped trace at the lower part within the boxed outline Also the effect of the anyCAP design on capacitor size can be noted from the tiny size of the C1 and C2 0 47uF input and output capacitors near the upper left of the boxed area For comparison purposes a
311. onstructed of solid semiconductor materials which exhibit a positive or negative temperature coefficient Although positive temperature coefficient devices are available the most commonly used thermistors are those with a negative temperature coefficient Figure 6 18 shows the resistance temperature characteristic of a commonly used NTC Negative Temperature Coefficient thermistor The thermistor is highly non linear and of the three temperature sensors discussed is the most sensitive RESISTANCE CHARACTERISTICS OF A 10kQ NTC THERMISTOR 40 ALPHA THERMISTOR INCORPORATED 30 RESISTANCE TEMPERATURE CURVE A 10 ko THERMISTOR 13A1002 C3 THERMISTOR RESISTANCE 20 10 0 20 40 60 80 100 TEMPERATURE Figure 6 18 The thermistor s high sensitivity typically 44 000ppm C at 25 C as shown Figure 6 19 allows it to detect minute variations in temperature which could not be observed with an RTD or thermocouple This high sensitivity is a distinct advantage over the RTD in that 4 wire Kelvin connections to the thermistor are not needed to compensate for lead wire errors To illustrate this point suppose a 10kQ NTC thermistor with a typical 25 C temperature coefficient of 44 000ppm C were substituted for the 100Q Pt RTD in the example given earlier then a total lead wire resistance of 21Q would generate less than 0 05 C error in the measurement This is roughly a factor of 500 improvement in err
312. op The capacitors should have low ESL and ESR and be rated to handle the required ripple current Low dropout linear post regulators provide both ripple reduction as well as better regulation and can be effective provided the sacrifice in efficiency is not excessive Finally it is difficult to predict the output ripple current analytically and there is no substitute for a prototype using the real world components Once the filter is proven to provide the desired ripple attenuation with some added safety margin care must be taken that parts substitutions or vendor changes are not made in the final production units without first testing them in the circuit for equivalent performance SUMMARY OF RESULTS Proper Layout and Grounding using Ground Plane Mandatory Low ESL ESR Capacitors Give Best Results External LC Filters Very Effective in Reducing Ripple Linear Post Regulation Effective for Noise Reduction and Best Regulation Completely Analytical Approach Difficult Prototyping is Required for Optimum Results Once Design is Finalized Do Not Switch Vendors or Use Parts Substitutions Without First Verifying Their Performance in Circuit High Frequency Localized Decoupling at IC Power Pins is Still Required Figure 8 41 LOCALIZED HiGH FREQUENCY POWER SUPPLY FILTERING The LC filters described in the previous section are useful in filtering switching regulator outputs However it may be desirable to place similar filters on
313. opout voltage is lowered the first term of Pp is reduced With intermediate dropout voltage rating of 1V a 1A load current will produce 1W of heat in this regulator which may require a heat sink for continuous operation It is this first term of the regulator power which usually predominates at least for loaded regulator conditions 2 26 REFERENCES AND LOW DROPOUT LINEAR REGULATORS The second term being proportional to Iground typically only 1 2 mA sometimes even less usually only becomes significant when the regulator is unloaded and the regulator s quiescent or standby power then produces a constant drain on the source VIN However it should be noted that in some types of regulators notably those which have very low pass devices such as lateral PNP transistors the Iground current under load can actually run quite high This effect is worst at the onset of regulation or when the pass device is in saturation and can be noted by a sudden Iground current spike where the current jumps upward abruptly from a lower low level All LDO regulators using bipolar transistor pass devices which can be saturated such as PNPs can show this effect It is much less severe in PNP regulators using vertical since these have a higher intrinsic and doesn t exist to any major extent in PMOS LDOs since PMOS transistors are controlled by voltage level not current In the example shown the regulator delivers 5V x 1A
314. or over an RTD 6 16 TEMPERATURE SENSORS TEMPERATURE COEFFICIENT OF 10kQ NTC THERMISTOR 60000 ALPHA THERMISTOR INCORPORATED 50000 27 RESISTANCE TEMPERATURE CURVE A 10 kQ THERMISTOR 13A1002 C3 THERMISTOR TEMPERATURE COEFFICIENT ppm C 40000 30000 20000 0 20 40 60 80 100 TEMPERATURE C Figure 6 19 However the thermistor s high sensitivity to temperature does not come without a price As was shown in Figure 6 18 the temperature coefficient of thermistors does not decrease linearly with increasing temperature as it does with RTDs therefore linearization is required for all but the narrowest of temperature ranges Thermistor applications are limited to a few hundred degrees at best because they are more susceptible to damage at high temperatures Compared to thermocouples and RTDs thermistors are fragile in construction and require careful mounting procedures to prevent crushing or bond separation Although a thermistor s response time is short due to its small size its small thermal mass makes it very sensitive to self heating errors Thermistors are very inexpensive highly sensitive temperature sensors However we have shown that a thermistor s temperature coefficient varies from 44 000 ppm C at 25 C to 29 000ppm C at 100 C Not only is this non linearity the largest source of error in a temperature measurement it also limits useful applications to very narrow temperature ranges
315. or the EOC pin for the charge completion signal In some cases the charge is continued for 30 60 minutes after EOC to top off the battery If this is desired the timer function should be started upon receiving the EOC After the allotted time the ADP3801 should be placed in shutdown to prevent constant trickle charging By using the high accuracy final battery voltage limit of the ADP3801 the circuit can guarantee safe Li Ion charging without requiring an expensive reference and amplifier 5 24 BATTERY CHARGERS REFERENCES 10 11 Bill Schweber Supervisory ICs Empower Batteries to Take Charge EDN Sept 1 1997 p 61 Doug Vargha A Designer s Guide to Battery Charging Switchover and Monitoring ED PIPS Supplement May 27 1993 p 89 Brian Kerridge Battery Management ICs EDN May 13 1993 p 100 Joe Buxton Li Ion Battery Charging Requires Accurate Voltage Sensing Analog Dialogue Vol 31 2 1997 p 3 Anne Watson Swager Smart Battery Technology Power Management s Missing Link EDN March 2 1995 p 47 ADP3810 ADP3811 Product Data Sheet Analog Devices Norwood MA http www analog com Frank Goodenough Battery Based Systems Demand Unique ICs ED July 8 1993 p 47 Pnina Dan Make the Right Battery Choice for Portables ED PIPS Supplement December 1996 p 39 ADP3801 ADP3802 Product Data Sheet Analog Devices Norwood MA http www analog com David Linden Editor Han
316. ors 8 61 HARDWARE DESIGN TECHNIQUES A DIAGNOSTIC FRAMEWORK FOR EMI Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA ANY INTERFERENCE PROBLEM CAN BE BROKEN DOWN INTO B The SOURCE of interference B The RECEPTOR of interference The PATH coupling the source to the receptor SOURCES PATHS RECEPTORS Microcontroller Radiated Microcontroller Analog EM Fields Analog Digital Crosstalk Digital Capacitive Inductive Communications Receivers ESD Conducted Communications Signal Other Electronic Transmitters Power Systems Power Ground Disturbances Lightning Figure 8 59 Interfering signals reach the receptor by conduction the circuit or system interconnections or radiation parasitic mutual inductance and or parasitic capacitance In general if the frequencies of the interference are less than 30MHz the primary means by which interference is coupled is through the interconnects Between 30MHz and 300MHz the primary coupling mechanism is cable radiation and connector leakage At frequencies greater than 300MHz the primary mechanism is slot and board radiation There are many cases where the interference is broadband and the coupling mechanisms are combinations of the above When all three elements exist together a framework for solving any EMI problem can be drawn from Figure 8 60 There are three types of interference with which t
317. ounding Philosophy for Mixed Signal Systems Electronic Design Special Analog Issue June 23 1997 p 29 Subject Index A AAVID 573300 heat sink thermal resistance vs airflow 8 57 for TO 263 8 57 AAVID 582002B12500 heat sink thermal resistance vs airflow 8 55 for TO 220 8 54 AAVID Thermal Technologies Inc 8 58 Absolute voltage output sensor 6 24 25 EMI RFI effects 6 25 with shutdown 6 23 thermal time constant 6 25 26 AD580 three terminal bandgap reference 2 5 6 Brokaw cell 2 6 AD586 buried zener reference 2 10 long term drift performance 2 14 low tolerance 2 14 AD587 buried zener noise reduction pin 2 18 AD588 buffer amplifier 2 16 buried zener reference 2 10 long term drift performance 2 14 low tolerance 2 14 AD592 current output sensors 6 21 22 AD594 Type J thermocouple 6 10 AD594 595 circuit 6 10 instrumentation amplifier thermocouple cold junction compensator 6 9 AD595 Type K thermocouple 6 10 AD596 597 monolithic set point controllers 6 10 AD620 Instrumentation Amplifier 8 86 AD688 Kelvin sensing circuit 2 16 AD780 long term drift performance 2 14 precision sigma delta ADC driver 2 23 AD815 Data Sheet Analog Devices 8 58 AD1580 shunt bandgap reference circuit 2 7 AD1582 1585 series bandgap reference circuit 2 9 connection diagram 2 9 specifications 2 8 Brokaw cell 2 9 AD3300 evaluation board 2 46 AD7416 circuit 6 36 AD7416 7417 7418 INDEX d
318. ource of the interference in meters For magnetic fields the loss depends also on the shielding material and the frequency of the interference Reflection loss for magnetic fields is given by fr o Rm dB 14 6 1010010 8 7 Hr 8 79 HARDWARE DESIGN TECHNIQUES and for plane waves gt 2 the reflection loss is given by R pw dB 168 1010210 E Eq 8 8 Absorption is the second loss mechanism in shielding materials Wave attenuation due to absorption is given by A dB 3 34 to ur f Eq 8 9 where t thickness of the shield material in inches This expression is valid for plane waves electric and magnetic fields Since the intensity of a transmitted field decreases exponentially relative to the thickness of the shielding material the absorption loss in a shield one skin depth 6 thick is 9dB Since absorption loss is proportional to thickness and inversely proportional to skin depth increasing the thickness of the shielding material improves shielding effectiveness at high frequencies Reflection loss for plane waves in the far field decreases with increasing frequency because the shield impedance Z5 increases with frequency Absorption loss on the other hand increases with frequency because skin depth decreases For electric fields and plane waves the primary shielding mechanism is reflection loss and at high frequencies the mechanism is absorption loss For these types of in
319. own in Figure 5 23 The output of the MUX is connected to an error amplifier that compares the divided down battery voltage to a 1 65V reference The accuracy of the final battery voltage is dependent upon the major functions shown in Figure 5 23 The accuracy of the reference the resistor divider and the amplifier must all be well controlled to give an overall accuracy of 0 75 The ADP3801 and 3802 are designed to charge two separate battery packs These batteries can be of different chemistries and have a different number of cells At any given time only one of the two batteries is being charged To select which battery is being monitored and therefore which battery is being charged the devices include a battery selector multiplexer as is shown in Figure 5 23 This two channel mux is designed to break before make to ensure that the two batteries are not shorted together momentarily when switching from one to the other An important feature for Li Ion battery chargers is an end of charge detect EOC The EOC signal operation is shown in Figure 5 24 When the charge current drops below 80mA for Reg 0 10 the EOC output pulls low The EOC threshold current IMIN is given by the equation 8mV IMIN 5 Ros 5 20 BATTERY CHARGERS INTERNAL MUX SELECTS FINAL BATTERY VOLTAGE BATSEL O BATA 9 i ADP3801 ADP3802 21 BAT PRG Figure 5 23 END OF CHAR
320. pecial thanks go to Wes Freeman Walter G Jung and Ed Grokulsky for thoroughly reviewing the material for content and accuracy Judith Douville compiled the index and printing was done by R R Donnelley and Sons Inc Walt Kester 1998 Copyright 1998 by Analog Devices Inc Printed in the United States of America All rights reserved This book or parts thereof must not be reproduced in any form without permission of the copyright owner Information furnished by Analog Devices Inc is believed to be accurate and reliable However no responsibility is assumed by Analog Devices Inc for its use Analog Devices Inc makes no representation that the interconnections of its circuits as described herein will not infringe on existing or future patent rights nor do the descriptions contained herein imply the granting of licenses to make use or sell equipment constructed in accordance therewith Specifications are subject to change without notice ISBN 0 916550 19 2 PRACTICAL DESIGN TECHNIQUES FOR POWER AND THERMAL MANAGEMENT SECTION 1 INTRODUCTION SECTION 2 REFERENCES AND LOW DROPOUT LINEAR REGULATORS BH Precision Voltage References BH Low Dropout Regulators SECTION 3 SWITCHING REGULATORS Applications of Switching Regulators Inductor and Capacitor Fundamentals Ideal Step Down Buck Converter Ideal Step Up Boost Converter Buck Boost Topologies Other Non Isolated Switcher Topologies Isolated Switching Regu
321. ple thermostat this allows infinite resolution of user control for control points and ON OFF hysteresis The device is placed in an air stream near the power IC such that both see the same stream of air and will thus have like temperature profiles assuming proper control of the stream This is shown in basic form by the layout diagram of Figure 6 36 6 32 TEMPERATURE SENSORS SYSTEM USE OF TMP12 AIRFLOW SENSOR PGA PACKAGE AIR FLOW PGA SOCKET POWER IC PC BOARD TMP12 B TMP12 HAS INTERNAL 250mW HEATER B TMP12 INTERNAL TEMPERATURE PROPORTIONAL TO AIR FLOW B TMP12 INTERNAL TEMPERATURE PROPORTIONAL TO POWER IC TEMPERATURE Figure 6 36 TMP12 TEMPERATURE RELATIONSHIPS 65 60 55 DIE TEMP 50 C 45 TMP12 DIE TEMP NO AIR FLOW B HIGH SET POINT 40 C LOW SET POINT D TMP12 DIE TEMP MAX AIR FLOW E SYSTEM AMBIENT TEMPERATURE 35 0 50 100 150 200 250 TMP12 Pp mW Figure 6 37 6 33 TEMPERATURE SENSORS With the TMP12 s internal 250mW heater ON and no airflow the TMP12 thermal profile will look like the curve A of Figure 6 37 and will show a 20 C rise above TA When airflow is provided this same dissipation results in a lower temperature D In programming the device for airspeed control the designer can set up to two switch points shown here symbolically by B and C which are HIGH and LOW setpoints respectively The basic idea is that when the IC
322. power continuously will require the entire heat sink situation to be re evaluated as what was adequate for 1 7W will simply not be adequate for 10W In fact the required heat sink is about 3 C W to support the 10W safely on a continuous basis which requires a much larger heat sink Note that a general overview of thermal design and heat sink selection is included in section 8 Sensing Resistors for LDO Controllers Current limiting in the ADP3310 controller is achieved by choosing an appropriate external current sense resistor Rg which is connected between the controller s VIN and IS source pins An internally derived 50 mV current limit threshold voltage appears between these pins to establish a comparison threshold for current limiting This 50mV determines the threshold where current limiting begins For a continuous current limiting a foldback mode is established with dissipation controlled by reducing the gate drive The net effect is that the ultimate current limit level is a factor of 2 3 of maximum The foldback limiting reduces the power dissipated in the pass transistor substantially To choose a sense resistor for a maximum output current Ip Rg is calculated as follows Rg 0 05 Kr I In this expression the nominal 50mV current limit threshold voltage appears in the numerator In the denominator appears a scaling factor Kp which can be either 1 0 or 1 5 plus the maximum load current For example if a
323. pplication circuit for the device is shown in Figure 3 37 and key specifications are summarized in Figure 3 38 A special boosted drive stage is used to saturate an NPN power switch providing a system efficiency higher than conventional bipolar buck switchers An external diode and capacitor provide the boosted voltage to the drive stage that is higher than the input supply voltage A shutdown signal places the device in a low power mode reducing the supply current to under 15pA The ADP3050 provides excellent line and load regulation maintaining 2 5 output voltage accuracy over an ambient operating range of 40 C to 85 C The ADP3050 package 8 pin SOIC footprint is thermally enhanced and has a junction to ambient thermal resistance of approximately 90 C W ADP3050 250kHz 1 5A BUCK REGULATOR 1N914 ViN O SWITCH 7 TO 24V ADP3050 5 100uF 1N5818 ooy Figure 3 37 For high current output switchers external power MOSFETs are often used as switches The basic buck and boost converter circuits using MOSFETs are shown in Figure 3 39 On resistances are typically 0 0060 0 10 depending upon power and efficiency requirements The MOSFETs are generally discrete devices and are rarely integrated onto the IC regulator The regulator generates the appropriate gate drive signal to the MOSFET 3 39 SWITCHING REGULATORS ADP3050 BUCK REGULATOR KEY SPECIFICATIONS Input Voltage Range 3 6V to 24V 3 3V 5V and Adjustable O
324. quired whatever the control technique Thus it tends to be necessary to implement some sort of current sensing even in VM controlled systems Now even though we speak of a CM controller as essentially controlling the inductor current more often than not the switch current is controlled instead since it is more easily sensed especially in a switching regulator and it is a representation of the inductor current for at least the on time portion of the switching cycle Rather than actually controlling the average switch current which is not the same as the average inductor current anyway it is often simpler to control the peak current which is the same for both the switch and the inductor in all the basic topologies The error between the average inductor current and the peak inductor current produces a non linearity within the control loop In most systems that is not a problem In other systems a more precise current control is needed and in such a case the inductor current is sensed directly and amplified and frequency compensated for the best response Other control variations are possible including valley rather than peak control hysteretic current control and even charge control a technique whereby the integral of the inductor current i e charge is controlled That eliminates even the phase shift of the output capacitance from the loop but presents the problem that instantaneous current is not controlled and therefore short ci
325. r a Good Prototype Figure 8 1 PROTOTYPING TECHNIQUES James Bryant Walt Kester The basic principle of a breadboard or prototype is that it is a temporary structure designed to test the performance of a circuit or system and must therefore be easy to modify 8 2 HARDWARE DESIGN TECHNIQUES There are many commercial prototyping systems but almost all of them are designed to facilitate the prototyping of digital systems where noise immunities are hundreds of millivolts or more Non copper clad Matrix board Vectorboard wire wrap and plug in breadboard systems are without exception unsuitable for high performance or high frequency analog prototyping because their resistance inductance and capacitance are too high Even the use of standard IC sockets is inadvisable in many prototyping applications An important consideration in selecting a prototyping method is the requirement for a large area ground plane This is required for high frequency circuits including switching power supplies as well as low speed precision circuits including references and low dropout linear regulators especially when prototyping circuits involving ADCs or DACs The differentiation between high speed and high precision mixed signal circuits is difficult to make For example 16 bit ADCs and DACs may operate on high speed clocks gt 10MHz with rise and fall times of less than a few nanoseconds while the effective throughput rate of the converters
326. r pole because it is a second and sometimes variable pole of a two pole system is the source of a major LDO application problem The Cy pole can strongly influence the overall frequency response of the regulator in ways that are both useful as well as detrimental Depending upon the relative positioning of the two poles in the frequency domain along with the relative value of the ESR of capacitor CT it is quite possible that the stability of the system can be compromised for certain combinations of and ESR Note that Cy is shown here as a real capacitor which is actually composed of a pure capacitance plus the series parasitic resistance ESR Without a heavy duty exercise into closed loop stability analysis it can safely be said that LDOs like other feedback systems need to satisfy certain basic stability criteria One of these is the gain versus frequency rate of change characteristic in the region approaching the system s unity loop gain crossover point For the system to be closed loop stable the phase shift must be less than 180 at the point of unity gain In practice a good feedback design needs to have some phase margin generally 45 or more to allow for various parasitic effects While a single pole system is intrinsically stable two pole systems are not necessarily so they may in fact be stable or they may also be unstable Whether or not they are stable for a given instance is highly dependent upon the specifics of their
327. r serves only to recover the energy in the leakage inductance i e that energy which is not perfectly coupled between the windings and delivering it to the load In that case the relationship between input and output voltage is given by VOUT VIN For non unity turns ratios the input output relationship is highly nonlinear due to transfer of energy occurring via both the coupling between the windings and the capacitor For that reason it is not analyzed here 3 23 SWITCHING REGULATORS SINGLE ENDED PRIMARY INDUCTANCE CONVERTER SEPIC Figure 3 23 This converter topology often makes an excellent choice in non isolated battery powered systems for providing both the ability to step up or down the voltage and unlike the boost converter the ability to have zero voltage at the output when desired The Zeta and C k converters not shown are two examples of non isolated converters which require capacitors to deliver energy from input to output i e rather than just to store energy or deliver only recovered leakage energy as the SEPIC can be configured via a 1 1 turns ratio Because capacitors capable of delivering energy efficiently in such converters tend to be bulky and expensive these converters are not frequently used ISOLATED SWITCHING REGULATOR TOPOLOGIES The switching regulators discussed so far have direct galvanic connections between the input and output Transformers can be used to supply galvani
328. r the components are placed in their desired positions the interconnections should be routed manually following good analog layout guidelines After the layout is complete the software verifies the connections per the schematic diagram net list Many design engineers find that they can use CAD techniques to lay out simple boards themselves or work closely with a layout person who has experience in analog circuit boards The result is a pattern generation tape or Gerber file which would normally be sent to a PCB manufacturing facility where the final board is made Rather than use a PC board manufacturer however automatic drilling and milling machines are available which accept the PG tape directly Reference 10 These systems produce single and double sided circuit boards directly by drilling all holes and use a milling technique to remove copper and to create insulation paths for the finished board The result is a board very similar to the final manufactured double sided PC board the chief exception being that there is no plated through hole capability and any vias between the two layers of the board must be wired and soldered on both sides Minimum trace widths of 25 mils 1 mil 2 0 001 and 12 mil spacing between traces are standard although smaller trace widths can be achieved with care The minimum spacing between lines is dictated by the size of the milling bit typically 10 to 12 mils Figures 8 4 and 8 5 show the top and bottom
329. ram 5 19 constant programmable charge current 5 20 internal multiplexer 5 18 5 20 output 5 20 21 output stage external PMOS transistor 5 22 separate battery pack charger 5 20 switch mode battery charger specifications 5 19 ADP3801 3802 Product Data Sheet 5 25 ADP3810 linear battery charger lithium ion cells external MOSFET 5 17 off line flyback battery charger diagram 5 14 lithium ion cells 5 14 overvoltage comparator 5 12 ADP3810 3811 battery charger controller IC 5 10 11 5 14 16 block diagram 5 11 circuitry performance details 5 13 current control details 5 15 current mode flyback converter topology 5 15 key features 5 11 off line charging circuit 5 16 simplified battery charger 5 12 ADP3810 3811 Product Data Sheet 5 25 ADP3820 charger lithium ion battery 5 17 linear regulator controller 5 17 18 ADR290 ADR298 series XFET reference specifications 2 12 topology characteristics 2 11 ADTO5 temperature sensor thermostatic switch 6 29 30 ADT14 setpoint controller quad 6 32 ADT20 21 22 programmable setpoint controllers internal hysteresis 6 32 ADT45 ADT50 absolute voltage output sensors 6 24 25 thermal time constant 6 25 26 70 platinum resistance temperature detector conditioning 6 14 15 ADuC810PC INDEX as MicroConverter 7 13 14 as processor 7 14 Airflow monitor using TMP12 6 32 35 Aluminum electrolytic capacitor general purpose 3 63 8 20 22 switching 3 63
330. range ADP3603 3604 3605 REGULATED 3V OUTPUT VOLTAGE INVERTERS Ron CONTROL Vout FEEDBACK V CONTROL SENSE LOOP Figure 4 17 4 16 SWITCHED CAPACITOR VOLTAGE CONVERTERS ADP3603 ADP3604 ADP3605 REGULATED INVERTERS KEY SPECIFICATIONS ADP3603 3604 ADP3605 Output Accuracy 3 2 Switching Frequency 120kHz 250kHz Turn On Turn Off Time 5ms 5ms Shutdown Current 1 4mA 10pA Output Current 50mA 120mA 120mA Quiescent Current 2 4mA 2mA Input Voltage 4 5V to 6V 4 5V to 6V Nominal Output 3V 3V Package SO 8 SO 8 Figure 4 18 ADP3603 3604 3605 APPLICATION CIRCUIT FOR 3V OUTPUT Vin 4 5V TO 6V Vout ADP3603 3604 3605 SHUTDOWN SEE TEXT Figure 4 19 SWITCHED CAPACITOR VOLTAGE CONVERTERS The regulated output voltage of the ADP3603 3604 3605 series can varied between and by connecting a resistor between the output and the VSENSE pin as shown in the diagram Regulation will be maintained for output currents up to about 30mA The value of the resistor is calculated from the following equation R VOUT 5s i The devices can be made to operate as standard inverters providing an unregulated output voltage if the VS NSE pin is simply connected to ground The ADP3607 ADP3607 5 are boost switched capacitor voltage regulators based on a regulated voltage doubling topology The ADP3607 5 is optimized for an output voltage of 5V for inp
331. rature errors associated with the TMP12 Although the open collector outputs can sink up to 20mA it is advised that currents be kept low at this node to limit any additional temperature rise The Q1 Q2 transistor buffer shown in the figure raises the current drive to 100mA allowing a 50 coil to be driven The relay type shown is general purpose and many other power interfaces are possible with the TMP12 If used as shown the relay contacts would be used to turn on a fan for airflow when the active low output at pin 7 changes indicating the upper setpoint threshold A basic assumption of the TMP12 s operation is that it will mimic another device in temperature rise Therefore a practical working system must be arranged and tested for proper airflow channeling minimal disturbances from adjacent devices etc Some experimentation should be expected before a final setup will result TMP12 50 C SETPOINT CONTROLLER 5V TMP12 D1 TO FAN OR IN4002 COOLING DEVICE vi TEMPERATURE AC 0 SENSOR AND XR R5 en ver VOLTAGE 3900 SPDT RELAY R1 om S10ko 2 5V COIL 500 MIN 1 VPTAT J 4 gt OMRON G2R 14 DC5 e 2 7Le Q Q1 Q2 2N2222 R3 FOR Tpys 2 C IREF 17pA 4 Y 5 SETPOINT HI 50 HYSTERESIS SETPOINT LO 35 C IF USED GENERATOR 1002 R1 52 3kQ R2 4 42kQ
332. rcuit protection is not inherent in the system All techniques offer various advantages and disadvantages Usually the best tradeoff between performance and cost simplicity is peak current control as used by the ADP1147 family This family also uses the current sense output to control a sleep or power saving mode of operation to maintain high efficiency for low output currents GATED OSCILLATOR PULSE BURST MODULATION CONTROL EXAMPLE All of the PWM techniques discussed thus far require some degree of feedback loop compensation This can be especially tricky for boost converters where there is more phase shift between the switch and the output voltage As previously mentioned a technique which requires no feedback compensation uses a fixed frequency gated oscillator as the switch control see Figure 3 29 This method is often incorrectly referred to as the Pulse Frequency Modulation PFM mode but is more correctly called pulse burst modulation PBM or gated oscillator control The output voltage VOUT is divided by the resistive divider R1 and R2 and compared against a reference voltage VREF The comparator hysteresis is required for stability and also affects the output voltage ripple When the resistor divider output voltage drops below the comparator threshold VREF minus the hysteresis voltage the comparator starts the gated oscillator The switcher begins switching again which then causes the output voltage to increase until
333. rd can serve as a guide to both the prototype and the final PC board layout Gerber files are generally available for all evaluation board layouts and may be obtained at no charge 8 12 HARDWARE DESIGN TECHNIQUES REFERENCES SIMULATION PROTOTYPING AND EVALUATION BOARDS 1 Paolo Antognetti and Guiseppe Massobrio Ed Semiconductor Device Modeling with SPICE McGraw Hill 1988 2 Amplifier Applications Guide Section 13 Analog Devices Inc Norwood MA 1992 3 Boyle et al Macromodelling of Integrated Circuit Operational Amplifiers IEEE Journal of Solid State Circuits Vol SC 9 no 6 December 1974 4 PSpice Simulation software MicroSim Corporation 20 Fairbanks Irvine CA 92718 714 770 3022 5 Jim Williams High Speed Amplifier Techniques Linear Technology Application Note 47 August 1991 6 Robert A Pease Troubleshooting Analog Circuits Butterworth Heinemann 1991 7 Vector Electronic Company 12460 Gladstone Ave Sylmar CA 91342 Tel 818 365 9661 Fax 818 365 5718 8 RDI Wainwright 69 Madison Ave Telford PA 18969 1829 Tel 215 723 4333 Fax 215 723 4620 Wainwright Instruments GmbH Widdersberger Strasse 14 DW 8138 Andechs Frieding Germany Tel 49 8152 3162 Fax 49 8152 40525 9 Schematic Capture and Layout Software PADS Software INC 165 Forest St Marlboro MA 01752 and ACCEL Technologies Inc 6825 Flanders Dr San Diego CA 92121 10 Prototype Board Cutters
334. reby a three resistor network is used in conjunction with two comparators and a single precision reference to check if the supply is within its required operating tolerance An added feature of this design is that the power supply voltages being monitored can be higher than the power supply voltage to the ADM9264 The allowable tolerance on the monitored voltages are as follows 12V 1V 0 5V 3 3V 0 3V 2 8V 0 2V The error output signals are available individually and are also gated into a common output PWROK Auxiliary inputs ERRX ERRY are provided which are also gated into the main PWROK signal Signals other than power supplies can be accomodated as inputs to the ADM9264 such as temperature sensors A block diagram of the ADM9264 is shown in Figure 7 8 key specifications in Figure 7 9 and an application circuit in Figure 7 10 7 6 HARDWARE MONITORING ADM9264 QUAD SUPPLY VOLTAGE MONITOR ore C O Vec PE SU2 SU3 SU4 O i ERR1 F e NZ VREF GND Figure 7 8 ADM9264 KEY SPECIFICATIONS Simultaneous Monitoring of 12V 5V 3 3V and 2 8V for Desktop PCs Limits Set at 12V 1V 5V 0 5V 3 3V 0 3V and 2 8V 0 2V Auxiliary Sensor Inputs Low Power 25pA Typical Internal Comparator Hysteresis 320mV for 12V 130mV for 5V 90mV for 3 3V and 80mV for 2 8V Power Supply Glitch Immunity 100mV 10s on Vec or SU1 SUA 2 5V to 6V Guaranteed 40 C to 85 C No
335. red to the battery during the constant current mode and the final 35 during the constant voltage mode Secondary charge termination is usually handled with a timer or if the cell temperature exceeds a maximum value TCO absolute temperature cutoff It should be emphasized that Li Ion batteries are extremely sensitive to overcharge Even slight overcharging can result in a dangerous explosion or severely decrease battery life For this reason it is critical that the final charge voltage be controlled to within about 50mV of the nominal 4 2V value 5 8 BATTERY CHARGERS Li lon FAST CHARGING CHARACTERISTICS 4 3 4 2 CELL VOLTAGE o fo mo ip ned Suman V CELL 4 0 Sees athe dis ee CURRENT CELL 4 CEL C 3 9 RUNE ARTES VOLTAGE Verspenfe Wee nerf CURRENT saint focus epe tied E tussis aan SE aet m vA ERI RERUM ETE IMIN 0 1 0 2 0 3 0 CHARGE TIME HOURS Figure 5 11 Battery packs which contain multiple Li Ion cells are generally manufactured with matched cells and voltage equalizers The external charging circuitry controls the charging current and monitors the voltage across the entire battery pack However the voltage across each cell is also monitored within the pack and cells which have higher voltage than others are discharged through shunt FETs If the voltage across any cell exceeds 4 2V charging must be terminated Li l
336. ree classes of capacitors useful in 10kHz 100MHz frequency range broadly distinguished as the generic dielectric types electrolytic film and ceramic These can in turn can be further sub divided A thumbnail sketch of capacitor characteristics is shown in the chart of Figure 3 62 3 62 SWITCHING REGULATORS CAPACITOR SELECTION Aluminum Aluminum Tantalum OS CON Polyester Ceramic Electrolytic Electrolytic Electrolytic Electrolytic Stacked Multilayer General Switching Film Purpose Type Size 100 pF 120 120 100 pF 1 pF 0 1 uF Rated 25V 25V 20 V 20 V 400 V 50V Voltage 0 60 0 18 0 0 120 0 020 0 110 0 120 ESR 100 kHz 100 kHz 100 kHz 100 kHz 1 MHz 1 MHz Operating 100 kHz 500 kHz 1 MHz 1MHz 10 MHz 1 GHz Frequency Upper frequency strongly size and package dependent Figure 3 62 The electrolytic family provides an excellent cost effective low frequency component because of the wide range of values a high capacitance to volume ratio and a broad range of working voltages It includes general purpose aluminum electrolytic types available in working voltages from below 10V up to about 500V and in size from 1 to several thousand uF with proportional case sizes All electrolytic capacitors are polarized and thus cannot withstand more than a volt or so of reverse bias without damage They also have relatively high leakage currents up to t
337. regulator By inspection of the inductor current we can write m Vsw gt E Zour VIN VD L L Jom or toff ZINC TOW VOUT VIN VD 3 54 SWITCHING REGULATORS However the switching frequency f is given by 1 E T ton toff d Ae Substituting this expression for to f in the previous equation for and solving for ton yields ae VIN VD f VoUT Vsw VD However I VIN VSW pp L on Combining the last two equations and solving for L yields L Bl VOUT VIN VD VIN VSW L for boost PWM constant f Vout Vsw Vp Ipp frequency converter For the boost converter the inductor input current I N can be related to the output current by Vi ITN Sour Jour Nominally make Ipp 0 2115 Note that for the boost PWM even though the input current is continuous while the output current pulsates we still base the inductance calculation on the peak to peak inductor ripple current As was previously suggested the actual selection of the inductor value in a switching regulator is probably the easiest part of the design process Choosing the proper type of inductor is much more complicated as the following discussions will indicate Fundamental magnetic theory says that if a current passes through a wire a magnetic field will be generated around the wire right hand rule The strength of this field is measured in amp
338. regulator must be used only with a specific size as well as type of output capacitor where the ESR is controlled with respect to both time and temperature to fully guarantee regulator stability Fortunately some recent Analog Devices LDO IC circuit developments have eased this burden on the part of the regulator user a great deal and will be discussed below in further detail Some examples of standard IC regulator architectures illustrate the points above regarding pass devices and allow an appreciation of regulator developments leading up to more recent LDO technologies The classic LM309 5V 1A three terminal regulator see Reference 1 was the originator in a long procession of regulators This circuit is shown in much simplified form in Figure 2 27 with current limiting and over temperature details omitted This IC type is still in standard production today not just in original form but in family derivatives such as the 7805 7815 etc and their various low and medium current alternates Using a Darlington pass connection for Q18 Q19 the design has never been known for low dropout characteristics 1 5V typical or for low quiescent current 5m4A It is however relatively immune to instability issues due to the internal compensation of C1 and the buffering of the emitter follower output This helps make it easy to apply The LM109 309 bandgap voltage reference actually used in this circuit consists of a more involved scheme as oppose
339. rent builds up in the primary winding and also in the secondary winding where it is transferred to the load through diode D1 When the switch is on the current in the inductor flows out of D1 from the transformer and is reflected back to the primary winding according to the turns ratio Additionally the current due to the input voltage applied across the primary inductance called the magnetizing current flows in the primary winding When the switch is opened the current in the inductor continues to flow through the load via the return path provided by diode D2 The load current is no longer reflected into the transformer but the magnetizing current induced in the primary still requires a return path so that the transformer can be reset Hence the extra reset winding and diode are needed The relationship between the input and output voltage is given by VIN p VOUT N 3 25 SWITCHING REGULATORS ISOLATED TOPOLOGY FORWARD CONVERTER BUCK DERIVED V Vour D D Duty Cycle Figure 3 25 There are many other possible isolated switching regulator topologies which use transformers however the balance of this section will focus on non isolated topologies because of their wider application in portable and distributed power systems SWITCH MODULATION TECHNIQUES Important keys to understanding switching regulators are the various methods used to control the switch For simplicity of analysis the examples previousl
340. require that hardware as well as software operate properly in spite of the many things that can cause a complex high performance system to crash or lock up The purpose of hardware monitoring is to monitor the critical items in a computing system and take corrective action should problems occur Microprocessor supply voltage and temperature are two critical parameters If the supply voltage drops below a specified minimum level further operations should be halted until the voltage returns to acceptable levels In some cases it is desirable to reset the microprocessor under brownout conditions It is also common practice to reset the microprocessor on power up or power down Switching to a battery backup may be required if the supply voltage is low Under low voltage conditions it may also be desirable to inhibit the microprocessor from writing to external CMOS memory by inhibiting the Chip Enable signal to the memory A summary of the concepts of power management thermal management and hardware monitoring is shown in Figure 1 1 1 1 INTRODUCTION POWER MANAGEMENT THERMAL MANAGEMENT AND HARDWARE MONITORING OVERVIEW B Power Management Switching Supplies Switched Capacitor Voltage Converters Battery Chargers Linear Low Dropout Regulators Voltage References Thermal Management Temperature Sensing Temperature Control Hardware Monitoring uP Supervision Supply Voltages Temperature Figure 1 1
341. resistance and as current flows in and out of them they dissipate power too Transistors bipolar or field effect are not ideal switches and have a voltage drop when they are turned on plus they cannot be switched instantly and thus dissipate power while they are turning on or off As we shall soon see switchers create ripple currents in their input and output capacitors Those ripple currents create voltage ripple and noise on the converter s input and output due to the resistance inductance and finite capacitance of the capacitors used That is the conducted part of the noise Then there are often ringing voltages in the converter parasitic inductances in components and PCB traces and an inductor which creates a magnetic field which it cannot perfectly contain within its core all contributors to radiated noise Noise is an inherent by product of a switcher and must be controlled by proper component selection PCB layout and if that is not sufficient additional input or output filtering or shielding 3 2 SWITCHING REGULATORS INTEGRATED CIRCUIT SWITCHING REGULATORS Advantages High Efficiency Small Flexible Step Up Boost Step Down Buck etc Disadvantages Noisy EMI RFI Peak to Peak Ripple Require External Components L s C s Designs Can Be Tricky Higher Total Cost Than Linear Regulators B Regulators vs Controllers Figure 3 1 Though switchers can be designed to accommodate
342. resistors R1 and R2 which are accurately trimmed for the standard Li Ion cell multiple cell voltages of 4 2V 1 cell 8 4V 2 cells 12 6V 3 cells and 16 8V 4 cells The value of the charging current is controlled by the voltage applied to the VCTRL input pin The charging current is constantly monitored by the voltage at the input pin The voltage is derived from a low side sense resistor placed in series with the battery The output of the ADP3810 OUT pin is applied to external circuitry such as a PWM which controls the actual charging current to the battery The output is a current ranging from 0 to 5mA which is suitable for driving an opto isolator in an isolated system 5 10 BATTERY CHARGERS ADP3810 3811 BLOCK DIAGRAM GND Ves Vrer Veer 1 5MO 80 O JMNA OVERVOLTAGE LOCKOUT ADP3810 ADP3811 Figure 5 14 ADP3810 3811 BATTERY CHARGER CONTROLLER KEY FEATURES Programmable Charge Current Battery Voltage Limits 4 2V 8 4V 12 6V 16 8 196 ADP3810 Adjustable ADP3811 Overvoltage Comparator 6 Over Final Voltage Input Supply Voltage Range 2 7V to 16V Undervoltage Shutdown for less than 2 7V Sharp Current to Voltage Control Transition Due to High Gain GM Stages SO 8 Package with Single Pin Compensation Figure 5 15 5 11 BATTERY CHARGERS The charging current is held constant until the battery voltage measured at the VSENSR input reaches the
343. results in a value of 5 17kQ for R The accuracy needed in the signal conditioning circuitry depends on the linearity of the network For the example given above the network shows a non linearity of 2 3 C 2 0 C The output of the network can be applied to an ADC to perform further linearization as shown in Figure 6 21 Note that the output of the thermistor network has a slope of approximately 10mV C which implies a 12 bit ADC has more than sufficient resolution 6 18 TEMPERATURE SENSORS LINEARIZED THERMISTOR AMPLIFIER 2264A Vout 0 994V T 0 C VouT 0 294V T 70 AVgu1 AT 10mV C AMPLIFIER OR ADC 10kQ NTC 5 17 THERMISTOR LINEARIZATION RESISTOR LINEARITY 2 C 0 C TO 70 C Figure 6 21 SEMICONDUCTOR TEMPERATURE SENSORS Modern semiconductor temperature sensors offer high accuracy and high linearity over an operating range of about 55 C to 150 C Internal amplifiers can scale the output to convenient values such as 10mV C They are also useful in cold junction compensation circuits for wide temperature range thermocouples All semiconductor temperature sensors make use of the relationship between a bipolar junction transistor s BJT base emitter voltage to its collector current q Ig where k is Boltzmann s constant T is the absolute temperature q is the charge of an electron and Ig is a current related to the geometry and the temperature of the transistors
344. rge valued RTDs exhibit slow response times Furthermore although the cost of RTDs is higher than that of thermocouples they use copper leads and thermoelectric effects from terminating junctions do not affect their accuracy And finally because their resistance is a function of the absolute temperature RTDs require no cold junction compensation Caution must be exercised using current excitation because the current through the RTD causes heating This self heating changes the temperature of the RTD and appears as a measurement error Hence careful attention must be paid to the design of the signal conditioning circuitry so that self heating is kept below 0 5 C Manufacturers specify self heating errors for various RTD values and sizes in still and in moving air To reduce the error due to self heating the minimum current should be used for the required system resolution and the largest RTD value chosen that results in acceptable response time Another effect that can produce measurement error is voltage drop in RTD lead wires This is especially critical with low value 2 wire RTDs because the temperature coefficient and the absolute value of the RTD resistance are both small If the RTD is located a long distance from the signal conditioning circuitry then the lead resistance can be ohms or tens of ohms and a small amount of lead resistance 6 12 TEMPERATURE SENSORS can contribute a significant error to the temperature measurement
345. ring an ADC MONITORING HIGH POWER MICROPROCESSOR OR DSP WITH TMP04 FAST MICROPROCESSOR DSP ETC IN PGA PACKAGE PGA SOCKET A PPS PC BOARD TMP04 IN SURFACE MOUNT PACKAGE Figure 6 32 Thermostatic Switches and Setpoint Controllers Temperature sensors used in conjunction with comparators can act as thermostatic switches ICs such as the ADT05 accomplish this function at low cost and allow a single external resistor to program the setpoint to 2 C accuracy over a range of 40 C to 150 C see Figure 6 33 The device asserts an open collector output when the ambient temperature exceeds the user programmed setpoint temperature The ADTO5 has approximately 4 C of hysteresis which prevents rapid thermal on off cycling The ADT05 is designed to operate on a single supply voltage from 2 7V to 6 29 TEMPERATURE SENSORS 7 0V facilitating operation in battery powered applications as well as industrial control systems Because of low power dissipation 200 3 3V self heating errors are minimized and battery life is maximized An optional internal 200kQ pull up resistor is included to facilitate driving light loads such as CMOS inputs The setpoint resistor is determined by the equation 39MO C RSET 903ko SET Topp C 2816 C The setpoint resistor should be connected directly between the pin Pin 4 and the GND pin Pin 5 If a ground plane is used the resistor may be connected direc
346. rn off again within a few microseconds however attractive such a procedure might be in terms of energy saving Regarding the second point a given reference IC may or may not be well suited for pulse loading conditions dependent upon the specific architecture Many references use low power and therefore low bandwidth output buffer amplifiers This makes for poor behavior under fast transient loads which may degrade the performance of fast ADCs especially successive approximation and flash ADCs Suitable decoupling can ease the problem but some references oscillate with capacitive loads or an additional external broadband buffer amplifier may be used to drive the node where the transients occur References like almost all other ICs today are fast migrating to such smaller packages such as SO 8 and the even more tiny SOT 23 enabling much higher circuit densities within a given area of real estate In addition to the system size reductions these steps bring there are also tangible reductions in standby power and cost with the smaller and less expense ICs CHOOSING VOLTAGE REFERENCES FOR HIGH PERFORMANCE SYSTEMS Tight Tolerance Improves Accuracy Reduces System Costs Temperature Drift Affects Accuracy Long Term Stability Low Hysteresis Assures Repeatability Noise Limits System Resolution Dynamic Loading Can Cause Errors Power Consumption is Critical to Battery Systems Tiny Low Cost Packages Increase Circuit Density Figure 2
347. rol the modulation of the switch The specific way in which the switch is modulated can be thought of separately and was just presented in the previous section In circuits using PBM for switch modulation the control technique typically used is a voltage mode hysteretic control In this implementation the switch is controlled by monitoring the output voltage and modulating the switch such that the output voltage oscillates between two hysteretic limits The ADP3000 switching regulator is an example of a regulator which combines these modulation and control techniques The most basic control technique for use with PWM is voltage mode VM control see Figure 3 27 Here the output voltage is the only parameter used to determine how the switch will be modulated An error amplifier first mentioned in the Buck Converter section monitors the output voltage its error is amplified with the required frequency compensation for maintaining stability of the control loop and the switch is modulated directly in accordance with that amplifier output The output voltage is divided down by a ratio matched resistor divider and drives one input of an amplifier A precision reference voltage VpEp is applied to the other input of the amplifier The output of the amplifier in turn controls the duty cycle of the PWM It is important to note that the resistor divider amplifier and reference are actually part of the switching regulator IC but are shown externa
348. ropout Regulators High Performance and Functionality B Small Size Light Weight Long Battery Life Fast Charging B Low Cost Figure 1 8 1 6 INTRODUCTION Temperature sensors and temperature control circuits are widespread in industrial applications such as process control In many cases the output signal levels are low level ones as in a thermocouple and low noise high gain conditioning is required before further processing Semiconductor temperature sensors are useful in many applications and offer high level output signals which reduces the burden on the signal conditioning circuitry see Figure 1 9 In addition semiconductor sensors are ideally suited to applications such as PCs because their operating temperature range power supply requirements and packaging closely match the other types of ICs in the system APPLICATIONS OF THERMAL MANAGEMENT Instrumentation Process Control IC Temperature Monitoring Airflow Control Battery Charging Heat Sink Design Figure 1 9 Finally proper hardware design techniques are critical to all modern systems Layout grounding and decoupling are extremely critical to successful system design as well as controlling EMI and RFI Also an understanding of thermal techniques for maintaining safe junction temperatures is critical due to the high power dissipated in many digital ICs Discussions regarding these practical issues conclude the book see Figure 1 10 1 7 INTR
349. rs where the switch is part of the IC The process is bipolar and this type of transistor is used as the switching element The ADP3000 and its relatives ADP1108 ADP1109 ADP1110 ADP1111 ADP1073 ADP1173 use this type of internal switch NPN SWITCHES IN IC REGULATORS ADP1108 1109 1110 1111 1073 1173 Isw V IN OUT BUCK ON VOLTAGE 1 5V 650mA BOOST ON VOLTAGE 1V I1A Figure 3 31 The diode is external to the IC and must be chosen carefully Current flows through the diode during the off time of the switching cycle This translates into an average current which causes power dissipation because of the diode forward voltage drop The power dissipation can be minimized by selecting a Schottky diode with a low forward drop 0 5V such as the 1N5818 type It is also important that the diode capacitance and recovery time be low to prevent additional power loss due to charging current and this is also afforded by the Schottky diode Power dissipation can be approximated by multiplying the average diode current by the forward voltage drop 3 34 SWITCHING REGULATORS The drop across the NPN switch also contributes to internal power dissipation The power neglecting switching losses is equal to the average switch current multiplied by the collector emitter on state voltage In the case of the ADP3000 series it is 1 5V at the maximum rated switch current of 650mA when operating in the buck mode In the boos
350. s This allows temperature measurements to be made over a range of approximately 50 C to 800 C The ADT70 contains the two matched current sources a precision rail to rail output instrumentation amplifier a 2 5V reference and an uncommitted rail to rail output op amp A shutdown function is included for battery powered equipment that reduces the quiescent current from 3mA to 10pA The gain or full scale range for the Pt RTD and ADT70 system is set by a precision external resistor connected to the instrumentation amplifier The uncommitted op amp may be used for scaling the internal voltage reference providing a Pt RTD open signal or over temperature warning providing a heater switching signal or other external conditioning determined by the user The ADT 0 is specified for operation from 40 C to 125 C and is available in 20 pin DIP and SOIC packages 6 14 TEMPERATURE SENSORS INTERFACING A Pt RTD TO A HIGH RESOLUTION ADC OR 5V DEPENDING ON ADC CONTROL REGISTER OUTPUT REGISTER Gz1 TO 128 SERIAL INTERFACE AD77XX SERIES 16 22 BITS TO MICROCONTROLLER Figure 6 16 CONDITIONING THE PLATINUM RTD USING THE ADT70 2 5V REFERENCE NU OUT 5mV C Note Some Pins Omitted 1V TO 5V for Clarity Figure 6 17 6 15 TEMPERATURE SENSORS THERMISTORS Similar in function to the RTD thermistors are low cost temperature sensitive resistors and are c
351. s however the inductor current increases exponentially corresponding to the drop in effective inductance It is therefore important in all switching regulator designs to determine the peak inductor current expected under the worst case conditions of input voltage load current duty cycle etc This worst case peak current must be less than the peak current rating of the inductor Notice that when inductor literature does not have a DC current rating or shows only an AC amps rating such inductors are often prone to saturation EFFECTS OF SATURATION ON INDUCTOR CURRENT Figure 3 56 From a simplified design standpoint the effects or presence of inductor saturation can best be observed with a scope and a current probe If a current probe is not available a less direct but still effective method is to measure the voltage across a small sense resistor in series with the inductor The resistor value should be 1Q or less depending on the inductor current and the resistor must be sized to dissipate the power In most cases 10 1W resistor will work for currents up to a few hundred mA and a 0 10 10W resistor is good for currents up to 10A Another inductor consideration is its loss Ideally an inductor should dissipate no power However in a practical inductor power is dissipated in the form of hysteresis loss eddy current loss and winding loss Figure 3 57 shows a typical B H curve for an inductor The enclosed area swept
352. s of the IC as shown External scaling resistors R1 and R2 set up the desired output voltage which is VOUT VREF 1 z 504A x R2 2 32 REFERENCES AND LOW DROPOUT LINEAR REGULATORS SIMPLIFIED SCHEMATIC OF LM317 ADJUSTABLE THREE TERMINAL REGULATOR ViN Vour R15 VREF 1 25V R1 R14 ADJ R2 VOUT VREF 1 50 x R2 R2 Y Figure 2 28 As can be noted the voltage output is a scaling of VREF by R2 R1 plus a small voltage component which is a function of the 50uA reference cell current Typically the R1 R2 values are chosen to draw gt 5mA making the rightmost term relatively small by comparison The design is internally compensated and in many applications will not necessarily need an output bypass capacitor Like the LM309 fixed voltage regulator the LM317 series has relatively high dropout voltage due to the use of Darlington pass transistors It is also not a low power IC quiescent current typically 3 5mA The strength of this regulator lies in the wide range of user voltage adaptability it allows Subsequent variations on the LM317 pass device topology modified the method of output drive substituting a PNP NPN cascade for the LM317 s Darlington NPN pass devices This development achieves a lower VMIN 1 5V or less see Reference 4 The modification also allows all of the general voltage programmability of the basic LM317 but at some potential increase in application sensitivity to output capacitance
353. s one or more internal and a second formed by external loading there is the potential for the cumulative gain phase to add in a less than satisfactory manner The potential for instability under certain output loading conditions is for better or worse a fact of life for most LDO topologies However the output capacitor which gives rise to the instability can for certain circumstances also be the solution to the same instability This seemingly paradoxical situation can be appreciated by realizing that almost all practical capacitors are actually as shown in Fig 2 29 a series combination of the capacitance and a parasitic resistance ESR While load resistance RT and CT do form a pole Cy and its ESR also form a zero The effect of the zero is to mitigate the de stabilizing effect of Cy for certain conditions For example if the pole and zero in 2 36 REFERENCES AND LOW DROPOUT LINEAR REGULATORS question are appropriately placed in frequency relative to the internal regulator poles some of the deleterious effects can be made to essentially cancel leaving little or no problematic instability see Reference 5 The basic problem with this setup is simply that the capacitor s ESR being a parasitic term is not at all well controlled As result LDOs which depend upon output pole zero compensation schemes must very carefully limit the capacitor ESR to certain zones such as shown by Figure 2 31 ZONED LOAD CAPACITOR ESR
354. s and minimize inductance Make track widths at least 200 mils for every inch of track length for lowest DCR and use 1 oz or 2 oz copper PCB traces to further reduce IR drops and inductance 3 Use short leads or better yet leadless components to minimize lead inductance This minimizes the tendency to add excessive ESL and or ESR Surface mount packages are preferred Make all connections to the ground plane as short as possible 4 Use a large area ground plane for minimum impedance 8 42 HARDWARE DESIGN TECHNIQUES 5 Know what your components do over frequency current and temperature variations Make use of vendor component models for the simulation of prototype designs and make sure that lab measurements correspond reasonably with the simulation While simulation is not absolutely necessary it does instill confidence in a design when correlation is achieved see Reference 15 The discussion above assumes that the incoming AC power is relatively clean an assumption not always valid The AC power line can also be an EMI entry exit path To remove this noise path and reduce emissions caused by the switching power supply or other circuits a power line filter is required It is important to remember that AC line power can potentially be lethal Do not experiment without proper equipment and training All components used in power line filters should be UL approved and the best way to provide this is to specify a packaged UL
355. s of the charge current setting The fact that the current remains at full charging until the battery is very close to its final voltage ensures fast charging times It should be noted however that the curves shown in Figure 5 18 reflect the performance of only 5 15 BATTERY CHARGERS the charging circuitry and not the I V characteristics when charging an actual battery The internal battery resistance will cause a more gradual decrease in charge current when the final cell voltage is reached see Figure 5 11 for example A detailed description of this off line charging circuit is contained in the ADP3810 3811 data sheet Reference 7 along with design examples for those interested CHARGE CURRENT VS VOLTAGE FOR FLYBACK CHARGER 2 IDEAL Li lon CELLS ZERO CELL RESISTANCE 1 0 0 9 0 8 lmt 07 A 06 0 5 0 4 0 3 4 5 5 5 5 6 6 5 7 7 5 8 8 5 Vout Figure 5 18 Off line chargers are often used in laptop computers as shown in Figure 5 19 Here there are many options The brick may consist of a simple AC DC converter and the charger circuit put inside the laptop In some laptops the charger circuit is part of the brick Ultimately the entire AC DC converter as well as the charger circuit can be put inside the laptop thereby eliminating the need for the brick entirely There are pros and cons to all the approaches and laptop computer designers wrestle with these tradeoffs for each new design 5 16 B
356. se AT calculate it as follows AT Ty Ta Pp x 0 154W x 165 C W 25 4 C With a maximum junction temperature of 125 C this yields a calculated maximum safe ambient operating temperature of 125 25 4 C or just under 100 C Since this temperature is in excess of the device s rated temperature range of 85 C the device will then be operated conservatively at an 85 C or less maximum ambient temperature These general procedures can be used for other devices in the series substituting the appropriate for the applicable package and applying the remaining operating conditions For reference a complete tutorial section on thermal management is contained in Chapter 8 In addition layout and PCB design can have a significant influence on the power dissipation capabilities of power management ICs This is due to the fact that the surface mount packages used with these devices rely heavily on thermally conductive traces or pads to transfer heat away from the package Appropriate PC layout techniques should then be used to remove the heat due to device power dissipation The following general guidelines will be helpful in designing a board layout for lowest thermal resistance in SOT 23 and SO 8 packages 1 PC board traces with large cross sectional areas remove more heat For optimum results use large area PCB patterns with wide and heavy 2 oz copper traces placed on the uppermost side of the PCB 2 Electrically c
357. shown in simplified form in Figure 2 5 AD1580 1 2V SHUNT TYPE BANDGAP REFERENCE HAS TINY SIZE IN SOT 23 FOOTPRINT Figure 2 5 In this circuit like transistors Q1 and Q2 form the bandgap core and are operated at a current ratio of 5 times determined by the ratio of R7 to R2 An op amp is formed by the differential pair Q3 Q4 current mirror Q5 and driver output stage Q8 Q9 In closed loop equilibrium this amplifier maintains the bottom ends of R2 R7 at the same potential As a result of the closed loop control described a basic voltage is dropped across R3 and a scaled PTAT voltage also appears as V1 which is effectively in series with The nominal bandgap reference voltage of 1 225V is then the sum of 915 Vpgpg and V1 The AD1580 is designed to operate at currents as low as 50 pA also handling maximum currents as high as 10 mA It is available in grades with voltage tolerances of 1 or 10 mV and with corresponding TC s of 50 or 100 ppm C 2 7 REFERENCES AND LOW DROPOUT LINEAR REGULATORS The AD1582 AD1585 series comprises a family of series mode IC references which produce voltage outputs of 2 5 3 0 4 096 and 5 0V Like the AD1580 the series uses a small geometry process to allow packaging within an SOT 23 The AD1582 series specifications are summarized in Figure 2 6 AD1582 AD1585 2 5 5V SERIES TYPE BANDGAP SERIES SPECIFICATIONS Vout 2 500 3 000 4 096 amp 5 000V 2 7V to 12V Supply Ra
358. sic LDO architecture of Fig 2 29 allow major improvements in terms of both DC and AC performance These developments are shown schematically in Figure 2 32 which is a simplified diagram of the Analog Devices ADP330X series LDO regulator family These regulators are also known as the anyCAP family so named for their relative insensitivity to the output capacitor in terms of both size and ESR They are available in power efficient packages such as the Thermal Coastline discussed below in both stand alone LDO and LDO controller forms and also in a wide span of output voltage options ADP330X anyCAP TOPOLOGY FEATURES IMPROVED DC amp AC PERFORMANCE OVER TRADITIONAL LDOs Vin Vout NONINVERTING WIDEBAND DRIVER Figure 2 32 Design Features Related to DC Performance One of the key differences in the ADP330X series is the use of a high gain vertical PNP pass device with all of the advantages described above with Figs 2 29 and 2 30 also see Reference 6 This allows the typical dropout voltages for the series to be on the order of 1mV mA for currents of 200mA or less It is important to note that the topology of this LDO is distinctly different from that of the generic form in Fig 2 29 as there is no obvious block The reason for this is the fact that the ADP330X series uses what is termed a merged amplifier reference design The operation of the integral amplifier and reference scheme illustrated in Fig 2 32 can
359. sic step up boost converter circuit is shown in Figure 3 14 During the switch on time the current builds up in the inductor When the switch is opened the energy stored in the inductor is transferred to the load through the diode The actual waveforms associated with the boost converter are shown in Figure 3 15 When the switch is on the voltage VIN appears across the inductor and the inductor current increases at a rate equal to VIN L When the switch is opened a voltage equal to VOUT VIN appears across the inductor current is supplied to the load and the current decays at a rate equal to VOUT VINL The inductor current waveform is shown in Figure 3 15B 3 15 SWITCHING REGULATORS BASIC STEP UP BOOST CONVERTER ERROR AMPLIFIER AND SWITCH CONTROL CIRCUIT ton tof SW ON SW OFF on off Figure 3 14 BASIC STEP UP BOOST CONVERTER WAVEFORMS Vout in iL ip lour Vsw lon loft lon A Lower Case Instantaneous Value Upper Case Average Value Figure 3 15 SWITCHING REGULATORS Note that in the boost converter the input current is continuous while the output current Figure 3 15D is pulsating This implies that filtering the output of a boost converter is more difficult than that of a buck converter Refer back to the previous discussion of buck converters Also note that the input current is the sum of the switch and diode current If a steady state condition exists see Fi
360. signers are not inclined to think of switching regulators as simply drop in solutions This presents the challenge to switching regulator manufacturers to provide careful design guidelines commonly used application circuits and plenty of design assistance and product support As the power levels increase ICs tend to grow in complexity because it becomes more critical to optimize the control flexibility and precision Also since the switches begin to dominate the size of the die it becomes more cost effective to remove them and integrate only the controller 3 1 SWITCHING REGULATORS The primary limitations of switching regulators as compared to linear regulators are their output noise EMI RFI emissions and the proper selection of external support components Although switching regulators do not necessarily require transformers they do use inductors and magnetic theory is not generally well understood However manufacturers of switching regulators generally offer applications support in this area by offering complete data sheets with recommended parts lists for the external inductor as well as capacitors and switching elements One unique advantage of switching regulators lies in their ability to convert a given supply voltage with a known voltage range to virtually any given desired output voltage with no first order limitations on efficiency This is true regardless of whether the output voltage is higher or lower than the
361. simulations such a model will behave very much like the actual op amp but not exactly The IC designer uses transistor and other device models based on the actual process upon which the component is fabricated Semiconductor manufacturers invest considerable time and money developing and refining these device models so that the IC designers can have a high degree of confidence that the first silicon will work and that mask changes costing additional time and money required for the final manufactured product are minimized However these device models are not published neither are the IC micromodels as they contain proprietary information which would be of use to other semiconductor companies who might wish to copy or improve on the design It would also take far too long for a simulation of a system containing several ICs each represented by its own micromodel to reach a useful result SPICE micromodels of analog ICs often fail to converge especially under transient conditions and multiple IC circuits make this a greater possibility 8 1 HARDWARE DESIGN TECHNIQUES For these reasons the SPICE models of analog circuits published by manufacturers or software companies are macromodels as opposed to micromodels which simulate the major features of the component but lack fine detail Most manufacturers of linear ICs including Analog Devices provide these macromodels for components such as operational amplifiers analog multipliers
362. specified in terms of wideband rms or peak to peak noise over a specified bandwidth The most useful way to specify noise as with op amps is a plot of noise voltage spectral density nV VHz versus frequency 2 16 REFERENCES AND LOW DROPOUT LINEAR REGULATORS Low noise references are important in high resolution systems to prevent loss of accuracy Since white noise is statistical a given noise density must be related to an equivalent peak to peak noise in the relevant bandwidth Strictly speaking the peak to peak noise in a gaussian system is infinite but its probability is infinitesimal Conventionally the figure of 6 6 x rms is used to define a practical peak value statistically this occurs less than 0 1 of the time This peak to peak value should be less than 1 2LSB in order to maintain required accuracy If peak to peak noise is assumed to be 6 times the rms value then for an N bit system reference voltage fullscale VREF reference noise bandwidth BW the required noise voltage spectral density Ej V VHz is given by E c REF _ 12 2N BW For a 10V 12 bit 100kHz system the noise requirement is a modest 643nV VHz Figure 2 15 shows that increasing resolution and or lower fullscale references make noise requirements more stringent The 100kHz bandwidth assumption is somewhat arbitrary but the user may reduce it with external filtering thereby reducing the noise Most good IC references have noise spectral densiti
363. ss of the conductive shield depends on two things First is the loss due to the reflection of the incident wave off the shielding material Second is the loss due to the absorption of the transmitted wave within the shielding material Both concepts are illustrated in Figure 8 74 The amount of reflection loss depends upon the type of interference and its wave impedance The amount of absorption loss however is independent of the type of interference It is the same for near and far field radiation as well as for electric or magnetic fields 8 78 HARDWARE DESIGN TECHNIQUES REFLECTION AND ABSORPTION ARE THE TWO PRINCIPAL SHIELDING MECHANISMS Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA INCIDENT RAY REFLECTED RAY TRANSMITTED RAY SHIELD MATERIAL ABSORPTIVE REGION Figure 8 74 Reflection loss at the interface between two media depends on the difference in the characteristic impedances of the two media For electric fields reflection loss depends on the frequency of the interference and the shielding material This loss can be expressed in dB and is given by Re dB 322 101og10 m Eq 8 6 where relative conductivity of the shielding material in Siemens per meter Ur relative permeability of the shielding material Henries per meter f frequency of the interference and r distance from s
364. ssor power supply voltage 1 3 4 7 1 purpose 1 1 7 1 scope 1 1 temperature 1 1 7 1 Heat sink 8 48 basics 8 51 FET pass transistor 8 53 thermal resistance case ambient measurement 8 48 junction ambient measurement 8 51 HFQ series capacitors 3 66 High efficiency switching regulators in cell phones 1 5 6 High Speed Design Techniques Analog Devices 8 13 8 17 8 88 High precision circuit 8 3 High speed circuit 8 3 Hysteretic current control 3 31 I IC regulator see Switching regulator Ice point junction 6 6 Immunity 8 66 Inductor considerations 3 48 core materials 3 59 current versus magnetic flux density 3 56 energy transfer 3 8 9 ferromagnetic core 3 56 saturation 3 57 fundamentals 3 8 10 magnetic considerations 3 55 59 magnetic core permeability 3 56 magnetic field strength oersteds 3 55 magnetic flux density gauss 3 55 manufacturer listing 3 70 power loss 3 57 58 eddy currents 3 58 magnetic hysteresis 3 58 winding resistance 3 58 power supply filters ferrites 8 23 self resonant frequency 3 59 Inductor current equations switch and diode voltage effects 3 35 36 Integrated circuit components linear regulators 1 1 switched capacitor voltage regulators 1 1 switching regulators 1 1 voltage references 1 1 Integrated circuit switching regulator see Switching regulator International EMI Emission Regulations list 8 87 J JFET transistor voltage reference 2 10 14
365. t rechargeable Battery considerations 5 1 Power line transient disturbances 8 70 72 EMI filters 8 70 71 Faraday shields 8 72 transformers 8 71 Power management cell phones 1 5 6 distributed power supply systems 1 1 integrated circuit components 1 1 localized regulators 1 1 overview 1 2 scope 1 1 Power MOSFET switches buck boost converters 3 39 3 41 selection criteria 3 40 41 Power supply analog ready 8 24 card entry filter 8 24 classical disadvantages 1 2 3 distributed advantages 1 3 analog circuits 1 3 4 EMI filtering 8 74 high frequency filtering 8 41 43 filtering layout summary 8 42 43 localized decoupling 8 42 noise reduction and filtering 8 19 44 Practical Analog Design Techniques Analog Devices 8 13 Precision voltage references 2 1 2 Pressman Abraham I 3 69 Printed circuit board see PCB Process control thermal management 1 7 Prototyping techniques 8 2 9 commercial digital systems 8 3 Pulse burst modulation 3 27 3 27 28 3 31 34 disadvantages 3 33 Pulse width modulation voltage feedback scheme 3 29 30 PWMS3845 current mode flyback controller 5 14 15 R Radiofrequency interference see RFI Ratiometric voltage output sensor 6 22 RDI Wainwright firm 8 87 Reay Richard J 2 57 Rechargeable Battery considerations 5 1 figures of merit 5 2 3 technologies 5 3 REF 195 low tolerance 2 14 Index 9 INDEX References packaging 2 2 precision voltage
366. t a few hundred kHz This impedance can be lowered with an external capacitor provided the op amp within the reference remains stable for such loading LINE SENSITIVITY Line sensitivity or regulation is usually specified in of input change and is lower than 50 86dB in the REF43 REF195 0680 and AD780 For DC and very low frequencies such errors are easily masked by noise As with op amps the line sensitivity or power supply rejection of references degrades with increasing frequency typically 30 to 50dB at a few hundred kHz For this reason the reference input should be highly decoupled LF and HF Line rejection can also be increased with a low dropout pre regulator such as one of the ADP3300 series parts Figure 2 14 summarizes the major reference specifications VOLTAGE REFERENCE DC SPECIFICATIONS TYPICAL VALUES AVAILABLE B Tolerance AD588 0 01926 AD586 AD780 REF195 0 04 B Drift TC 0586 0588 1 2 0780 ADR29X 3 ppm C Drift long term ADR29X 0 2 ppm 1000 hours AD588 25 ppm 1000 hours B Supply Range REF19X AD1582 AD1585 Vout Plus 0 5 V Load Sensitivity 100 100mohm B Line Sensitivity 50pV V 86 dB Figure 2 14 NOISE Reference noise is not always specified and when it is there is not total uniformity on how For example some devices are characterized for peak to peak noise in a 0 1 to 10Hz bandwidth while others are
367. t mode the NPN switch can be driven into saturation so the on state voltage is reduced and thus so is the power dissipation Note that in the case of the ADP3000 the saturation voltage is about 1V at the maximum rated switch current of 1A In examining the two configurations it would be logical to use a PNP switching transistor in the buck converter and an NPN transistor in the boost converter in order to minimize switch voltage drop However the PNP transistors available on processes which are suitable for IC switching regulators generally have poor performance so the NPN transistor must be used for both topologies In addition to lowering efficiency by their power dissipation the switching transistors and the diode also affect the relationship between the input and output voltage The equations previously developed assumed zero switch and diode voltage drops Rather than re deriving all the equations to account for these drops we will examine their effects on the inductor current for a simple buck and boost converter operating in the continuous mode as shown in Figure 3 32 EFFECTS OF SWITCH AND DIODE VOLTAGE ON INDUCTOR CURRENT EQUATIONS Vout Vin VOUT Vsw Vour Vp ton L L INDUCTOR CURRENT lon VIN VSW _ VOUT MN L ton L toff Figure 3 32 In the buck converter the voltage applied to the inductor when the switch is on is equal to VIN VOUT Vsw where Vgw
368. tant of a temperature sensor is defined to be the time required for the sensor to reach 63 2 of the final value for a step change in the temperature Figure 6 28 shows the thermal time constant of the ADT45 ADT50 series of sensors with the SOT 23 3 package soldered to 0 338 x 0 307 copper PC board as a function of air flow velocity Note the rapid drop from 32 seconds to 12 seconds as the air velocity increases from 0 still air to 100 LFPM As a point of reference the thermal time constant of the ADT45 ADT50 series in a stirred oil bath is less than 1 second which verifies that the major part of the thermal time constant is determined by the case The power supply pin of these sensors should be bypassed to ground with a 0 1 ceramic capacitor having very short leads preferably surface mount and located as close to the power supply pin as possible Since these temperature sensors operate on very little supply current and could be exposed to very hostile electrical environments it is important to minimize the effects of EMI RFI on these devices The effect of RFI on these temperature sensors is manifested as abnormal DC shifts in the output voltage due to rectification of the high frequency noise by the internal IC junctions In those cases where the devices are operated in the presence of high frequency radiated or conducted noise a large value tantalum electrolytic capacitor gt 2 2uF placed across the 0 1 ceramic may offer additional n
369. te equal to V L where V is the applied voltage and L is the value of the inductance This energy is stored in the inductor s magnetic field and if the switch is opened the magnetic field collapses and the inductor voltage goes to a large instantaneous value until the field has fully collapsed INDUCTOR AND CAPACITOR FUNDAMENTALS L IO dt dt L dt dt C i v 0 t 0 t Current Does Not Voltage Does Not Change Instantaneously Change Instantaneously Figure 3 7 When a current is applied to an ideal capacitor the capacitor is gradually charged and the voltage builds up linearly over time at a rate equal to I C where I is the applied current and C is the value of the capacitance Note that the voltage across an ideal capacitor cannot change instantaneously Of course there is no such thing as an ideal inductor or capacitor Real inductors have stray winding capacitance series resistance and can saturate for large currents Real capacitors have series resistance and inductance and may break down under large voltages Nevertheless the fundamentals of the ideal inductor and capacitor are critical in understanding the operation of switching regulators An inductor can be used to transfer energy between two voltage sources as shown in Figure 3 8 While energy transfer could occur between two voltage sources with a resistor connected between them the energy transfer would be inefficient due to the power loss in the resistor and t
370. ter component for switchers There are many different types of capacitors and an understanding of their individual characteristics is absolutely mandatory to the design of effective practical supply filters There are generally three classes of capacitors useful in 10kHz 100MHz filters broadly distinguished as the generic dielectric types electrolytic film and ceramic These can in turn can be further sub divided A thumbnail sketch of capacitor characteristics is shown in the chart of Figure 8 15 Aluminum Aluminum Tantalum OS CON Polyester Ceramic Electrolytic Electrolytic Electrolytic Electrolytic Stacked Multilayer General Switching Film Purpose Type Size 100 pF 120 pF 120 uF 100 pF 1 pF 0 1 pF Rated 25V 25V 20 V 20 V 400 V 50 V Voltage 0 60 0 180 0 120 0 020 0110 0 120 ESR 100 kHz 100 kHz 100 kHz 100 kHz 1 MHz 1 MHz Operating 100 kHz 500 kHz 1MHz 1MHz 10 MHz 1 GHz Frequency 8 20 Upper frequency strongly size and package dependent Figure 8 15 HARDWARE DESIGN TECHNIQUES With any dielectric a major potential filter loss element is ESR equivalent series resistance the net parasitic resistance of the capacitor ESR provides an ultimate limit to filter performance and requires more than casual consideration because it can vary both with frequency and temperature in some types Another capacitor loss element is ESL equivalent series i
371. ter with a higher precision external one The converter in question may have been trimmed during manufacture to deliver its specified performance with a relatively inaccurate internal reference In such a case using a more accurate external reference with the converter may actually introduce additional gain error For example the early AD574 had a guaranteed uncalibrated gain accuracy of 0 125 when using an internal 10V reference which itself had a specified accuracy of only 1 It is obvious that if such a device having an internal 2 23 REFERENCES AND LOW DROPOUT LINEAR REGULATORS reference which is at one end of the specified range is used with an external reference of exactly 10V then its gain will be about 1 in error THE AD780 IS IDEAL FOR DRIVING PRECISION SIGMA DELTA ADCs 5V ANALOG AVpp REF IN AD77XX ADC REF IN NE AGND NOTE ONLY REFERENCE CONNECTIONS SHOWN Figure 2 22 REFERENCES Voltage References 1 2 24 Bob Widlar New Developments in IC Voltage Regulators IEEE Journal of Solid State Circuits Vol SC 6 February 1971 Paul Brokaw A Simple Three Terminal IC Bandgap Voltage Reference IEEE Journal of Solid State Circuits Vol SC 9 December 1974 Paul Brokaw More About the AD580 Monolithic IC Voltage Regulator Analog Dialogue 9 1 1975 Dan Sheingold Section 20 2 within Analog Digital Conversion Handbook 3d Edition Prentice Hall 1986 Walt Jung
372. terference high conductivity materials such as copper or aluminum provide adequate shielding At low frequencies both reflection and absorption loss to magnetic fields is low thus it is very difficult to shield circuits from low frequency magnetic fields In these applications high permeability materials that exhibit low reluctance provide the best protection These low reluctance materials provide a magnetic shunt path that diverts the magnetic field away from the protected circuit Some characteristics of metallic materials commonly used for shielded enclosures are shown in Figure 8 75 A properly shielded enclosure is very effective at preventing external interference from disrupting its contents as well as confining any internally generated interference However in the real world openings in the shield are often required to accommodate adjustment knobs switches connectors or to provide ventilation see Figure 8 76 Unfortunately these openings may compromise shielding effectiveness by providing paths for high frequency interference to enter the instrument 8 80 HARDWARE DESIGN TECHNIQUES CONDUCTIVITY AND PERMEABILITY FOR VARIOUS SHIELDING MATERIALS MATERIAL RELATIVE RELATIVE CONDUCTIVITY PERMEABILITY Copper 1 1 Aluminum 1 0 61 Steel 0 1 1 000 Mu Metal 0 03 20 000 Conductivity Ability to Conduct Electricity Permeability Ability to Absorb Magnetic Energy Figure 8 75 ANY OPENING IN AN ENCLOS
373. the bond inside the IC it is obvious that in practical circuits the problem is even more complex and it is necessary to take extreme care to ensure that all the junction pairs in the circuitry around a thermocouple except the measurement and reference junctions themselves are at the same temperature Thermocouples generate a voltage albeit a very small one and do not require excitation As shown in Figure 6 6D however two junctions T1 the measurement junction and T2 the reference junction are involved If T2 T1 then V2 V1 and the output voltage V 0 Thermocouple output voltages are often defined with a reference junction temperature of 0 C hence the term cold or ice point junction so the thermocouple provides an output voltage of at 0 C To maintain system accuracy the reference junction must therefore be at a well defined temperature but not necessarily 0 C A conceptually simple approach to this need is shown in Figure 6 7 Although an ice water bath is relatively easy to define it is quite inconvenient to maintain Today an ice point reference and its inconvenient ice water bath is generally replaced by electronics A temperature sensor of another sort often a semiconductor sensor sometimes a thermistor measures the temperature of the cold junction and is used to inject a voltage into the thermocouple circuit which compensates for the difference between the actual cold junction temperature and its ideal value
374. the individual PC boards where the power first enters the board Of course if the switching regulator is placed on the PC board then the LC filter should be an integral part of the regulator design Localized high frequency filters may also be required at each IC power pin see Figure 8 42 This simple filter can be considered an option one which is exercised 8 41 HARDWARE DESIGN TECHNIQUES dependent upon the high frequency characteristics of the associated IC and the relative attenuation desired It uses Z1 a leaded ferrite bead such as the Panasonic EXCELSA89 providing a resistance of more than 800 at 10MHz increasing to over 100Q at 100MHz The ferrite bead is best used with a local high frequency decoupling cap right at the IC power pins such as a 0 1uF ceramic unit shown HIGH FREQUENCY LOCALIZED DECOUPLING 5V TO ADDITIONAL FROM STAGES CARD ENTRY FILTER zi LEADED FERRITE BEAD PANASONIC EXCELSA39 OPTIONAL ANALOG IC 0 1uF CERAMIC Figure 8 42 The following list summarizes the switching power supply filter layout construction guidelines which will help ensure that the filter does the best possible job 1 Pick the highest electrical value and voltage rating for filter capacitors which is consistent with budget and space limits This minimizes ESR and maximizes filter performance Pick chokes for low AL at the rated DC current as well as low DCR 2 Use short and wide PCB tracks to decrease voltage drop
375. the comparator threshold is reached VREF plus the hysteresis voltage at which time the oscillator is turned off When the oscillator is off quiescent current drops to a very low value 3 31 SWITCHING REGULATORS for example 95 the ADP1073 making PBM controllers very suitable for battery powered applications SWITCH CONTROL USING GATED OSCILLATOR PULSE BURST MODULATION PBM FIXED ON OFF FREQUENCY SWITCH GATED OSCILLATOR CONTROL SWITCHING REG R2 pa ets INDUCTOR HYSTERESIS DIODE VREF NOTE RESISTORS AMPLIFIER OSCILLATOR AND VREF INCLUDED IN SWITCHING REGULATOR IC Figure 3 29 A simplified output voltage waveform is shown in Figure 3 30 for a PBM buck converter Note that the comparator hysteresis voltage multiplied by the reciprocal of the attenuation factor primarily determines the peak to peak output voltage ripple typically between 50 and 100mV It should be noted that the actual output voltage ripple waveform can look quite different from that shown in Figure 3 30 depending on the design and whether the converter is a buck or boost A practical switching regulator IC using the PBM approach is the ADP3000 which has a fixed switching frequency of 400kHz and a fixed duty cycle of 80 This device is a versatile step up step down converter It can deliver an output current of 100mA in a 5V to 3V step down configuration and 180mA in a 2V to 3 3V step up configuration Input supp
376. the time the pump capacitor is charged by the input voltage the output capacitor C2 must supply the load current The load current flowing out of C2 causes a droop in the output voltage which corresponds to a component of output voltage ripple Higher switching frequencies allow smaller capacitors for the same amount of droop There are however practical limitations on the switching speeds and switching losses and switching frequencies are generally limited to a few hundred kHz The voltage doubler works similarly to the inverter however the pump capacitor is placed in series with the input voltage during its discharge cycle thereby accomplishing the voltage doubling function In the voltage doubler the average input current is approximately twice the average output current The basic inverter and doubler circuits provide no output voltage regulation however techniques exist to add regulated capability and have been implemented in the ADP3603 3604 3605 3607 4 1 SWITCHED CAPACITOR VOLTAGE CONVERTERS BASIC SWITCHED CAPACITOR VOLTAGE INVERTER AND VOLTAGE DOUBLER INVERTER C1 O Vout 2Vin DOUBLER Figure 4 1 There are certain advantages and disadvantages of using switched capacitor techniques rather than inductor based switching regulators An obvious key advantage is the elimination of the inductor and the related magnetic design issues In addition these converters typically have relatively low noise and m
377. ting OSC IN low selects 100ms while leaving it floating selects 1 6sec With OSC SEL low OSC IN can be driven by an external clock signal or an external capacitor can be connected between OSC IN and 7 3 HARDWARE MONITORING GND This capacitor then sets both the reset active pulse timing and the watchdog timeout period The ADM8691 series supervisory circuit contains a high degree of functionality There are many applications however where all these features are not required Figure 7 4 lists some popular supervisory products and the various functions available in each TYPICAL SUPERVISORY PRODUCTS Batt CE Power Low Watch Vcc Manual Switch Gate Fail Line dog Monitor Reset Monitor Monitor Timer Reset Gen ADM869x X X X X X X ADM1232 X X X ADM707 X X X X ADM809 810 X ADM811 812 X X Figure 7 4 The ADM9261 is a triple power supply monitor IC which allows simultaneous monitoring of a 9V and two 3 3V supplies and is designed primarily for pager systems An error signal is generated if any of the supply voltages falls below an acceptable minimum value Limits are set at 4V for the 9V supply SUI input 3 0V for the SU2 3 3V input and 2 8V for the SU3 3 3V input Power supplies greater than Vcc can be monitored because the ADM9261 has on chip thin film resistor input attenuators Key features of the design are the comparator hysteresis 3 and glitch immunity 100mV 20ys Glitch i
378. tion at low frequencies lt 1MH2 or for transients with rise and fall times greater than 300ns Most motor noise and lightning transients are in this range so isolation transformers work well for these types of disturbances Although the isolation between input and output is galvanic isolation transformers do not provide sufficient protection against extremely fast transients lt 10ns or those caused by high amplitude electrostatic discharge 1 to 3ns As illustrated in Figure 8 69 isolation transformers can be designed for various levels of differential or common 8 71 HARDWARE DESIGN TECHNIQUES mode protection For differential mode noise rejection the Faraday shield is connected to the neutral and for common mode noise rejection the shield is connected to the safety ground FARADAY SHIELDS IN ISOLATION TRANSFORMERS PROVIDE INCREASING LEVELS OF PROTECTION Reprinted from EDN Magazine January 20 1994 CAHNERS PUBLISHING COMPANY 1995 A Division of Reed Publishing USA STANDARD TRANSFORMER NO SHIELD m NOTE CONNECTION FROM SECONDARY 3 TO SAFETY GROUND TO ELIMINATE GROUND TO NEUTRAL VOLTAGE SINGLE FARADAY SHIELD L CONNECT TO SAFETY GROUND FOR 3 COMMON MODE PROTECTION SINGLE FARADAY SHIELD CONNECT TO NOISY SIDE NEUTRAL WIRE FOR DIFFERENTIAL MODE PROTECTION e TRIPLE FARADAY SHIELD CONNECT TO SAFETY GROUND FOR gt z COMMON MODE CONNECT TO NE
379. tion that all ground connections are made with direct connections to the ground plane This method works extremely well when the regulator and the load are on the same PC board and the load is distributed around the board rather than located at one specific point If the load is not distributed the connection from Vou sense should be connected directly to the load as shown by the dotted line the diagram This ensures the regulator provides the proper voltage at the load regardless of the drop in the trace connecting the pass transistor output to the load GROUNDING AND SIGNAL ROUTING FOR LOW DROPOUT REGULATOR METHOD 2 ALTERNATE CONNECTION Rs SEE TEXT IS GATE Vin ADP3310 LDO Vout LINEAR REG GND POWER MMON FORME m SHORT CONNECTION TO GROUND PLANE SHORT HEAVY TRACES Figure 8 11 Switching regulators present major challenges with respect to layout grounding and filtering The discussion above on linear regulators applies equally to switchers although the importance of DC voltage drops may not be as great There is no way to eliminate high frequency switching currents in a switching regulator since they are necessary for the proper operation of the regulator What one must do however is to recognize the high switching current paths and take proper measures to ensure that they do not corrupt circuits on other parts of the 8 15 HARDWARE DESIGN TECHNIQUES board or system Figure 8 12 shows a ge
380. tly to this plane at the closest available point The setpoint resistor can be of nearly any resistor type but its initial tolerance and thermal drift will affect the accuracy of the programmed switching temperature For most applications a 1 metal film resistor will provide the best tradeoff between cost and accuracy Once has been calculated it may be found that the calculated value does not agree with readily available standard resistors of the chosen tolerance In order to achieve a value as close as possible to the calculated value a compound resistor can be constructed by connecting two resistors in series or parallel ADT05 THERMOSTATIC SWITCH Vg 2 7V TO 7V SOT 23 5 2 C Setpoint Accuracy 4 C Preset Hysteresis Specified Operating Range 40 C to 150 C Power Dissipation 2000W 3 3V Figure 6 33 6 30 TEMPERATURE SENSORS The TMP01 is a dual setpoint temperature controller which also generates a PTAT output voltage see Figure 6 34 and 6 35 It also generates a control signal from one of two outputs when the device is either above or below a specific temperature range Both the high low temperature trip points and hysteresis band are determined by user selected external resistors TMP01 PROGRAMMABLE SETPOINT CONTROLLER TEMPERATURE V SENSOR AND VOLTAGE REFERENCE OVER WINDOW COMPARATOR UNDER HYSTERESIS VPTAT GENERATOR Figure 6 34 The TMPO01 consists of a bandgap v
381. to protect the device against accidental overload conditions For normal operation device power dissipation should be externally limited by means of heat sinking air flow etc so that junction temperatures will not exceed 125 C A capacitor C3 connected between pins 2 and 4 can be used for an optional noise reduction NR feature This is accomplished by AC bypassing a portion of the regulator s internal scaling divider which has the effect of reducing the output noise 10 dB When this option is exercised only low leakage 10 100nF capacitors should be used Also input and output capacitors should be changed to 1 and 4 7uF values respectively for lowest noise and the best overall performance Note that the noise reduction pin is internally connected to a high impedance node so connections to it should be carefully done to avoid noise PC traces and pads connected to this pin should be as short and small as possible 2 44 REFERENCES AND LOW DROPOUT LINEAR REGULATORS LDO Regulator Thermal Considerations To determine a regulator s power dissipation calculate it as follows Pp VIN VoUTY IL VIN Iground where Ij and Iground are load and ground current and Vy and the input and output voltages respectively Assuming I 50mA Iground 0 5mA VIN 8V and VoyT 5V the device power dissipation is Pp 8 5 0 05 8 0 0005 0 150 004 0 154 W To determine the regulator s temperature ri
382. tor C1 switched continuously between the source V1 and C2 in parallel with the load The conditions shown are after a steady state condition has been reached The charge transferred each cycle is Aq C1 V1 V2 This charge is transferred at the switching frequency f This corresponds to an average current current charge transferred per unit time of I f Aq 1 V2 or V1 V2 CONTINUOUS SWITCHING STEADY STATE CHARGE TRANSFERRED CYCLE C1 V1 V2 CHARGE TRANSFERRED f C1 V1 V2 V1 V2 V1 V2 TIME R v D ID Figure 4 7 Notice that the quantity 1 f C1 can be considered an equivalent resistance R connected between the source and the load The power dissipation associated with this virtual resistance R is essentially forced to be dissipated in the switch on resistance and the capacitor ESR regardless of how low those values are reduced It should be noted that capacitor ESR and the switch on resistance cause additional power losses as will be discussed shortly In a typical switched capacitor voltage inverter a capacitance of 10 switched at 100kHz corresponds to R 10 Obviously minimizing R by increasing the frequency minimizes power loss in the circuit However increasing switching frequency tends to increase switching losses The optimum switched capacitor operating frequency is therefore highly process and device dependent Therefore
383. unce 1 4 mils copper traces separated by 0 021 FR 4 e 4 7 dielectric material The characteristic impedance and one way transit time of such a signal trace would be 88Q and 1 7ns ft 7 ns respectively 8 76 HARDWARE DESIGN TECHNIQUES REFERENCES ON EMI RFI 10 11 EDN s Designer s Guide to Electromagnetic Compatibility EDN January 20 1994 material reprinted by permission of Cahners Publishing Company 1995 Designing for EMC Workshop Notes Kimmel Gerke Associates Ltd 1994 Systems Application Guide Chapter 1 pg 21 55 Analog Devices Incorporated Norwood MA 1994 Henry Ott Noise Reduction Techniques In Electronic Systems Second Edition New York John Wiley amp Sons 1988 Ralph Morrison Grounding And Shielding Techniques In Instrumentation Third Edition New York John Wiley amp Sons 1986 Amplifier Applications Guide Chapter XI pg 61 Analog Devices Incorporated Norwood MA 1992 B Slattery and J Wynne Design and Layout of a Video Graphics System for Reduced EMI Analog Devices Application Note AN 333 Paul Brokaw An IC Amplifier User Guide To Decoupling Grounding And Making Things Go Right For A Change Analog Devices Application Note Order Number E1393 5 590 A Rich Understanding Interference Type Noise Analog Dialogue 16 3 1982 pp 16 19 A Rich Shielding and Guarding Analog Dialogue 17 1 1983 pp 8 13 EMC Test amp Design Cardiff Publishing C
384. unction is to cascade two switching regulators a boost regulator followed by a buck regulator as shown in Figure 3 22 The example shows some practical voltages in a battery operated system The input from the four AA cells can range from 6V charged to about 3 5V discharged The intermediate voltage output of the boost converter is 8V which is always greater than the input voltage The buck regulator generates the desired 5V from the 8V intermediate voltage The total efficiency of the combination is the product of the individual efficiencies of each regulator and can be greater than 85 with careful design An alternate topology is use a buck regulator followed by a boost regulator This approach however has the disadvantage of pulsating currents on both the input and output and a higher current at the intermediate voltage output 3 22 SWITCHING REGULATORS CASCADED BUCK BOOST REGULATORS EXAMPLE VOLTAGES INTERMEDIATE Vine VOLTAGE 4 AA CELLS Vout 8 5V 3 5 6V BOOST BUCK REGULATOR REGULATOR Figure 3 22 OTHER NON ISOLATED SWITCHER TOPOLOGIES The coupled inductor single ended primary inductance converter SEPIC topology is shown in Figure 3 23 This converter uses a transformer with the addition of capacitor Cc which couples additional energy to the load If the turns ratio N the ratio of the number of primary turns to the number of secondary turns of the transformer in the SEPIC converter is 1 1 the capacito
385. urrent handling capability of the MOSFET This current must be supplied by the input power supply and adds to the overall regulator power dissipation It can be a significant contributor to efficiency reduction up to 2 or 396 for output currents of 100 to 200mA Note that gate charge loss increases directly with both input voltage and operating frequency This is the principal reason why highest efficiency circuits which utilize this topology operate at moderate frequencies of 200kHz or less Furthermore it argues against using a larger MOSFET than necessary to control on resistance I2R loss at the maximum expected output current 3 40 SWITCHING REGULATORS POWER MOSFET SWITCHES isw BUCK Vout BOOST NCH DRIVE Figure 3 39 Power MOSFET switches allow current levels greater than 1A at high efficiencies greater than 90 using ICs such as the ADP1147 buck converter controller The input voltage for the ADP1147 can range from 3 5V to 14V Two output voltage versions are available 3 3 ADP1147 3 3 ADP1147 5 The ADP1147 regulator controller operates in a constant off time variable frequency control mode with current mode control Operating in the constant off time mode maintains constant inductor ripple current thereby easing the output filter design High efficiency is maintained at low output currents by switching automatically into a power saving PBM mode A typical step down applicati
386. us while the input current is pulsating Obviously this has implications regarding input and output filtering If one is concerned about the voltage ripple created on the power source which supplies a buck converter the input filter capacitor not shown is generally more critical that the output capacitor with respect to ESR ESL 3 11 SWITCHING REGULATORS If a steady state condition exists see Figure 3 11 the basic relationship between the input and output voltage may be derived by inspecting the inductor current waveform and writing VIN VOUT VOUT OUT L oU ton n toff Solving for VOUT t Vout VIN VIN D ton toff where D is the switch duty ratio more commonly called duty cycle defined as the ratio of the switch on time ton to the total switch cycle time ton 6 This is the classic equation relating input and output voltage in a buck converter which is operating with continuous inductor current defined by the fact that the inductor current never goes to zero INPUT OUTPUT RELATIONSHIP FOR BUCK CONVERTER on lott iL lour Write by Inspection from Inductor Output Current Waveforms toff ViN V V m VOUT ston Rearrange and Solve for t E Vout VIN 22 Vin D ton toff Figure 3 11 Notice that this relationship is independent of the inductor value L as well as the switching frequency 1 toy 1 and the load current Decreasi
387. useful in controlling resonant peaks In most electrolytic capacitors ESR degrades noticeably at low temperature by as much as a factor of 4 6 times at 55 C vs the room temperature value For circuits where ESR is critical to performance this can lead to problems Some specific electrolytic types do address this problem for example within the HFQ switching types the 10 ESR at 100kHz is more than 2x that at room temperature The OSCON electrolytics have a ESR vs temperature characteristic which is relatively flat As noted all real capacitors have parasitic elements which limit their performance The equivalent electrical network representing a real capacitor models both ESR 8 22 HARDWARE DESIGN TECHNIQUES and ESL as well as the basic capacitance plus some shunt resistance see Figure 8 16 In such a practical capacitor at low frequencies the net impedance is almost purely capacitive At intermediate frequencies the net impedance is determined by ESR for example about 0 12Q to 0 4Q at 125kHz for several types Above about 1MHz these capacitor types become inductive with impedance dominated by the effect of ESL All electrolytics will display impedance curves similar in general shape to that of Figure 8 17 The minimum impedance will vary with the ESR and the inductive region will vary with ESL which in turn is strongly effected by package style CAPACITOR EQUIVALENT CIRCUIT AND PULSE RESPONSE i Ipeak 1
388. uses the difference in the J1 J2 pinchoff voltages to differ by 500mV With the pinchoff voltage of two such FETs purposely skewed a differential voltage will appear between the gates for identical current drive conditions and equal source voltages This voltage AVp is AVP where Vp Vpo are the pinchoff voltages of FETs J1 and J2 respectively 2 10 REFERENCES AND LOW DROPOUT LINEAR REGULATORS ADR290 ADR293 2 048 5V XFET REFERENCE TOPOLOGY FEATURES HIGH STABILITY AND LOW POWER VOUT Ave ipud 4 IPTAT R3 Vour IPTAT Figure 2 9 Note that within this circuit the voltage AVp exists between the gates of the two FETs We also know that with the overall feedback loop closed the op amp axiom of zero input differential voltage will hold the sources of the two JFET at same potential These source voltages are applied as inputs to the op amp the output of which drives feedback divider R1 R3 As this loop is configured it stabilizes at an output voltage from the R1 R2 tap which does in fact produce the required AVp between the J1 J2 gates In essence the op amp amplifies AVp to produce Vout where a R VOUT avp 1 IpTAT R3 As can be noted this expression includes the basic output scaling leftmost portion of the right terms plus a rightmost temperature dependent term including IpTAT The portion of the expression compensates for a basic negative temperatur
389. ut conditions It is also common practice to reset the microprocessor on power up or power down Switching to a battery backup may be required if the supply voltage is low Under low voltage conditions it is mandatory to inhibit the microprocessor from writing to external CMOS memory by inhibiting the Chip Enable signal to the external memory Many microprocessors can be programmed to periodically output a watchdog signal Monitoring this signal gives an indication that the processor and its software are functioning properly and that the processor is not stuck in an endless loop The need for hardware monitoring has resulted in a number of ICs traditionally called microprocessor supervisory products which perform some or all of the above functions These devices range from simple manual reset generators with debouncing to complete microcontroller based monitoring sub systems with on chip temperature sensors and ADCs The ADM8691 series see Figures 7 1 and 7 2 are examples of traditional microprocessor supervisory circuits Comparator accuracy and glitch immunity is key to the circuit s operation The ADM8691 series provides the following functionality 1 Power on reset output during power up power down and brownout conditions Circuitry remains operational with Vcc as low as 1V 2 Battery backup switching for CMOS RAM CMOS microprocessor or other low power logic 3 A reset pulse is generated by the optional watchdog timer if th
390. ut a Type J thermocouple is more sensitive Second the Seebeck coefficients provide a quick guide to a thermocouple s linearity Using Figure 6 5 the system designer can choose a Type K thermocouple for its linear Seebeck coefficient over the range of 400 C to 800 C or a Type S over the range of 900 C to 1700 C The behavior of a thermocouple s Seebeck coefficient is important in applications where variations of temperature rather than absolute magnitude are important These data also indicate what performance is required of the associated signal conditioning circuitry To use thermocouples successfully we must understand their basic principles Consider the diagrams in Figure 6 6 THERMOCOUPLE BASICS A THERMOELECTRIC VOLTAGE C THERMOCOUPLE MEASUREMENT Metal A Metal A V1 V2 Metal A Thermoelectric V1 v2 EMF Metal B Y B THERMOCOUPLE D THERMOCOUPLE MEASUREMENT Copper Copper Metal A R Metal A Metal A Metal A V1 T1 T2 V2 V1 V2 Metal B Metal B R Total Circuit Resistance I V1 V2 R E V V1 V2 If T3 4 Figure 6 6 If we join two dissimilar metals at any temperature above absolute zero there will be a potential difference between them their thermoelectric e m f or contact potential which is a function of the temperature of the junction Figure 6 6A If we join the two wires at two places two junctions are formed Figure 6 6B If the two junctions are at different temperatures th
391. ut on a supply for example The general expression for VouyT is shown the figure where is the reference voltage Amplifier standby current can be further reduced below 20 if an amplifier from the OP181 281 481 or the OP193 293 498 series is used This choice will be at some expense of current drive but can provide very low quiescent current if necessary All devices shown operate from voltages down to 3V except the OP279 which operates at 5V REFERENCE PULSE CURRENT RESPONSE The response of references to dynamic loads is often a concern especially in applications such as driving ADCs and DACs Fast changes in load current invariably perturb the output often outside the rated error band For example the reference input to a sigma delta ADC may be the switched capacitor circuit shown in Figure 2 19 The dynamic load causes current spikes in the reference as the capacitor is charged and discharged As a result noise may be induced on the ADC reference circuitry 2 20 REFERENCES AND LOW DROPOUT LINEAR REGULATORS SWITCHED CAPACITOR INPUT OF SIGMA DELTA ADC PRESENTS A DYNAMIC LOAD TO THE VOLTAGE REFERENCE hii SIGMA DELTA ADC Figure 2 19 Although sigma delta ADCs have an internal digital filter transients on the reference input can still cause appreciable conversion errors Thus it is important to maintain a low noise transient free potential at the ADC s reference input Be
392. utput Versions 0 50 Saturating NPN Switch 250kHz Switching Frequency Current Mode Control Cycle by Cycle Current Limit Shutdown Feature Reduces Current to 15yA 8 Pin SOIC Thermally Enhanced Package 90 C W Figure 3 38 The main selection criteria for the power MOSFET is the peak current rating threshold voltage and the on resistance The minimum regulator input voltage determines whether a standard threshold or logic level threshold MOSFET must be used For input voltages greater than 8V a standard threshold MOSFET with a threshold voltage of less than 4V can be used If the input voltage is expected to drop below 8V a logic level MOSFET is recommended In applications involving high current outputs and input voltages less than 8V it may be necessary to drive the MOSFET gates with circuits which operate on a higher voltage such as 12V If this voltage is not available in the system it can be derived from the input voltage using charge pump techniques described in a later section since the current requirements of the drive circuits are typically fairly low The I2R loss in this type of regulator can be quite low because of the low MOSFET on resistance however one source of internal power dissipation which must not be overlooked is the gate charge required to turn the MOSFET on and off The gate drive signal must overcome the gate capacitance typically 1000 to 3000pF and is directly proportional to physical size and the c
393. uts between 3V and 5V The ADP3607 output is adjustable with an external resistor A block diagram is shown in Figure 4 20 and key specifications in Figure 4 21 The device uses a feedback control scheme similar to the ADP3603 3604 3605 to maintain output voltage regulation for VOUT lt 2VIN ADP3607 SWITCHED CAPACITOR BOOST REGULATOR Cp Cp 8 Vout FEEDBACK VsENSE CONTROL LOOP 7 GND Figure 4 20 4 18 SWITCHED CAPACITOR VOLTAGE CONVERTERS ADP3607 ADP3607 5 BOOST REGULATOR KEY SPECIFICATIONS Input Voltage Range 3V to 5V Output Voltage 5V ADP3607 5 Adjustable Output Voltage ADP3607 lt 2Vin Output Current 50mA Accuracy 296 Switching Frequency 250kHz Quiescent Current 2mA Shutdown Current 10 Turn On Turn Off Time 50ps Package 8 Pin SOIC Figure 4 21 A typical application circuit is shown in Figure 4 22 The Schottky diode connecting the input to the output is required for proper operation during start up and shutdown If VSENSE is connected to ground the devices operate as unregulated voltage doublers The output voltage of each device can be adjusted with an external resistor The equation which relates output voltage to the resistor value for the ADP3607 is given by VOUT 1V for Vout lt 2VIN The ADP3607 should be operated with an output voltage of at least 3V in order to maintain regulation Although the ADP3607 5 is optimized for a
394. verse breakdown zener 2 3 Distance from source electric field strength 8 66 Dobkin Robert C 2 57 Drift voltage references 2 14 15 Dropout voltage 2 25 E Early effect 6 19 Eddy currents 3 58 EDN s Designer s Guide to Electromagnetic Compatibility 8 77 8 86 Effective temperature differential 8 45 Electromagnetic compatibility 8 59 77 Electromagnetic interference see EMI Embedding 8 75 EMC Design Workshop Notes 8 44 EMC EMI regulations design impact 8 61 EMC Test amp Design 8 77 EMI frequency 8 64 model FAT ID 8 63 frequency amplitude time impedance distance 8 63 source receptor path 8 61 62 passive components 8 64 65 path cable radiation 8 62 conduction 8 62 connector leakage 8 62 Index 5 INDEX interconnects 8 62 radiation 8 62 slot and board radiation 8 62 regulations 8 59 61 automotive equipment 8 61 commercial equipment conducted interference 8 59 FCC VDE VCCI 8 59 radiated emissions 8 59 computers 8 60 industrial and process control equipment 8 60 61 medical equipment 8 60 military equipment 8 60 signal rise time 8 64 source physical aspects of equipment 8 64 source path receptor 8 19 type circuit system emission conduction radiation 8 62 63 circuit system immunity susceptibility 8 63 internal 8 63 EMI RFI considerations 8 59 77 diagnosis 8 61 64 Energy transfer using inductor 3 8 9 Erisman Brian 3 1 4 1 E S D Prevention Manual
395. w noise linear power supplies During the last decade however switching power supplies have become much more common in electronic systems As a consequence they also are being used for analog supplies Good reasons for the general popularity include their high efficiency low temperature rise small size and light weight In spite of these benefits switchers do have drawbacks most notably high output noise This noise generally extends over a broad band of frequencies resulting in both conducted and radiated noise as well as unwanted electric and magnetic fields Voltage output noise of switching supplies are short duration voltage transients or spikes Although the fundamental switching frequency can range from 20kHz to 1MHz the spikes can contain frequency components extending to 100MHz or more While specifying switching supplies in terms of RMS noise is common vendor practice as a user you should also specify the peak or p p amplitudes of the switching spikes with the output loading of your system The following section discusses filter techniques for rendering a switching regulator output analog ready that is sufficiently quiet to power precision analog circuitry with relatively small loss of DC terminal voltage The filter solutions presented are generally applicable to all power supply types incorporating switching element s in their energy path This includes various DC DC converters as well as popular 5V PC type supplies An
396. w is to allow high power dissipation levels while maintaining safe junction temperatures There are many tradeoffs which can be made between airflow and heat sink area and this section examines some of them A thermal model of an IC and a heat sink is shown in Figure 8 50 The critical parameter is the junction temperature Ty which must be kept below 150 C for most ICs The model shows the various thermal resistances and temperatures at various parts of the system TA is the ambient temperature Tg is the heat sink temperature is the IC case temperature and Ty is the junction temperature The heat sink is usually attached to the IC in such a manner as to minimize the difference between the IC case temperature and the heat sink temperature This is accomplished by a variety of means including thermal grease machined surface contact area etc In any case the thermal resistance between the heat sink and the IC case can usually be made less than 1 C W HEAT SINK BASICS Y T Ts 0sA Oya 9cs Osa Ts Tae To Te 0 JA 9 c Te T sa 9c Pp DEVICE POWER DISSIPATION 9c MAXIMUM JUNCTION TEMPERATURE T TA MAX z MAXIMUM AMBIENT TEMPERATURE J TJMAX TA MAX oja 4 m MAX Figure 8 50 The junction to ambient thermal resistance 0J A is therefore the sum of the three thermal resistance terms 9JA
397. y With 5mA of output current the output stage can drive a variety of optocouplers A current mode flyback converter topology is used on the secondary side Only a single diode is needed for rectification and no filter inductor is required The diode also prevents the battery from back driving the charger when input power is disconnected The Reg resistor senses the average current which is controlled via the Vcg input The Vcc source to the ADP3810 3811 can come from a direct connection to the battery as long as the battery voltage remains below the specified 16V operating range If the battery voltage is less than 2 7V e g with a shorted battery or a battery discharged below its minimum voltage the ADP3810 3811 will be in Undervoltage Lock Out UVLO and will not drive the optocoupler In this condition the primary PWM circuit will run at its designed current limit The Vcc of the ADP3810 3811 is boosted using the additional rectifier and 3 3V zener diode This circuit keeps above 2 7V as long as the battery voltage is at least 1 5V with a programmed charge current of 0 1A For higher programmed charge current the battery voltage can drop below 1 5V and is still maintained above 2 7V The charge current versus charge voltage characteristics for three different charge current settings are shown in Figure 5 18 The high gain of the internal amplifiers ensures the sharp transition between current mode and voltage mode regardles
398. y OUT Lower Case Instantaneous Value ip lour OUT Upper Case Average Value 0 Figure 3 17 The inductor L resonates with the stray switch capacitance and diode capacitance Cow Cp as in the case of the buck converter The ringing is dampened by circuit resistances and if needed a snubber The current at which a boost converter becomes discontinuous can be derived by observing the inductor current same as input current waveform of Figure 3 18 3 18 SWITCHING REGULATORS BOOST CONVERTER POINT OF DISCONTINUOUS OPERATION INDUCTOR CURRENT AND INPUT CURRENT IPEAK Vin VOUT DISCONTINUOUS MODE IF VOUT VIN 1 lin lt et IN lt gt 2L off ViN Vour 1 loUT 2 y VouT 21 ton Figure 3 18 The average input current at the point of discontinuous operation is IIN IPEAK 2 Discontinuous operation will occur if IIN lt IPEAK 2 However I Vi V IIN PEAK our IN e toff Also VIN IIN VOUT IOUT and therefore V V IN Vout VIN us VOUT VOUT 2L However VOUT _ 1 _ 1 _ ton toff VIN 1 D 1 ton toff ton toff 3 19 SWITCHING REGULATORS Solving for ___VIN f VOUT an ton toff Substituting this value for tofr into the previous expression for Ip the criteria for discontinuous operation of a boost converter is established VIN VQUT VIN 9 Criteria for discontinuous
399. y discussed used a simple fixed frequency pulse width modulation PWM technique There can be two other standard variations of the PWM technique variable frequency constant on time and variable frequency constant off time In the case of a buck converter using a variable frequency constant off time ensures that the peak to peak output ripple current also the inductor current remains constant as the input voltage varies This is illustrated in Figure 3 26 where the output current is shown for two conditions of input voltage Note that as the input voltage increases the slope during the on time increases but the on time decreases thereby causing the frequency to increase Constant off time control schemes are popular for buck converters where a wide input voltage range must be accomodated The ADP1147 family implements this switch modulation technique 3 26 SWITCHING REGULATORS CONTROL OF BUCK CONVERTER USING CONSTANT OFF TIME VARIABLE FREQUENCY PWM iL lout gt Vin Vout iL lout WAVEFORMS Vin VOUT Vour L VOUT PEAK TO PEAK LARGER L 2 RIPPLE VIN Figure 3 26 In the case of a boost converter however neither input ramp slopes nor output ramp slopes are solely a function of the output voltage see Figure 3 15 so there is no inherent advantage in the variable frequency constant off time modulation method with respect to maintaining constant output ripple current St
400. y reversible by flipping connections and reversing the drive current However a basic limitation of all shunt regulators is that load current must always be less usually appreciably less than the driving current Ip SIMPLE DIODE REFERENCE CIRCUITS Vs D1 o FORWARD BIASED ZENER AVALANCHE DIODE DIODE Figure 2 2 In the second circuit of Figure 2 2 a zener or avalanche diode is used and an appreciably higher output voltage realized While true zener breakdown occurs below 5V avalanche breakdown occurs at higher voltages and has a positive temperature coefficient Note that diode reverse breakdown is referred to almost universally today as zener even though it is usually avalanche breakdown With a D1 breakdown voltage in the 5 to 8V range the net positive TC is such that it equals the negative TC of forward biased diode D2 yielding a net TC of 100ppm C or less with proper bias current Combinations of such carefully chosen diodes formed the basis of the early single package temperature compensated zener references such as the 1N821 1N829 series The temperature compensated zener reference is limited in terms of initial accuracy since the best TC combinations fall at odd voltages such as the 1N829 s 6 2V And 2 3 REFERENCES AND LOW DROPOUT LINEAR REGULATORS the scheme is also limited for loading since for best TC the diode current must be carefully controlled Unlike a fundamentally lower voltage lt

Power and Thermal Management Design - Sylvain LARRIBE

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