In-Fixture Measurements Using Vector Network Analyzers
Contents
1. Coaxial connectors Figure 4 This is an example of a typical fixture used in the R amp D application It incorpo rates calibration standards and has a section where the DUT can be attached Removing fixture errors There are three fundamental techniques for removing errors introduced by a fixture modeling de embedding and direct measurement Each has relatively simple and more complicated versions that require greater work but yield more accurate measurements The relative performance of the fixture compared to the specifications of the DUT being measured will determine what level of cali bration is required to meet the necessary measurement accuracy Calibration based on modeling uses mathematical corrections derived from an accurate model of the fixture Often the fixture is measured as part of the process of providing an accurate model Modeling Port extensions 1 4 assume no loss flat magnitude Two port calibration linear phase Mathematically extend constant impedance reference plane 1 P a De embedding Accus external software required gt lt Two port calibration Accurate S parameter data from model or measurement Figure 5 Modeling requires that we have data regarding the fixture characteristics The simplest way to use this data is with the port extension feature of the network analyzer First you pe2. 11 Characterizing calibration standards for SOLT calibration Short Open Load Thru SOLT Calibration SOLT calibration is attractive for RF fixtures e simpler and less expensive fixtures and standards relatively easy to make broadband calibration standards e short thru are easiest e open requires characterization load is hardest quality determines corrected directivity Figure 10 Most network analyzers already contain standard calibration kit definition files that describe the characteristics of a variety of calibration standards These cal ibration kit definitions usually cover the major types of coaxial connectors used for component and circuit measurements for example Type N 7 mm 3 5 mm and 2 4 mm Most high performance network analyzers allow the user to modify the definitions of the calibration standards This capability is especially impor tant for fixture based measurements because the in fixture calibration stan dards rarely have the same attributes as the coaxial standards Custom calibration standards such as those used with fixtures require the user to characterize the standards and enter the definitions into the network analyzer The calibration kit definition must match the actual standards for accurate mea surements Definitions of the in fixture calibration standards can be stored in the analyzer as a custom user defined calibration kit While there are many characteristics used to desc
3. We can distinguish between capacitive and inductive mismatches and see non Z transmission lines TDR can help us determine the magnitude of and distance to reflections of the fix ture and the calibration standards Once the fixture has been designed and fab ricated we can use TDR to effectively evaluate how well we have minimized reflections Using TDR to Evaluate Fixture and Standards what is TDR e time domain reflectometry e analyze impedance versus time e distinguish between inductive and capacitive transitions with gating analyze transitions e analyzer standards inductive transition Zo impedance time capacitive transition ENS non Zo transmission line Figure 16 TDR measurements using a vector network analyzer start with a broadband sweep in the frequency domain The inverse Fourier transform is used to trans form the frequency domain data to the time domain yielding TDR measure ments The spatial resolution is inversely proportional to the frequency span of the measurement The wider the frequency span the smaller the distance that can be resolved For this reason it is generally necessary to make microwave measurements on the fixture to get sufficient resolution for analyzing the vari ous transmissions TDR Basics Using a Network Analyzer e start with broadband frequency sweep 41822 Re 50 mU REFQU often requires microwave VNA ee 1 3 d inverse FFT to compu
4. reflections due to the load standard provided we can get enough spatial resolu tion this may require the use of microwave vector network analyzers The smoother trace on the plot on the left shows the gated response of a load stan dard with a fairly typical match of about 38 dB at 1 GHz and around 30 dB at 2 GHz The right hand plot shows that the load standard looks somewhat inductive which is fairly typical It is possible to adjust our load standard to compensate for the unavoidable par asitic characteristics that degrade the reflection response Time domain gating is an excellent tool for helping to determine the proper compensation For example we see the effect in both the time and the frequency domains of adding a small capacitance to cancel out some of the inductance of the load standard 27 Connectors on Fixtures e transition at the connector launch causes reflection due to mismatch when cal standards are inserted in fixture connector match is removed when each cal standard has connectors consistency is very important Figure 20 When using PCB based fixtures performance at the connector transition is important and the consistency between connectors is critical To minimize the effect of connector mismatch when using multiple connectors on a fixture a pair for each calibration standard there must be consistency between the con nectors and their mechanical attachments to the fixture Tim
5. raw source match and load match for TRL LRM calibration The TRO calibration 254464 4 herd sees dee ee wee aa Oe WR ee Requirements for TRL standards lisse Fabricating and defining calibration standards for TRL LRM Using TDR to evaluate fixtures and standards 0 0000 00000 Bi sing active parts iss esses he eR PREMO RR PER Oe Conclusio vv 4 ek oY RY GEE YE he BRE ENE VS EEE OUEY hd OER OEE VERE Introduction The need for fixtures This application note describes the use of vector network analyzers when mak ing measurements of components in fixtures We will explain the need for fix tures the selection of fixtures measurement error how to minimize the errors basic fixture construction and the construction and characterization of required calibration standards if commercial fixtures are not available for your device Size weight and cost constraints along with higher operating frequencies and advances in technology are driving the use of smaller and more integrated pack aged parts at the assembly level Now there are many nonstandard surface mount technology SMT packages for many RF lt 3 GHz applications The physical dimensions of these parts vary greatly due to differing technologies power handling requirements environmental conditions and design criteria With the wide variety of component sizes and shapes no single fixture fits all Making quality RF measurements
6. standards that will be used with the fixture The standard definition describes the electrical characteristics delay attenua tion and impedance of each calibration standard These electrical characteris tics can be derived mathematically from the physical dimensions and material of each calibration standard or from the actual measured response A Standard Definitions table see Figure 1 lists the parameters that are used by the net work analyzer to specify the mathematical model Standard Definitions System Z Calibration Kit Label Disk File Name 4Ensure system Z of network analyzer is set to this value Open short load delay thru or arbitrary impedance Load or arbitrary impedance only 3 Arbitrary impedance only device terminating impedance Open standard types only Figure 1 Standard class assignment The standard class assignment organizes calibration standards into a format that is compatible with the error models used in measurement calibration A class or group of classes corresponds to one of seven calibration types used in the network analyzer A Standard Class Assignments table Figure 2 lists the class assignments for each standard type Standard Class Assignments Calibration Kit Label Disk File Name Standard Class Label Standard Reference Numbers Transmission Isolation TRL line or match C
7. Forward ransmission Figure 2 HP Application Note 1287 8 Applying Error Correction to Network Analyzer Measurements will provide a more in depth discussion of network analyzer basics Fixtures for R amp D versus manufacturing Fixturing in R amp D versus Manufacturing Manufacturing R amp D quick insertion alignment clamping rugged for high volume use compliant contacts usually mechanically sophisticated solder parts onto fixture ruggedness not an issue for low volumes e soldering handles leaded leadless parts often simple e g PCB with connectors Figure 3 Fixtures intended for manufacturing applications look different than those used in R amp D since the basic design goals are different In manufacturing high throughput is the overriding concern A fixture that allows quick insertion alignment and clamping is needed It must be rugged since many thousands of parts will be inserted in the fixture over its lifetime Fixtures designed for man ufacturing use tend to be mechanically sophisticated For R amp D applications the fixtures can be much simpler and less rugged They can be PCB based and since we are usually testing only a few devices we can get by with soldering parts in and out of the fixture Typical PCB Fixture with Cal Standards Load standard Short standard Contact to DUT Open standard li Thru standard Launches Transitions
8. HEWLETT U PACKARD In Fixture Measurements Using Vector Network Analyzers Application Note 1287 9 Table of contents Higingoeiuinuto REA 3 The need for fixtures 2h G25 Soa Ede uitae ES epe Ae xU eden a 3 Measurement errors 2603 ces aca h Sag ee uedmexeemaea qugacreqaugee 4 Measurement calibration 0 0 0 eee ees 4 Calibration Kit So zd ede RR 88 Ea ioe y hw AR EER ER AES TTE 5 Standard definition 0 0000 eee 5 Standard class assignment 6 60 4664 n ew ede xe eee vd 6 Fixtures for R amp D versus manufacturing 000 00200 7 Removing fixture errors 0 0 0 tee 8 Characterizing calibration standards for SOLT calibration 12 Characterizing a Short i cc ee ev lk Ped Ro ne ERR PR ee eels 13 Characterizing an Open eiieeii oue qox sede Re de Vor OY oe es Se ie VOD Y d 13 How to determine open capacitance llli 13 Characterizing a load 225224344 c eR RUE RP RERCPpS PR ER 15 Characterizing a thru 22x eee bia ino dha Oe Ohi wm qxenmeeoshdda Dd dyes 16 TRL LRM calibration pes ei gesiene eee n 17 TRIG terminology 5 2 06 9 04 aput Gee eed e pe Se Gea WR aay ee ede Snes 17 How TRL LRM calibration works 0 0 0 0 000 0 eee 17 TRE errormodel 22 22 1 054p erREe Bhan be eru ee er s 17 USOT AULOM EAM deter eee dts oats deme eee eee ee 18 Source match and load match 2 0 0 0 000 ee 18 How true TRL LRM works four sampler receiver architecture only 19 Improving
9. Non zero the same the average impedance is used length Attenuation of the thru need not be known If the thru is used to set the reference plane the insertion phase or electrical length must be well known and specified If a non zero length thru is specified to have zero delay the reference plane is established in the middle of the thru 20 Types REFLECT LINE MATCH LINE LINE MATCH MATCH Requirements continued Reflection coefficient T magnitude is optimally 1 0 but need not be known Phase of I must known and specified to within VA wavelength or 90 During computation of the error model the root choice in the solution of a quadratic equation is based on the reflection data An error in definition would show up as a 180 error in the measured phase I must be identical on both ports If the reflect is used to set the reference plane the phase response must be well known and specified Zo of the line establishes the reference impedance of the mea surement i e S11 S s 0 The calibration impedance is defined to be the same as Zo of the line If the Zo is known but not the desired value i e not equal to 50 Q the SYS TEMS Zp selection under the TRL LRM options menu is used Insertion phase of the line must not be the same as the thru zero length or non zero length The difference between the thru and line must be between 20 and 160 n x 180 Measurement uncertainty
10. Source match and load match The first step in the TRL two port calibration process is the same as the trans mission step for a full two port calibration For the thru step the test ports are connected together directly zero length thru or with a short length of trans mission line non zero length thru and the transmission frequency response and port match are measured in both directions by measuring all four S para meters For the reflect step identical high reflection coefficient standards typically open or short circuits are connected to each test port and measured S11 and 22 For the line step a short length of transmission line different in length from the thru is inserted between port I and port 2 and again the frequency response and port match are measured in both directions by measuring all four S parameters In total 10 measurements are made resulting in 10 independent equations However the TRL error model has only eight error terms to solve for The characteristic impedance of the line standard becomes the measurement refer ence and therefore has to be assumed ideal or known and defined precisely At this point the forward and reverse directivity Epp and Epp transmission tracking Erp and Erp and reflection tracking Epp and Epp terms may be derived from the TRL error terms This leaves the isolation Exp and Exp source match Esr and Egg and load match Hy and Ej terms to discuss Two addit
11. analyzer user guide 29 Biasing active parts Making in fixture measurements of active parts requires that DC bias be sup plied along with the RF signal Traditionally when bias was needed for testing transistors external bias tees were used in the main RF signal paths This approach is still valid today although internal bias tees are provided by most vector network analyzers Biasing Active Parts e can use bias tees if RF and DC share same line many network analyzers contain internal bias tees if separate fixture needs extra connectors pins or wires proper bypassing is important to prevent oscillation Figure 23 Many packaged amplifiers and RFICs require that DC power be supplied on separate pins This means that the fixture must provide extra connectors DC feedthroughs wires or pins for the necessary bias These bias connections should present a low DC impedance Discrete elements can be placed directly on the fixture near the DUT to provide proper RF bypassing and isolation of the DC supply pins Good RF bypassing techniques can be essential as some ampli fiers will oscillate if RF signals couple onto the supply lines 30 Transistor Bias Example to port one n bias tee ama Collector current Risse Helier monitor 10K Q ooo 100 9 Vbase 4 Vcollector 4 BAe Collector voltage monitor Two port calibration was perfo
12. ance of the thru should match the impedance of the transmission lines used with the other standards all of which should be 50 ohms Notice in figure 12 the PC board is wider for the transmission line where the DUT will be soldered Since we want the two halves of line to be equal in elec trical length to the thru line the PCB must be widened by the length of the DUT Thru Standard DUT placed here thru e thru is a simple transmission line desire constant impedance and minimal mismatch at ends PCB is widened by the length of the DUT to insure that both lines are of equal length Figure 14 With a properly designed PC board fixture the short or open defines a cali bration plane to be in the center of the fixture This means the thru will have a length of zero which is usually not the case for fixtures used in manufacturing applications where a set of calibration standards is inserted into a single fix ture Since the length is zero we do not have to worry about characterizing the loss of the thru or its phase shift 16 TRL LRM calibration TRL terminology How TRL LRM calibration works TRL error model Notice that the letters TRL LRL LRM TRM are often interchanged depending on the standards used For example LRL indicates that two lines and a reflect standard are used TRM indicates that a thru reflection and match standards are used All of these refer to the same bas
13. ange Copyright O 1999 Hewlett Packard Company Printed in U S A 5 99 5968 5329E 32
14. atch is especially troublesome for low loss transmission measurements such as measuring a filter passband or a cable and for reflection measure ments Using response calibration for transmission measurements on low loss devices can result in considerable measurement uncertainty in the form of rip ple Measurement accuracy will depend on the relative mismatch of the test fixture in the network analyzer compared to the DUT When measuring transmission characteristics with fixtures considerable mea surement accuracy improvement can be obtained by performing a two port correction at the ends of test cables This calibration improves the effective source and load match of the network analyzer thus helping to reduce the mea surement ripple the result of reflections from the fixture and analyzer s test ports 10 Two Port Calibration Two port calibration corrects for all major sources of systematic measurement errors R A Directivity Crosstalk MU Um IN x Ag pur fy Frequency response pu ems reflection tracking A R Source Load transmission tracking B R Mismatch Mismatch Six forward and six reverse error terms yields 12 error terms for two port devices Figure 9 Two port calibration provides much more accurate measurements compared to a response calibration It also requires more calibration standards There are two basic types of two port calibration Short Open Load Thru SOLT and the Thru Re
15. ation er TIPS e at end of test cable measure load store data in Pad memory display data mem Del measure short add port extension until flat 180 phase measure open read capacitance from admittance Smith chart enter capacitance coefficient s in cal kit definition of open START 050 000 000 TOP 6 000 000 000 GHz Hz watch out for negative capacitance due to long or inductive short adjust with negative offset delay in open or positive offset delay in short Figure 11 Determining the fringing capacitance is only necessary above approximately 300 MHz The fringing capacitance can be measured as follows 1 Perform a one port calibration at the end of the test cable Use a connector type that is compatible with the fixture For example use APC 3 5 mm standards for a fixture using SMA connectors 2 Connect the fixture and measure the load standard This data should be stored in memory and the display changed to data minus memory This step subtracts out the reflection of the fixture connector assuming good consistency between connectors so that we can characterize just the open An alternative is to use time domain gating to remove the effect of the connector 13 3 Measure the short standard Set the port extension to get a flat 180 degrees phase response To fine tune the value of port extension set the phase off set value for the trace to 180 degrees and expand degrees p
16. ation solves for forward error terms directivity source match and reflection tracking Likewise the S22 1 PORT calibration solves for the same terms in the reverse Full 2 PORT and TRL 2 PORT calibrations include forward and reverse error terms of both ports plus transmission tracking and isolation Calibration kit Standard definition The type of measurement calibration selected by the user depends on the device to be measured for example one port or two port device and the extent of accuracy enhancement desired Further a combination of calibra tions can be used in the measurement of a particular device The accuracy of subsequent DUT measurements is dependent on the accuracy of the test equipment how well the known devices are modeled and the exact ness of the error correction model Measurement accuracy is largely dependent upon calibration standards and a set of calibration standards is often supplied as a calibration kit Each standard has precisely known or predictable magnitude and phase response as a function of frequency For the network analyzer to use the standards of a calibration kit the response of each standard must be mathematically defined and then organized into a standard class that corresponds to the error model used by the network analyzer Hewlett Packard currently supplies calibration kits for most coaxial components However when measuring non coaxial components it is necessary to create and define the
17. because some reflection always occurs at some fre quency especially with non coaxial actual standards At RF we can build a good load using standard surface mount resistors Usually it is better to use two 100 ohm resistors in parallel instead of a single 50 ohm resistor because the parasitic inductance is cut in half For example 0805 size SMT resistors have about 1 2 nH series inductance and 0 2 pF parallel capacitance Two parallel 100 ohm 0805 resistors have nearly a 20 dB better match than a single 50 ohm resistor 15 Characterizing a thru Load Standard CH Sil 5dB REFOdB CH2 MEM log MAG 5dB REFOdB two 100 ohm Paii resistors Cor One 50 ohm SMT resistor 1 24 229dB 1 GHz 2 14 792 dB 3 GHz 2 1 Two 100 ohm SMT resistors PRm SEIT Cor 1 41 908 dB 1GHz 2 32 541 dB 3GHz START 300000 MHz STOP 6 000 000 000 MHz ideal zero reflection at all frequencies can only approximate at best usually somewhat inductive two 100 ohm resistors in parallel better than a single 50 ohm resistor Figure 13 The thru standard is usually a simple transmission line between two coaxial connectors on the fixture A good thru should have minimal mismatch at the connector launches and maintain a constant impedance over its length which is generally the case for PCB thrus The imped
18. degrees phase reference The electrically shorter open will then appear to have positive phase The remedy for this is to decrease the port extension until the phase is monoto nically negative The model for the open will then have a normal positive capacitance value The value of the negative offset delay that needs to be included in the open standard definition is simply the amount by which port extension was reduced for instance the difference in the port extension values between the short and the open In effect we have now set the reference plane at the short Alternatively the offset delay of the open can be set to zero and a small positive offset delay can be added to the model of the short stan dard This will set an effective reference plane at the open 14 Characterizing a load Port Extensions port extension feature of network analyzer removes linear portion of phase response accounts for added electrical length of fixture doesn t correct for loss or mismatch mismatch can occur from e launches e variations in transmission line impedance After port extensions Fixture response without applied fixture phase port extensions response is flat Phase 45 Div Frequency Frequency Figure 12 An ideal load reflects none of the incident signal thereby providing a perfect termination over a broad frequency range We can only approximate an ideal load with a real termination
19. e domain measure ments are useful for analyzing both connector performance and repeatability See Figure 21 28 Connector Performance CH1 S11 log MAG 10dB REFOdB 1900 GHz bal EA right angle 1_ 23 753 dB PRM connector 1_ 32 297 dB Cor Ht pum mi rA edge connector frequency domain Gat edge connector with gap CH1 START 099 751 243 GH STOP 20 049 999 843 GH Comparing match CH2 S11 R s U REFOU 1 996 mU efrightangle and z mW pU edge mount connectors PRm euge connectat e right angle with and without gap with gap connector Cor a ar ence nid time domain X edge connector CH2 START 500 ps STOP 1 ns Figure 21 Connector Consistency CH1 Sq1 M log MAG 5 dB REF 10 dB lal PRm right angle 1 900 GHz Cor connector 1_ 33 392 dB 1 43 278 dB frequency domain Gat ge c nnecto CH1 START 099 751 243 GHz STOP 20 049 999 843 GHz Use data memory CH2 S M Re 49 6 mU REF50mU 1 80261 mU to check consistency of connectors PRm Cor 1 M time domain Gat d right angle connector CH2 START 500 ps STOP 1 ns Figure 22 For information on making time domain measurements and using the gating feature please see your network
20. ect standard The loss term must also be specified The line standard must meet specific frequency related criteria in conjunction with the length used by the thru standard In particular the insertion phase of the line must not be the same as the thru The optimal line length is 1 4 wave length 90 degrees relative to a zero length thru at the center frequency of interest and between 20 and 160 degrees of phase difference over the frequen cy range of interest Note these phase values can be N x 180 degrees where N is an integer If two lines are used LRL the difference in electrical length of the two lines should meet these optimal conditions Measurement uncertain ty will increase significantly when the insertion phase nears zero or is an inte ger multiple of 180 degrees and this condition is not recommended For a transmission media that exhibits linear phase over the frequency range of interest the following expression can be used to determine a suitable line length of 14 wavelength at the center frequency which equals the sum of the start frequency and stop frequency divided by 2 Electrical length cm LINE 0 length THRU Electrical length cm _ 15000 x VF fl MHz f2 MHz let fl 1000 MHz f2 2000 MHz VF Velocity Factor 1 for this example Thus the length to initially check is 5 cm 22 Next use the following to verify the insertion phase at fl and f2 Phase degrees 860 x
21. er division scale Mismatch and directivity reflections may cause a slight ripple so use your best judgment for determining the flattest trace or use marker statistics set the mean value to zero 4 Set the network analyzer display format to Smith chart the marker function to Smith chart format G jB admittance and then measure the open stan dard Markers now read G jB instead of the R jX of an impedance Smith chart Admittance must be used because the fringing capacitance is modeled as a shunt element not a series element The fringing capacitance typically 0 03 0 25 pF can be directly read at the frequency of interest using a trace marker At RF a single capacitance value Co is generally adequate for the calibration kit definition of the open In some cases a single capacitance number may not be adequate as capacitance can vary with frequency This is typically true for the measurements that extend well into the microwave frequency range Because capacitance varies with frequency at fre quencies above 3 GHz it may be better to use a TRL LRM calibra tion When measuring the fringing capacitance a problem can arise if the short stan dard is electrically longer than the open standard The measured impedance of the open circuit then appears to be a negative capacitor indicated by a trace that rotates backwards counter clockwise on the Smith chart This problem is a result of using an electrically longer short standard as the 180
22. er user manual A shareware program that simplifies the process of modifying calibration kit coefficients is available at www vnahelp com LA eackana For more information about Hewlett Packard test and measure ment products applications services and for a current sales office listing visit our web site http www hp com go tmdir You can also contact one of the following centers and ask for a test and measurement sales representative United States Hewlett Packard Company Test and Measurement Call Center P O Box 4026 Englewood CO 80155 4026 1 800 452 4844 Canada Hewlett Packard Canada Ltd 5150 Spectrum Way Mississauga Ontario L4W 5G1 905 206 4725 Europe Hewlett Packard European Marketing Centre P O Box 999 1180 AZ Amstelveen The Netherlands 81 20 547 9900 Japan Hewlett Packard Japan Ltd Measurement Assistance Center 9 1 Takakura Cho Hachioji Shi Tokyo 192 Japan Tel 81 426 56 7832 Fax 81 426 56 7840 Latin America Hewlett Packard Latin American Region Headquarters 5200 Blue Lagoon Drive 9th Floor Miami Florida 33126 U S A 305 267 4245 4220 Australia New Zealand Hewlett Packard Australia Ltd 31 41 Joseph Street Blackburn Victoria 3130 Australia 1 800 629 485 Asia Pacific Hewlett Packard Asia Pacific Ltd 17 21 F Shell Tower Times Square 1 Matheson Street Causeway Bay Hong Kong Tel 852 2599 7777 Fax 852 2506 9285 Data Subject to Ch
23. f x 1 V where f frequency l length of line v velocity speed of light x velocity factor which can be reduced to the following using frequencies in MHz and length in centimeters Phase degrees approx 0 012 x f MHz x cm VF So for an air line velocity factor approximately 1 at 1000 MHz the insertion phase is 60 degrees for a 5 cm line it is 120 degrees at 2000 MHz This line would be a suitable line standard For microstrip and other fabricated standards the velocity factor is significant In those cases the phase calculation must be divided by that factor For exam ple if the dielectric constant for a substrate is 10 and the corresponding effec tive dielectric constant for microstrip is 6 5 then the effective velocity factor equals 0 39 1 square root of 6 5 Using the first equation with a velocity factor of 0 39 the initial length to test would be 1 95 cm This length provides an insertion phase at 1000 MHz of 60 degrees at 2000 MHz 120 degrees the insertion phase should be the same as the air line because the velocity factor was accounted for when using the first equation Another reason for showing this example is to point out the potential problem in calibrating at low frequencies using TRL For example 1 4 wavelength is Length cm _7500 x VF Sc where Je center frequency Thus at 50 MHz Length cm 7500 150 cm or 1 5m 50 MHz 23 Such a line sta
24. flect Line TRL These are named after the types of standards used in the calibration process A calibration at the coaxial ports of the network analyzer removes the effects of the network analyzer and any cables or adapters before the fixture however the effects of the fixture itself are not accounted for An in fixture calibration is preferable but high quality SOLT standards are not readily available to allow a conventional full two port calibration of the system at the desired measurement plane of the device In microstrip a short circuit is inductive an open circuit radiates energy and a high quality purely resistive load is difficult to produce over a broad frequency range The TRL two port calibration is an alternative to the traditional SOLT Full two port calibration technique that utilizes simpler more convenient standards for device measurements in the microstrip environ ment In all measurement environments the user must provide calibration standards for the desired calibration to be performed The advantage of TRL is that only three standards need to be characterized as opposed to four in the traditional SOLT full two port calibrations Further the requirements for characterizing the T R and L standards are less stringent and these standards are more easily fabricated For more information on network analyzer calibrations please see HP Application Note 1287 83 Applying Error Correction to Network Analyzer Measurements
25. ic method The TRL LRM calibration is used in a network analyzer with a three sampler receiver architecture and relies on the characteristic impedance of simple transmission lines rather than on a set of discrete impedance standards Since transmission lines are relatively easy to fabricate in a microstrip for example the impedance of these lines can be determined from the physical dimensions and substrate s dielectric constant 8 Term TRL Model 1001 ERF go EDF E33 EDR 11 Esp ELR 522 Esp ELF 10 E32 ETF E01 23 ETR 8 term TRL error model and generalized coefficients Figure 15 For TRL two port calibration a total of 10 measurements are made to quantify eight unknowns not including the two isolation error terms Assume the two transmission leakage terms Ex and Exp are measured using the conventional technique Although this error model is slightly different from the traditional Full two port 12 term model the conventional error terms may be derived from it For example the forward reflection tracking Epp is represented by the product of 4 and 9 Also notice that the forward source match Ege and reverse load match Ej are both represented by while the reverse source match Egg and forward load match Ej are both represented by 3 In order to solve for these eight unknown TRL error terms eight linearly inde pendent equations are required 17 Isolation
26. ional measurements are required to solve for the isolation terms Exe and Exp Isolation is characterized in the same manner as the full two port calibration Forward and reverse isolation are measured as the leakage or crosstalk from port 1 to port 2 with each port terminated The isolation part of the calibration is generally only necessary when measuring high loss devices greater than 70 dB Note If an isolation calibration is performed the fixture leakage must be the same during the isolation calibration and the measurement A TRL calibration assumes a perfectly balanced test set architecture as shown by the 4 term which represents both the forward source match Esp and reverse load match Ezp and by the 59 term which represents both the reverse source match Egg and forward load match E p However in any switching test set the source and load match terms are not equal because the transfer switch presents a different terminating impedance as it is changed between port 1 and port 2 18 How true TRL LRM works four sampler receiver architecture only Improving raw source match and load match for TRL LRM calibration For network analyzers that are based on a three sampler receiver architecture it is not possible to differentiate the source match from the load match terms The terminating impedance of the switch is assumed to be the same in either direction Therefore the test port mismatch cannot be f
27. librate with the external bias tees in place no bias applied during cal ibration to remove their effects from the measurement Because the bias tees must be placed after the attenuators they essentially become part of the fixture Therefore their mismatch effects on the measure ment will not be improved by the attenuators Although the fixed attenuators improve the raw mismatch of the network ana lyzer system they also degrade the overall measurement dynamic range This effective mismatch of the system after calibration has the biggest effect on reflection measurements of highly reflective devices Likewise for well matched devices the effects of mismatch are negligible This can be shown by the fol lowing approximation Reflection magnitude uncertainty Ep EpS Es 8 ELS21Sp Transmission magnitude uncertainty Ex E485 E33 989 Ey S059 Where Ep effective directivity Eg effective reflection tracking Es effective source match E effective load match E effective crosstalk Er effective transmission tracking S 4 S parameters of the device under test When building a set of TRL standards for a microstrip or fixture environment the requirements for each of these standard types must be satisfied Types Requirements THRU No loss Characteristic impedance Zo need not be known Zero length So Sy 1 Z 0 S11 S22 0 THRU Zo of the thru must be the same as the line if they are not
28. modify the corresponding source and load match terms for both forward and reverse The four sampler receiver architecture configuration with TRL establishes a higher performance calibration method over TRL when making in fixture mea surements because all significant error terms are systematically reduced With TRL the source and load match terms are essentially those of the raw uncor rected performance of the hardware A technique that can be used to improve the raw test port mismatch is to add high quality fixed attenuators as closely as possible to the measurement plane The effective match of the system is improved because the fixed attenuators usually have a return loss that is better than that of the network analyzer Additionally the attenuators provide some isolation of reflected signals The attenuators also help to minimize the difference between the port source match and load match making the error terms more equivalent With the attenuators in place the effective port match of the system is improved so that the mismatch of the fixture transition itself dominates the measurement errors after a calibration 19 The TRL calibration Requirements for TRL standards If the device requires bias it will be necessary to add external bias tees between the fixed attenuators and the fixture The internal bias tees of the ana lyzer will not pass the bias properly through the external fixed attenuators Be sure to ca
29. ndard would not only be difficult to fabricate but its long term stability and usability would be questionable as well Thus at lower frequencies and or very broad band measurements fabrication of a match or termination may be deemed more practical Since a termina tion is in essence an infinitely long transmission line it fits the TRL model mathematically and is sometimes referred to as a TRM calibration The TRM calibration technique is related to TRL with the difference being that it bases the characteristic impedance of the measurement on a matched Zo termination instead of a transmission line for the third measurement stan dard Like the TRL thru standard the TRM THRU standard can either be of zero length or non zero length The same rules for thru and reflect standards used for TRL apply for TRM TRM has no inherent frequency coverage limitations which makes it more convenient in some measurement situations Additionally because TRL requires a different physical length for the thru and the line standards its use becomes impractical for fixtures with contacts that are at a fixed physical distance from each other For more information on how to modify calibration constants for TRL LRM and how to perform a TRL or LRM calibration refer to the Optimizing Measurement Results in the network analyzer users manual 24 Using TDR to evaluate fixtures and standards Time domain reflectometry TDR is a helpful tool
30. on devices with standard coaxial connectors is relatively easy Very accurate measurements can be made using commercial calibration kits and standard error correction routines found in most network analyzers Devices without connectors are difficult to measure since some sort of test fixture is required to provide electrical and mechanical connection between the device under test DUT and the coaxial connector based test equipment In addition in fixture calibration standards are often required to achieve the level of measurement accuracy demanded by many of today s devices An ideal fixture would provide a transparent connection between the test instrument and the device being tested It would allow direct measurement of the DUT without imposition of the fixture s characteristics In parametric terms this would mean the fixture would have no loss a flat frequency response with linear phase no mismatches be a precisely known electrical length and have infinite isolation between input and output zero crosstalk If we could make such a fixture calibration would be unnecessary Since it is impossible to make an ideal fixture we can only approximate the ideal case We need to do this by optimizing the performance of the test fixture relative to the performance of the DUT We can try to make the loss of the fix ture smaller than the specified gain or insertion loss uncertainty of the DUT The bandwidth of the fixture needs to be wider than
31. reflections of the calibration standards or just the calibration standards by themselves Figure 18 shows the performance of a thru standard used in a fixture intended for manufacturing use The time domain plot on the left shows significant mis match at the input and output of the thru The plot on the right shows perfor mance of the thru in the frequency domain with and without gating We see about a 7 dB improvement in return loss at 947 MHz using time domain gat ing resulting in a return loss for the thru of about 45 dB The gated measure ment provides a more accurate characterization of the thru standard 26 Characterizing and Adjusting Load CH1 S11 amp M log MAG 5dB REFOdB PRm a C 1 38805 dB 947 MHz load mismatch _ load in frequency domain due to inductance Gate with and without gating CH1Sq1 Re 100 mJ REF 0 U n PRm al 2 Cor 1 61 951 mU 707 ps sd i 3 2 159 74 mU 749 ps ART 050 GHz STOP 6 000 GHz use time domain gating to see load reflections independent from fixture use time domain to compensate for imperfect load e g try to cancel out inductance START 5ns STOP 1 5 ns Figure 19 Time domain gating can be a very useful tool for evaluating how well the load is performing We can gate out the response of the fixture and just look at the
32. rform a full two port calibration at the points indicated in figure 5 This calibration establishes the reference plane at the junction of the test port cables The fixture is then connected to the test port cables and the reference plane is then mathematically adjusted to the DUT using the port extension feature of the network analyzer If the fixture performance is consid erably better than the specifications of the DUT this technique may be suffi cient De embedding Two port calibration De embedding Accurate S parameter data e requires external software from model or measurement accuracy is determined by quality of fixture model Figure 6 De embedding requires an accurate linear model of the fixture or measured S parameter data of the fixture External software is needed to combine the error data from a calibration done without the fixture using coaxial standards with the modeled fixture error If the error terms of the fixture are generated solely from a model the overall measurement accuracy depends on how well the actual performance of the fixture matches the modeled performance For fixtures that are not based on simple transmission lines determining a precise model is usually harder than using the direct measurement method Direct Measurement 44 Various calibration standards lt Measurement plane for cal standards and DUT e measure standards to determine systematic e
33. ribe calibration standards only a few need to be modified for most fixture applications For a properly designed PCB fixture only the fringing capacitance of the open standard and the delay of the short need to be characterized 12 Characterizing a short Characterizing an open How to determine open capacitance The electrical definition of an ideal short is unity reflection with 180 degrees of phase shift All of the incident energy is reflected back to the source perfectly out of phase with the reference A simple short circuit from a single conductor to ground makes a good short standard For example the short can be a few vias plated through holes to ground at the end of a micro strip transmission line If coplanar transmission lines are used the short should go to both ground planes To reduce the inductance of the short avoid excessive length A good RF ground should be near the signal trace If the short is not exactly at the contact plane of the DUT an offset length can be entered in terms of electrical delay as part of the user defined calibration kit The open standard is typically realized as an unterminated transmission line Electrical definition of an ideal open has unity reflection with no phase shift The actual model for the open however does have some phase shift due to fringing capacitance Determining Open Capacitance 1 228 23 uS 1 2453 mS 209 29 fF 947 000 e perform one port calibr
34. rmed prior to taking S parameter data of the transistor Figure 24 This is an example of how bias could be supplied to a transistor The power supplies are not shown but they would be connected to the V base and the V collector nodes The V base controls the collector current and V collec tor controls the collector to emitter voltage on the transistor For the base resistors it is important to use a fairly large value such as a 10K ohms so that the voltage adjustment is not too sensitive You may find it convenient to use two digital voltmeters to monitor the collector current and collector to emitter voltage simultaneously 31 Conclusion We have covered the principles of in fixture testing of components with vector network analyzers It is time to determine the source of the fix ture Is the fixture available commercially or must it be designed and built Inter Continental Microwave is a Hewlett Packard Channel Partner experienced in designing and manufacturing test fixtures that are com patible with Hewlett Packard network analyzers Inter Continental Microwave contact information Inter Continental Microwave 1515 Wyatt Drive Santa Clara Ca 95054 1586 Tel 408 727 1596 Fax 408 727 0105 Fax on Demand 408 727 2763 Internet www icmicrowave com If it is necessary to design and build the fixture more information on calibration kit coefficient modification can be found in the appropriate network analyz
35. rrors Various e two major types of calibrations calibration response normalization calibration standards two port calibration vector error correction short open load thru SOLT e thru reflect line TRL Figure 7 Direct measurement usually involves measuring physical calibration standards and calculating error terms This method is based on how precisely we know the characteristics of our calibration standards The number of error terms that can be corrected varies considerably depending on the type of calibration used Normalization only removes one error term while full two port error correction accounts for all 12 error terms Direct measurements have the advantage that the precise characteristics of the fixture do not need to be known beforehand They are measured during the calibration process The simplest form of direct measurement is a response cal ibration which is a form of normalization A reference trace is placed in memo ry and subsequent traces are displayed as data divided by memory A response calibration only requires one standard each for transmission a thru and reflec tion a short or open Response Calibration PCM 5 ne ee C doo gt Reference X errors due to mismatch Figure 8 However response calibration has a serious inherent weakness due to the lack of correction for source and load mismatch and coupler bridge directivity Mism
36. source of measurement uncertainty The six systematic errors in the forward direction are directivity source match reflection tracking load match transmission tracking and isolation The reverse error model is a mirror image giving a total of 12 errors for two port measurements Calibration is the process for removing these errors from net work analyzer measurements Measurement A more complete definition of measurement calibration using the network ana calibration lyzer and a description of error models are included in the network analyzer operating manual The basic ideas are summarized here Measurement calibration is a process in which a network analyzer measures precisely known devices and stores the vector differences between the mea sured and the actual values The error data is used to remove the systematic errors from subsequent measurements of unknown devices There are six types of calibrations available with the vector network analyzer response response amp isolation S11 1 PORT S22 1 PORT FULL 2 port and TRL 2 PORT Each of these calibration types solves for a different set of systematic measurement errors A RESPONSE calibration solves for the systematic error term for reflection or transmission tracking depending on the S parameter that is activated on the network analyzer at the time of the calibration RESPONSE amp ISOLATION adds correction for crosstalk to a simple RESPONSE calibration An S11 1 PORT cal ibr
37. te time domain _ Lj resolution inversely proportionate to frequency span CH1 START 0 s STOP 1 5 ns Figure 17 For example it may be necessary to measure a fixture designed for use at 3 GHz with a frequency span of 0 05 GHz to 20 GHz or even 40 GHz to get the needed resolution 25 Time Domain Gating TDR and gating can remove undesired reflections only useful for broadband devices a load or thru for example and broadband fixture define gate to only include DUT use two port calibration CH1 Sj1 amp Mlog MAG 5dB REFOdB at ends of test cables va PRm Cor CH1 MEM Re 20 mU REF OU al fan L 1 48 729 mU 638 ps ae 2 24 961 hU 668 ps Gate 1 45 113 dB 0 947 GHz 8 992 mm 3 10 891 mU 721 ps 2 15 78 dB 6 000 GHz thru in time domain thru in frequency domain with and without gating CH1 START Os STOP 1 5 ns START 050 000 000 GHz STOP 20 050 000 000 GHz Figure 18 As long as we have enough spatial resolution we can see the reflections of the connector independently of the reflections of the calibration standards With time domain we can isolate various sections of the fixture and see the effects in the frequency domain For example we can choose to look at just the con nector launches without interference from the
38. the desired measurement bandwidth of the DUT Mismatch can be minimized with good design and the use of effective measurement tools such as time domain reflectometry TDR to identify the mismatches in the fixture The electrical length of the fixture can be measured Fixture crosstalk need only be less than the isolation of the device under test Since we can only approximate the perfect fixture the type of calibration required for any particular application will depend solely on how stringent the DUT specifications are Before we discuss calibration we need to briefly discuss what factors con tribute to measurement uncertainty Measurement errors Errors in network analyzer measurements can be separated into three categories Drift errors occur when the test system s performance changes after a calibra tion has been performed They are primarily caused by temperature variation and can be removed by recalibration Random errors vary as a function of time Since they are not predictable they cannot be removed by calibration The main contributors to random errors are instrument noise switch repeatability and connector repeatability The best way to reduce random errors is by decreasing the IF bandwidth or by using trace averaging over multiple sweeps Systematic errors include mismatch leakage and system frequency response In most microwave or RF measurements systematic errors are the most signifi cant
39. ully corrected An assumption is made that forward source match Esp reverse load match Eip 11 reverse source match Esp forward load match Eip 25 For a fixture TRL can eliminate the effects of the fixture s loss and length but does not completely remove the effects due to the mismatch of the fixture This is in contrast to the pure TRL technique used by instruments equipped with four sampler receiver architecture Note Because the TRL technique relies on the characteristic impedance of transmission lines the mathematically equivalent method LRM for line reflect match may be substituted for TRL Since a well matched termination is in essence an infinitely long transmission line it is well suited for low RF frequency calibrations Achieving a long line standard for low frequencies is often physically impossible The TRL implementation with four sampler receiver architecture requires a total of 14 measurements to quantify 10 unknowns as opposed to only a total of 12 measurements for TRL Both include the two isolation error terms Because of the four sampler receiver architecture additional correction of the source match and load match terms is achieved by measuring the ratio of the two reference receivers during the thru and line steps These measurements characterize the impedance of the switch and associated hardware in both the forward and reverse measurement configurations They are then used to
40. will increase significantly when the insertion phase nears 0 or an integer multiple of 180 Optimal line length is V4 wavelength or 90 of insertion phase relative to the thru at the middle of the desired frequency span Usable bandwidth for a single thru line pair is 8 1 frequency span start frequency Multiple thru line pairs Zg assumed identical can be used to extend the bandwidth to the extent transmission lines are available Attenuation of the line need not be known Insertion phase must be known and specified within 1 4 wavelength or 90 Zo of the match establishes the reference impedance of the measurement T must be identical on both ports 21 Fabricating and defining calibration standards for TRL LRM When calibrating a network analyzer the actual calibration standards must have known physical characteristics For the reflect standard these characteristics include the offset in electrical delay seconds and the loss ohms second of delay The characteristic impedance OFFSET Zo is not used in the calcula tions because it is determined by the line standard The reflection coefficient magnitude should optimally be 1 0 but need not be known since the same reflection coefficient magnitude must be applied to both ports The thru standard may be a zero length or known length of transmission line The value of length must be converted to electrical delay just as for the refl
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